CA1106924A - Signal transmission circuit - Google Patents
Signal transmission circuitInfo
- Publication number
- CA1106924A CA1106924A CA314,249A CA314249A CA1106924A CA 1106924 A CA1106924 A CA 1106924A CA 314249 A CA314249 A CA 314249A CA 1106924 A CA1106924 A CA 1106924A
- Authority
- CA
- Canada
- Prior art keywords
- circuit
- sampling
- signal
- signal transmission
- transmission circuit
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
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Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
- H04N25/00—Circuitry of solid-state image sensors [SSIS]; Control thereof
- H04N25/40—Extracting pixel data from image sensors by controlling scanning circuits, e.g. by modifying the number of pixels sampled or to be sampled
- H04N25/44—Extracting pixel data from image sensors by controlling scanning circuits, e.g. by modifying the number of pixels sampled or to be sampled by partially reading an SSIS array
- H04N25/447—Extracting pixel data from image sensors by controlling scanning circuits, e.g. by modifying the number of pixels sampled or to be sampled by partially reading an SSIS array by preserving the colour pattern with or without loss of information
Abstract
ABSTRACT OF THE DISCLOSURE
A signal transmitting circuit is disclosed, wherein a sample-hold circuit, a low pass filter and a sampling circuit are connected in series in the signal transmission line.
Assuming that a sampling frequency at both the sample-hold circuit and sampling circuit is fc, a cut-off frequency of the low pass filter is preferably selected about the same as the sampling frequency fc. This causes the compensation of the frequency response in the transmission line, and particularly the circuit is applicable to a CCD (charge-coupled device) image read-out circuit. The output of the CCD is something like a sampled signal, so that the application of this system to the CCD signal is very normal and useful.
A signal transmitting circuit is disclosed, wherein a sample-hold circuit, a low pass filter and a sampling circuit are connected in series in the signal transmission line.
Assuming that a sampling frequency at both the sample-hold circuit and sampling circuit is fc, a cut-off frequency of the low pass filter is preferably selected about the same as the sampling frequency fc. This causes the compensation of the frequency response in the transmission line, and particularly the circuit is applicable to a CCD (charge-coupled device) image read-out circuit. The output of the CCD is something like a sampled signal, so that the application of this system to the CCD signal is very normal and useful.
Description
~1~692~
BACKGRO~ OF T~E INVENTION
Field of the Invention:
.
The present invention relates generally to a signal transmission circuit in which a plurality of sampling circuits are interposed, ~nd is directed more particularly to such a signal transmission circuit suitable for use in the signal processing circuit of a solid state television camera in T~7hich a semiconductor device such as a CCD and a BBD (bucket brigade device) is employed.
Descri tion of the Prior Art:
P
By way of example, a prior art solid state television camera using a CCD imager will be now considered. With such a prior art television camera, the output signal from an imaging device is passed through a circuit consisting of a sampling circuit and/or sample-hold circuit connected in cascade to be a final signal. In this case, it is noted that the signal is apt to be deteriorated by the sampling action. This may be considered from the fact that minus components lying above the sampling frequency, which is generated by the first sample-hold action, cancel out effective signal components existing lower than the sampling frequency by the second sampling action, which will be described later.
Fundamentally speaking, the output from a solid state image sensor is a sampled type signal and low in signal level, so that even a little signal deterioration can not be neglected.
In.a charge transfer device such as a CCD, there are many cases where the sample-hold function is requi.red in deriving a signal from the device itself. Accordinglv, the sampling action, which is carried out once after the signal is derived, deteriorates the rrequency response of the solid state imaging syste~ in the $2~
prior art system.
OBJECTS Al~ SUMMARY OF THE INV~NTION
An object of the present invention is to provide a ;
signal transmission circuit having a sampling and/or sample-hold circuit.
Another object of the invention is to provide a novel signal transmission circuit in which circuits with the sampling function are connected in series.
According to the fundamental construction of the inven-tion, a sample-hold circuit, a low pass filter and a sampling circuit (or may be a sample-hold circuit) are connected in series. A sampling pulse train applied to the respective cir-cuits are the same in frequency but may be different in phase and window width.
In the usual manner, the sample-hold circuit shows low pass filter characteristics to a signal to be transmitted therethrough. In this invention, such low pass filter char-acteristics of the sample-hold circuit are able to be cancelled or compensated for by interposing a low pass filter having a cut-off frequency of about the sampling frequency. In this case, the last connected sampling circuit determines the phase of an output signal. Accordingly, by adjusting the window phase and window width of the sampling pulse applied to the last connected sampling circuit, the phase of the output signal from the CCD
imaging device is determined.
Therefore, the signal transmission circuit of the invention can be applied to an image pick-up system which uses a plurality of image sensor chips to improve its resolution.
In such a system it is required that the phase of an output signal is arranged in response to a chip displacement. When . ~
:
the present invention is applied to such a phase arrangement of the output signal, the phase arrangement of the output signal can be performed while signal compensation is achieved.
The additional, and other objects, features and ad-vantages of the present invention will become clear from the following description taken in conjunction with the accompanying drawings through which the like reference numerals and letters indicate the same elements.
BRI~F DESCRIPTION OF THE DRAWINGS
`, Figure 1 is a schematic diagram showing the arrange-ment relation of imaging elements;
Figure 2 is a graph showing the frequency character-istics of a picked-up output;
Figure 3 is a systematic block diagram showing an image sensing apparatus;
Figures 4A to 4G are diagrams used for explaining a sampling operation;
Figures 5A and SB are diagrams used for the explana-tion of the operation of a prior circuit;
Figure 6 is a block diagram showing the fundamental construction of the present invehtion;
Figures 7A, 7B and 7C are diagrams showing the fre-quency characteristics thereof;
Figures 8 and 9 are respectively block diagrams showing solid state television cameras to which the invention is applied;
Figures 10 A to 10D and llA to llB are respectively explanatory diagrams on alignment errors; and Figure 12 is a block diagram showing another example of the invention.
BACKGRO~ OF T~E INVENTION
Field of the Invention:
.
The present invention relates generally to a signal transmission circuit in which a plurality of sampling circuits are interposed, ~nd is directed more particularly to such a signal transmission circuit suitable for use in the signal processing circuit of a solid state television camera in T~7hich a semiconductor device such as a CCD and a BBD (bucket brigade device) is employed.
Descri tion of the Prior Art:
P
By way of example, a prior art solid state television camera using a CCD imager will be now considered. With such a prior art television camera, the output signal from an imaging device is passed through a circuit consisting of a sampling circuit and/or sample-hold circuit connected in cascade to be a final signal. In this case, it is noted that the signal is apt to be deteriorated by the sampling action. This may be considered from the fact that minus components lying above the sampling frequency, which is generated by the first sample-hold action, cancel out effective signal components existing lower than the sampling frequency by the second sampling action, which will be described later.
Fundamentally speaking, the output from a solid state image sensor is a sampled type signal and low in signal level, so that even a little signal deterioration can not be neglected.
In.a charge transfer device such as a CCD, there are many cases where the sample-hold function is requi.red in deriving a signal from the device itself. Accordinglv, the sampling action, which is carried out once after the signal is derived, deteriorates the rrequency response of the solid state imaging syste~ in the $2~
prior art system.
OBJECTS Al~ SUMMARY OF THE INV~NTION
An object of the present invention is to provide a ;
signal transmission circuit having a sampling and/or sample-hold circuit.
Another object of the invention is to provide a novel signal transmission circuit in which circuits with the sampling function are connected in series.
According to the fundamental construction of the inven-tion, a sample-hold circuit, a low pass filter and a sampling circuit (or may be a sample-hold circuit) are connected in series. A sampling pulse train applied to the respective cir-cuits are the same in frequency but may be different in phase and window width.
In the usual manner, the sample-hold circuit shows low pass filter characteristics to a signal to be transmitted therethrough. In this invention, such low pass filter char-acteristics of the sample-hold circuit are able to be cancelled or compensated for by interposing a low pass filter having a cut-off frequency of about the sampling frequency. In this case, the last connected sampling circuit determines the phase of an output signal. Accordingly, by adjusting the window phase and window width of the sampling pulse applied to the last connected sampling circuit, the phase of the output signal from the CCD
imaging device is determined.
Therefore, the signal transmission circuit of the invention can be applied to an image pick-up system which uses a plurality of image sensor chips to improve its resolution.
In such a system it is required that the phase of an output signal is arranged in response to a chip displacement. When . ~
:
the present invention is applied to such a phase arrangement of the output signal, the phase arrangement of the output signal can be performed while signal compensation is achieved.
The additional, and other objects, features and ad-vantages of the present invention will become clear from the following description taken in conjunction with the accompanying drawings through which the like reference numerals and letters indicate the same elements.
BRI~F DESCRIPTION OF THE DRAWINGS
`, Figure 1 is a schematic diagram showing the arrange-ment relation of imaging elements;
Figure 2 is a graph showing the frequency character-istics of a picked-up output;
Figure 3 is a systematic block diagram showing an image sensing apparatus;
Figures 4A to 4G are diagrams used for explaining a sampling operation;
Figures 5A and SB are diagrams used for the explana-tion of the operation of a prior circuit;
Figure 6 is a block diagram showing the fundamental construction of the present invehtion;
Figures 7A, 7B and 7C are diagrams showing the fre-quency characteristics thereof;
Figures 8 and 9 are respectively block diagrams showing solid state television cameras to which the invention is applied;
Figures 10 A to 10D and llA to llB are respectively explanatory diagrams on alignment errors; and Figure 12 is a block diagram showing another example of the invention.
2~L
DESCRIPTION OF THE PREFERRED_E~BODI~ENTS
In a solid state television camera, there is a case where a sample-hold circuit and a circuit having the sampling function (sampling circuit, sample-hold circuit or the like) are connected in cascade in the signal transmission line or path for generating a desired video signal from an output of a solid state imaging element.
For example, such a solid state television camera can be exemplified as an example with the above circuit in which three imaging devices pick up (sample) an object to cancel side band components existing in picked-up outputs so as to suppress bad affects on a reproduced picture to expand a frequency band.
Such an example is disclosed in the U.S. Patent3,975,760.
The feature of this example is that when the same object is picked up, the sampling points by picture elements of three image sensing devices are shifted or displaced. For example, as shown in Figure 1, in view of the spatial positional relation of three image sensing devices lR, lG and lB, they are sequentially displaced in the horizontal direction with respect to the object by, for example, ~ X for the arranging pitch _ of picture cells or elements 2, which are arranged in the horizontal direction, corresponding to a scanning time TH, while in view of the phase relation when the image sensing device lR is taken as a reference, the image sensing device lG such as a CCD chip is given with a phase difference of ~12 from the CCD chip lR and the CCD chip lB is given with a phase difference of al3 from the CCD chip lR, respectively.
The above example is a color television camera, sa that output signals corresponding to the primary colors R(red), G
(green) and B (blue) are derived from the CCD chips lR, lG and lB, ~ 2 respectively.
In the case where the spatial relation between the objective image and the CC~ chips lP~ to lB is selected as above, if the sampling timing of the respective CCD chips lR to lG is selected, or the signals are read out therefrom, satisfying the above relation in view of time, the output level and the positional relation of the sampling pulses become as shown in Figure 2.
Figure 2 shows the composite signal, in which YB re-presents a base band component, YS represents a side band com-ponent, and the phase relation among carriers (sampling pulses) CR, CG and CB of the side band component YS becomes as sho~7n in Figure 2. When the spatial relation completely satisfies the above condition, ~12 = 120 and ~13 = 240, respectively. Therefore, when the object is a black and white image, in the composite out-put the side band component YS is cancelled and hence the com-posite output becomes only the base band component YB.
Figure 3 shows an example of the circuit system which will achieve the above purpose. In this example, the light from an object O is focussed through half mirrors HM and full mirrors FM on the respective imaging devices lR, lG and lB.
In this case, color filters FR, FG and FB are located in front of the imaging devices lR, lG and lB, respectively.
The horizontal shift registers (not shown) provided in the respective imaging devices lR to lB are supplied with a clock pulse PC (with the frequency of, for example l/TH) for sampling (reading out), respectively. Generally speaking, the clock pulses applied to the respective horizontal shift registers has to be selected equal in phase to the spatial phase. But in .
the Figure 3 embodiment, the clock pulses PC with the same phase are applied to the respective horizontal shift registers and in ~6 ~
turn, the matching to the spatial phase is established after the first sampling.
Namely, a picked-up output SR from the imaging device lR is fed through a first sample-hold circuit 5R for wave-shaping to a second sample-hold circuit 7R for phase-matching.
In this case, the clock pulse PC is used as the sampling signal.
Similarly, the other picked-up outputs SG and SB from the other imaging devices lG and lB are wave-shaped by sample-hold circuits 5& and 5B and thereafter adjusted in phase by sample-hold circuits 7G and 7B, respectively. That is, phases ~12 and ~13 of the sampling signals applied to the sample-hold circuits 7G and 7B are adjusted to be coincident with the spatial phases 312 and ~13. In Figure 3, 8G and 8B designate phase-adjusting circuits for the above purpose, respectively. The respective picked-up outputs SR, SG and SB thus phase-adjusted are applied, as well known, to an encoder (which includes a matrix circuit) which then produces a desired color video signal.
It is ascertained that when the pair of sample-hold circuits 5 and 7 are interposed in the transmission systems of the picked-up outputs SR to SB, the frequency characteristics thereof are deteriorated by the sample-hold circuits 5 and 7.
The reason why the frequency characteristics are deteriorated is explained as follows. First, the characteristics of the sample-hold circuits will be now considered. The char-acteristics can be deemed such that, as shown in Figures 4A to 4G, an input signal f(x) (refer to Figure 4A) is sampled by an ideal sampling pulse Pl(x) (refer to Figure 4B) with the pulse width of zero and thereafter a sampled output l(x) (refer to Figure 4C) is held for a constant time period Tl (refer to Figure 4D). This holding effect can be considered as a kind of low pass effects.
, . ~ _ , . ., . . . _ _ . _ ., . ., , ... _, 6C~2 4 The frequency characteristics can be expressed Tl(sin ~ llp/~ Tlp)e j2~PTl/2 when the input signal f(x) is expressed as f(x) = cos 2~px (where ~ represents the repeating frequency). But, when the sample-hold circuits are connected in cascade, there is not obtained such an output that the above operation is repeated twice. Although the theoretical detailed explanation is omitted, the conclusion is as follows. In the output signal, the holding effect of the signal is determined not by the former sample-hold circuit but by the latter sample-hold circuit, and the output signal is delivered under such a state that the output signal is delayed for the input signal f(x) by a phase difference ~ (refer to Figure 4E) between the sampling pulses Pl(x) and P2(x) of both the sample-hold circuits.
Accordingly, even if the sample-hold circuits are connected in cascade over several stages, the hold effect of an output signal S2(x) (refer to Figure 4G) is determined by the sample-hold circuit at the final stage.
If the input signal is taken as f(x) and the ideal sampling pulse train is taken as p(x), respectively, the follow-ing equation (l) is established, P(x) = ~(x-n TH) (1) Thus, the output signal S2(x) is expressed as follows:
S2(x) = [f(x- ~ ) x pl(x- ~ )] * rect ( T~
........ (2) whère rect(x) = 1 ¦xl <
= O lxl > 1 ~ ........ (3) and * represents convolution products.
In the equation (2), the left half term of the right side thereof is a signal 02(x) itself (refer to Figure 4F), which ~ 2 is subjected to the delay caused by the phase difference ~
between the sampling pulses Pl(x) and P2(x), and the remaining right half term of the same side is one representing the holding effect with the width of T2. That is, the final output S2(x) is obtained by applying the holding effect to the sampled output 02(x).
Next, the frequency characteristics of the output signal will be now considered. Now, the frequency characteris-tics of the output signal 02(x) is considered in two ways.
From the equation (2), the following equation (4) is derived.
02(x) = f (X-E) X ~ -n~H~ ----...(4) The frequency component thereof can be expressed as follows by the Fourier transformation.
2(f) = 1 e -j2~fE ~ F(f- n TH n H ................................. (5) While the frequency component of the output signal Sl(x) is expressed by the following equation (6), since it is obtained by holding the output signal l(x) with the holding width (window width) (f) T 1 -j2~f 1 ~F(f- m ........ (6) Since the output signal 02(x) is obtained by sampling signal Sl(x) with the ideal sampling pulse train P2(x), the frequency component 2(f) of the former can be also expressed as follows:
2(f) = Sl(f) '` T (f Q ) e~i2~ TH
T1 1 ~ sin ~T1(~- _ ) -j2~(f- Q )_~_ 'H
e i TH x ~ F(f- T ~ ~ . . (7) m H
Since the equations (5) and (7) must be the same, there-fore, M + Q = n. Thus, from the equations (5) and (7) the follow-ing equation (8) is obtained.
TTl~f ~ ~ e i (f T ) - ~ j2 Q
= e~i2~f~ -------(8) In order to make the effect clear, if = - ~ - - the equation (8) can be rewritten as follows:
T sin ~Tl(f - Q
1 ~ H = 1 TH ~Tl ~f ~ Q ) ........ (9) Now, if the input signal f(x) can be expressed by the following expression (1), f(x) = cos 2~px ........ (10) the equation (9) can be rewritten as follows:
1 ~ sin ~T 1 ( T - P) " ' 1 = 1 H 1( T - P) H ........ (11) Since the equation (11) is established irrespective of the repeating frequency P of the object, an output with a con-stant amplitude can be, of course, delivered.
By the way, the left side of the equation (11) can be expressed as follows:
~ $ 2~
Tl (sin ~rTl p sin rrTl( T - P) SinrrTl(T + Pj) TH l ~rT lP ~rT 1 ( T -P) rrT 1 ( T + p) Ij ........ (12) If it is assumed that Tl _ TH in the equation (12), 2N-1 < Tl( T + p) < 2N ........ (13) When N=l, 2, ... in the equation (13), sin ~rTl ( T +- P) . - - -- < O
~rTl ( T -+ P) ........ (14) since the following conditions (15) and (16) are established, sin ~rTl( TH + P) ~ sin ~rTl( n+l +p ¦ H l l H
sin lrTl( TH ~ P) ~ sin ~rTl( nr+l _ p rrT 1 ( T - P) l l rrTl ( T - P) the left term _ of the equation (11) can be expressed as follows:
T1 ( sin rrTlp sin ~Tl( TH P~ +
H ~ lP rrTl ( 1 - p) Tl (sin ~rTlp sin ~rTl( T - P) ) T ~ ïrTlp ~rTl( 1 - p) ~ s in ~rT 1 ( T + p) S in ~rT 1 ( T P ) TH ~ ~rTl( 1 - + p) 1 T H P
........ (17) ~ 2 The second term of the right side of the equation (17 is a negative component.
Due to the fact that there exists the negative com-ponent expressed by the above equation, the output level L
is forced to be reduced and consequently, the frequency char-acteristics are deteriorated.
Now, the deterioration of frequency characteristics will be described briefly. Since the sample-hold circuit is a kind of modulating circuit, the sampling pulse P2(x) for the sample-hold circuit at the rear stage is modulated by the respective frequency components (p, l/TH - p, l/TH + p, .... ) of the first held output Sl(x) (refer to Figures 5A and 5B).
The modulating signal contains positive and negative components, so that the composite modulated components thereby are affected by the negative modulating signal, then the output level thereof is made constant and the frequency characteristics are deter-mined. Though the frequency characteristics are determined by the next holding effect, the characteristics are such that the high frequency band component is deteriorated, for which improvement is desired.
When the sample-hold circuit at the rear stage is a mere sampling circuit, the holding effect is determined by the first sample-hold circuit and a phenomenon similar to that set forth above is accompanied.
The present invention, in consideration of the fore-going, is to improve the frequency characteristics and in theory to remove the component expressed by the second term of the right side of the equation (17). That is, with the present invention, as shown in Figure 6 which shows the fundamental construction thereof, a filter 13 having the low pass characteristic is interposed between a sample-hold circuit 11 and a circuit 12 having the sampling function which are connected in cascade.
The cut-off frequency of the filter 13 is selected to be 1/TH
in theoretical point of view, but possible somewhat higher or lower than 1ITH in accordance with a purpose of the circuit with no problem. By this, the negative component of the right side of the equation (17) can be removed completely, so that the output level in the case where the filter 13 is interposed is increased as compared with the case where the filter 13 is not used. Thus, the frequency characteristics, especially those lower than the intermediate frequency band can be improved much.
Figure 7 shows the frequency characteristics of the invention. A curve 16 in Figure 7A represents the case where the circuit 12 at the rear stage is a sampling circuit 12S, i.e., the frequency characteristic 2(f) at the time when the sampling is achieved by the ideal sampling pulse, from which it will be apparent that, especially the intermediate frequency band is boosted and hence the effect of the filter 13 is performed sufficiently. When the circuit 12 is made of a sample-hold circuit 12H, its frequency characteristics vary in accordance with the width T2 of the hold pulse. In Figure 7B, a curve 17a represents the frequency characteristic in the case where the width T2 is selected as -~- TH and a curve 17b represents the case where the width T2 is selected as -~- TH, respectively.
Accordingly, the frequency characteristics of the whole circuit in which the sample-hold circuit 12H is used become as shown in Figure 7C. In Figure 7C, a curve 18a repre-sents the frequency characteristic in the case of T2 = 2 TH, a curve 18b represents that in the case of T2 = -~- TH, and the frequency characteristics in the case where the filter 13 is not.
used, are the same as those shown in Figure 7B.
As may be apparent from the comparison of the invention _ _ . .. ...... . ....
~ 2~
with the prior art, when the filter 13 is interposed, the response in the intermediate frequency band and higher are boosted. As a result, the frequency characteristics are im-proved much.
An example of the present invention, which is applied to the signal transmission system of a solid state television camera, will be described.
Figure 8 is a systematic block diagram showing an example of the solid state television camera, in which the fundamental circuit of the invention is applied to each of the transmission paths of the picked-up signals S~, SG and SB, respectively, and in which the elements or parts corresponding to those of Figure 3 are marked with the same reference numerals and letters, respectively.
Now, the transmission path or line for the picked-up signal SR will be described. As shown in Figure 8, a low pass filter 13R and a sampling circuit 12SR are connected in this order to the output terminal of the second sample-hold circuit 7R or at the rear stage thereof to form together with the first sample-hold circuit 5R a transmission circuit or a response compensating circuit lOR in this example. When the filter 13 for improving the frequency characteristics is interposedJ the time delay caused by a former stage sample-hold circuit llR, consisting of the sample-hold circuits 5R and 7R, disappears, so that the phase ~11 in view of time of the sampling pulse fed to the sampling circuit 12SR at the rear stage is compensated for by a phase shifter 9R in the delay caused by the sample-hold circuit 7R and filter 13R from the spatial phase all.
The other signal transmission systems are similarly formed, so that their detailed description will be omitted.
The respective picked-up outputs SR, SG and SB, which are response-compensated for, are applied to a matrix circuit 15 which then produces a luminance signal Y and a pair of color difference signals (R-Y) and (B-Y) (in which R = SR and B = SB).
The luminance signal Y is fed through a low pass filter 16 and a delay circuit 17 to an adding circuit 18, while the pair of color difference signals (R-Y~ and (B-Y) are fed through low pass filters 19A and l9B to an encoder 20 to be converted into a desired carrier chrominance signal. The carrier chrominance signal is fed to the adding circuit 18 which is also supplied with the luminance signal Y and a composite synchronizing signal, so that a desired video signal is provided by the adding circuit 18 and then d~livered to its terminal 18a.
Since the response compensation is the necessary process for the luminance signal system, if the circuit of the invention is applied to the transmission path for the luminance signal Y developed by the matrix circuit 15, the same effect is obtained.
Figure 9 is a block diagram showing an example of this case. In this example, the response compensation is achieved by the first and second sample-hold circuits 5, 7 and the low pass filter 13 and the sampling circuit 12SR which are provided at the rear stage of the matrix circuit 15. In Figure 9, 22 designates a phase shifter and 23 designates a pulse width control circuit~ respectively.
The phase relation among the respective picked-up outputs SR to SB, when they are read out from the imaging devices lR to lB with the spatial phase li of the imaging devices lG and lB being kept, is shown in Figures lOA, lOB and lOC. Accordingly, if the alignment among the imaging devices lR to lB is perfect, the side band components are cancelled by composing the picked-up outputs SR to SB to eliminate influences ,, .................... . ... , ,, , .. ., .... . ., . ._~ .. . .. ... . . . . . .. .
on picture quality when a black and white object is picked up.
However, if any error exists in the alignment, the side band components are not cancelled and remain in a base band component YB with the result that the picture quality can not be improved and also the band width can not be expanded. In a practical problem, the imaging device is small and, if it requires several hundreds of picture elements in the horizontal direction, the mechanical displacement corresponding to the time -~- TH is difficult to be provided. This results in that the imaging device becomes low in yield and hence expensive.
For example, when the spatial phase al2 of the imaging device lB is displaced somewhat from the normal phase (which is shown in Figure lOB by the dotted line), the composite output SO corresponding to the luminance signal Y becomes as shown in Figure lOD. The phase 4of the side band component in Figure lOD shows the phase rotation after the side band component is composed.
The remaining of the side band component due to the alignment error can be easily removed by utilizing the above response compensator while the res~onse is compensated when the error is small. That is, the pulse width T2 and phase of the sampling pulse P2(x) applied to the sampling circuit 12S pro-vided at the rear stage of the filter 13 is sufficient to be selected as desired. The reason will be described with refer-ence to Figure 9.
An output signal Qli(x) from the sample-hold circuits 7R, 7G and 7B will be explained first. When i = 1, Qll(x) = SR.
Hence, Q12(X) = SG~ and Q13(x) = SB~ respectively- If it is assumed that the displacement from the reference phase in space .
is taken as ~ as set forth above, that in time is taken as ~ and the imaging device lR is taken as the reference as shown in Figure 1, the imaging device lG is displaced from the device lR
by 12 in space and ~12 in time. Similarly, the remaining imaging device lB is displaced from the reference device lR by 13 in space and ~l3 in time. Therefore, the output signal Qli(x) is expressed as follows:
Qli(x)=Ki[Ri(p)cos ~2~Px+(9li ~li)TH
+Ri(l/TH-p) cos ¦21r(l/TH-p)X~ (~li- Yli) THP Sli,~
+Ri(l/TH+p) cos ~ 2fr(l/TH+P)X+(~ li) THP Yli~
+Ri(2/TH-p) cos {21r(2/TH+p)x (~ li) T~p li}
+ ..... ] ........ (18) where Ki = - sin n T op TH ~ToP ...... (19) To corresponding to the opening width of the picture element, Li corresponding to the brightness of the object (the delay effect by the holding is neglected) and Ri() represents the total frequency characteris-tics of the transmission path.
In order to compensate for the alignment error in a desired frequency band, the spatial phase li and time phase ~li must be made coincident. Accordingly, the time phase ~12 for the device lB having an alignment error is varied in correspond-ence with a phase 912 including a displacement different from the normal phase and consequently defined as follows:
~ li .................................... ..(20) At this time, the output signal Qli(x) of the equation (18) becomes as follows:
Qli(x) = Ki(p)cos 2~px + Ki(l/lH-p)cos ~ 2~T(llTH-p)x- ~li~
+ Ki(l/lH+P)CS ~ 21r(l/TH+p)X_ ~li l5 + Ki(2/TH-p)COS ~2Tr(2lTH-p)x-2~
+ ................................ ........ (21) Accordingly, if the composite output signal Q(x) of the output signals Qll(x) to Q13(x) (the composite output signal corresponds to the luminance signal Y from the matrix circuit) is passed through the filter 13 having the cut-off frequency of l/TH to eliminate the components higher than l/TH, the com-posite output signal Q(x) can be expressed as follows:
Q(x) = K(p) cos 2~px + K(1/TH_P) [COS ~ 21T(1/TH-P)X ~
+ cos.~2~(1/TH-p)x- ~12 ~+ cos~ 2~(1/TH-p)x- ~l3} ]
........ (22) ..Q(x)=K(p) cos 2~px + K~ (1/TH-P)COS~2~(1/TH-P)X- ~ ~
........ (23) Figure lOD shows the phase relation of the equation (23).
When the output signal Q(x) expressed by the equation (23) is applied to the sampling circuit 12S, it becomes as follows where, however, the pulse width of the sa~pling pulse P2(x) is taken as T2 and its displacement from the sampling pulse applied to the reference imaging device lP~ is taken as Q (time phase is 2~ Q~ .
sin ~
Qo(x) = 12 ~Q(x) ~ 2Q(x) I cos(2~ ~ ~ )3 H
TH [K(p)cos2~px+K (l/TH-p)cos ~2~(1/TH-p)x~
sin ~
+ K (l/lH-p) IH cos (2~px+ ~-2~ T~ ) sln lr + K(p) I cos ~2~(1/TH-p)x-2~ 1 } ]
~r --........ (24) In the equation (24), the components of the first and second terms are the base band component of the final output signal Qo(x) and the components of the third and fourth terms are the side band component.
Accordingly, if the displacement ~ of the sampling pulse from the reference phase is selected by the following equation (25) 2~ ~ = y _ ~
IH ................................... (25) and the base and side band components at the frequency (l/TH-p) are selected by the following equation (26) sin ~ 12 K(p) T H = K'(l/lH-p) IH ................................... (26) the phase of the base band component and the phase of the side band components can be shown in Figures llA and llB, respect-ively. That is, the phases thereof are opposite at the frequency (l/lH-p) and hence in the output signal Qo(X) there is no fre-quency component of (l/TH-p), i.e., side band component. In -rs-__ , ,, _. _ _ . .. , . . .. , ,., _ , . . _ , _ ...
this case, however, the base band component is lowered some-what.
As set forth above, if the equations (20), (25) and (26) are satisfied, the alignment error can be compensated for.
Since during the sampling operation the ratio between the base band component and the side band component can be varied easily by changing the window width of the sampling pulse P2(x), the equation (26) can be satisfied only by select-ing the pulse width ~2. And, if the displacement phase ~ of the sampling pulse P2(x) is controlled in accordance with the phase ~ based upon the alignment error, the equation (25) is satisfied. A practical example, which will compensate for the alignment error, is substantially the same as that shown in Figure 9, in which if only the phase shifter 22 and the pulse width control circuit 23 are controlled in accordance with the alignment error, the desired purpose can be achieved.
Even if a balanced modulator is used as a circuit, which has the sampling function, in place of the sampling cir-cuit in the example of Figure 9, the band compensation and the alignment compensation can be similarly carried out. Figure 12 shows such an example. To achieve this purpose, it is enough to compose the modulated output and modulating output.
In Figure 12, 25 designates an adding or composing circuit, 26 designates a low pass filter, and 27 designates a de ~y circuit, respectively.
Since the modulating output is the equation (14) itself, a modulated output Mo(x) becomes as follows:
Mo(x) = A[K(l/TH-p) cos (2~px-2~ ~ + ~) + K(p) cos ~2~ H-p)x-2~ ~ ~ ]
........ (27) When the equation (25) is satisfied similar to the above and the level is selected equal, the phase relation of the modulating output becomes the same as that shown in Figure llA, and the phase relation of the modulated output becomes the same as that shown in Figure llB. Accordingly, when both the modulating and modulated outputs are composed, the side band component due to the alignment error can be positively removed.
Even with the example of Figure 8, the alignment error can be compensated for by suitably selecting the phase and pulse width of the sampling pulse P2(x). In this case, the circuit 12 which has the sampling function may be a sample-hold circuit.
As described in the foregoing, according to the con-struction of the present invention, the frequency band higher than the intermediate frequency band is boosted and the band com-pensation becomes possible, so that the present invention can be applied to various transmission systems in addition to the above examples.
When the present invention is constructed such as shown in Figures 10 and 12, the alignment error can be compensated for electrically. As a result, the yield of the imaging device can be increased much.
It will be apparent that many modifications and varia-tions could be effected by one skilled in the art without de-parting from the spirit or scope of the novel concepts of the present invention, so that the scope or spirits of the invention should be determined by the appended claims only.
DESCRIPTION OF THE PREFERRED_E~BODI~ENTS
In a solid state television camera, there is a case where a sample-hold circuit and a circuit having the sampling function (sampling circuit, sample-hold circuit or the like) are connected in cascade in the signal transmission line or path for generating a desired video signal from an output of a solid state imaging element.
For example, such a solid state television camera can be exemplified as an example with the above circuit in which three imaging devices pick up (sample) an object to cancel side band components existing in picked-up outputs so as to suppress bad affects on a reproduced picture to expand a frequency band.
Such an example is disclosed in the U.S. Patent3,975,760.
The feature of this example is that when the same object is picked up, the sampling points by picture elements of three image sensing devices are shifted or displaced. For example, as shown in Figure 1, in view of the spatial positional relation of three image sensing devices lR, lG and lB, they are sequentially displaced in the horizontal direction with respect to the object by, for example, ~ X for the arranging pitch _ of picture cells or elements 2, which are arranged in the horizontal direction, corresponding to a scanning time TH, while in view of the phase relation when the image sensing device lR is taken as a reference, the image sensing device lG such as a CCD chip is given with a phase difference of ~12 from the CCD chip lR and the CCD chip lB is given with a phase difference of al3 from the CCD chip lR, respectively.
The above example is a color television camera, sa that output signals corresponding to the primary colors R(red), G
(green) and B (blue) are derived from the CCD chips lR, lG and lB, ~ 2 respectively.
In the case where the spatial relation between the objective image and the CC~ chips lP~ to lB is selected as above, if the sampling timing of the respective CCD chips lR to lG is selected, or the signals are read out therefrom, satisfying the above relation in view of time, the output level and the positional relation of the sampling pulses become as shown in Figure 2.
Figure 2 shows the composite signal, in which YB re-presents a base band component, YS represents a side band com-ponent, and the phase relation among carriers (sampling pulses) CR, CG and CB of the side band component YS becomes as sho~7n in Figure 2. When the spatial relation completely satisfies the above condition, ~12 = 120 and ~13 = 240, respectively. Therefore, when the object is a black and white image, in the composite out-put the side band component YS is cancelled and hence the com-posite output becomes only the base band component YB.
Figure 3 shows an example of the circuit system which will achieve the above purpose. In this example, the light from an object O is focussed through half mirrors HM and full mirrors FM on the respective imaging devices lR, lG and lB.
In this case, color filters FR, FG and FB are located in front of the imaging devices lR, lG and lB, respectively.
The horizontal shift registers (not shown) provided in the respective imaging devices lR to lB are supplied with a clock pulse PC (with the frequency of, for example l/TH) for sampling (reading out), respectively. Generally speaking, the clock pulses applied to the respective horizontal shift registers has to be selected equal in phase to the spatial phase. But in .
the Figure 3 embodiment, the clock pulses PC with the same phase are applied to the respective horizontal shift registers and in ~6 ~
turn, the matching to the spatial phase is established after the first sampling.
Namely, a picked-up output SR from the imaging device lR is fed through a first sample-hold circuit 5R for wave-shaping to a second sample-hold circuit 7R for phase-matching.
In this case, the clock pulse PC is used as the sampling signal.
Similarly, the other picked-up outputs SG and SB from the other imaging devices lG and lB are wave-shaped by sample-hold circuits 5& and 5B and thereafter adjusted in phase by sample-hold circuits 7G and 7B, respectively. That is, phases ~12 and ~13 of the sampling signals applied to the sample-hold circuits 7G and 7B are adjusted to be coincident with the spatial phases 312 and ~13. In Figure 3, 8G and 8B designate phase-adjusting circuits for the above purpose, respectively. The respective picked-up outputs SR, SG and SB thus phase-adjusted are applied, as well known, to an encoder (which includes a matrix circuit) which then produces a desired color video signal.
It is ascertained that when the pair of sample-hold circuits 5 and 7 are interposed in the transmission systems of the picked-up outputs SR to SB, the frequency characteristics thereof are deteriorated by the sample-hold circuits 5 and 7.
The reason why the frequency characteristics are deteriorated is explained as follows. First, the characteristics of the sample-hold circuits will be now considered. The char-acteristics can be deemed such that, as shown in Figures 4A to 4G, an input signal f(x) (refer to Figure 4A) is sampled by an ideal sampling pulse Pl(x) (refer to Figure 4B) with the pulse width of zero and thereafter a sampled output l(x) (refer to Figure 4C) is held for a constant time period Tl (refer to Figure 4D). This holding effect can be considered as a kind of low pass effects.
, . ~ _ , . ., . . . _ _ . _ ., . ., , ... _, 6C~2 4 The frequency characteristics can be expressed Tl(sin ~ llp/~ Tlp)e j2~PTl/2 when the input signal f(x) is expressed as f(x) = cos 2~px (where ~ represents the repeating frequency). But, when the sample-hold circuits are connected in cascade, there is not obtained such an output that the above operation is repeated twice. Although the theoretical detailed explanation is omitted, the conclusion is as follows. In the output signal, the holding effect of the signal is determined not by the former sample-hold circuit but by the latter sample-hold circuit, and the output signal is delivered under such a state that the output signal is delayed for the input signal f(x) by a phase difference ~ (refer to Figure 4E) between the sampling pulses Pl(x) and P2(x) of both the sample-hold circuits.
Accordingly, even if the sample-hold circuits are connected in cascade over several stages, the hold effect of an output signal S2(x) (refer to Figure 4G) is determined by the sample-hold circuit at the final stage.
If the input signal is taken as f(x) and the ideal sampling pulse train is taken as p(x), respectively, the follow-ing equation (l) is established, P(x) = ~(x-n TH) (1) Thus, the output signal S2(x) is expressed as follows:
S2(x) = [f(x- ~ ) x pl(x- ~ )] * rect ( T~
........ (2) whère rect(x) = 1 ¦xl <
= O lxl > 1 ~ ........ (3) and * represents convolution products.
In the equation (2), the left half term of the right side thereof is a signal 02(x) itself (refer to Figure 4F), which ~ 2 is subjected to the delay caused by the phase difference ~
between the sampling pulses Pl(x) and P2(x), and the remaining right half term of the same side is one representing the holding effect with the width of T2. That is, the final output S2(x) is obtained by applying the holding effect to the sampled output 02(x).
Next, the frequency characteristics of the output signal will be now considered. Now, the frequency characteris-tics of the output signal 02(x) is considered in two ways.
From the equation (2), the following equation (4) is derived.
02(x) = f (X-E) X ~ -n~H~ ----...(4) The frequency component thereof can be expressed as follows by the Fourier transformation.
2(f) = 1 e -j2~fE ~ F(f- n TH n H ................................. (5) While the frequency component of the output signal Sl(x) is expressed by the following equation (6), since it is obtained by holding the output signal l(x) with the holding width (window width) (f) T 1 -j2~f 1 ~F(f- m ........ (6) Since the output signal 02(x) is obtained by sampling signal Sl(x) with the ideal sampling pulse train P2(x), the frequency component 2(f) of the former can be also expressed as follows:
2(f) = Sl(f) '` T (f Q ) e~i2~ TH
T1 1 ~ sin ~T1(~- _ ) -j2~(f- Q )_~_ 'H
e i TH x ~ F(f- T ~ ~ . . (7) m H
Since the equations (5) and (7) must be the same, there-fore, M + Q = n. Thus, from the equations (5) and (7) the follow-ing equation (8) is obtained.
TTl~f ~ ~ e i (f T ) - ~ j2 Q
= e~i2~f~ -------(8) In order to make the effect clear, if = - ~ - - the equation (8) can be rewritten as follows:
T sin ~Tl(f - Q
1 ~ H = 1 TH ~Tl ~f ~ Q ) ........ (9) Now, if the input signal f(x) can be expressed by the following expression (1), f(x) = cos 2~px ........ (10) the equation (9) can be rewritten as follows:
1 ~ sin ~T 1 ( T - P) " ' 1 = 1 H 1( T - P) H ........ (11) Since the equation (11) is established irrespective of the repeating frequency P of the object, an output with a con-stant amplitude can be, of course, delivered.
By the way, the left side of the equation (11) can be expressed as follows:
~ $ 2~
Tl (sin ~rTl p sin rrTl( T - P) SinrrTl(T + Pj) TH l ~rT lP ~rT 1 ( T -P) rrT 1 ( T + p) Ij ........ (12) If it is assumed that Tl _ TH in the equation (12), 2N-1 < Tl( T + p) < 2N ........ (13) When N=l, 2, ... in the equation (13), sin ~rTl ( T +- P) . - - -- < O
~rTl ( T -+ P) ........ (14) since the following conditions (15) and (16) are established, sin ~rTl( TH + P) ~ sin ~rTl( n+l +p ¦ H l l H
sin lrTl( TH ~ P) ~ sin ~rTl( nr+l _ p rrT 1 ( T - P) l l rrTl ( T - P) the left term _ of the equation (11) can be expressed as follows:
T1 ( sin rrTlp sin ~Tl( TH P~ +
H ~ lP rrTl ( 1 - p) Tl (sin ~rTlp sin ~rTl( T - P) ) T ~ ïrTlp ~rTl( 1 - p) ~ s in ~rT 1 ( T + p) S in ~rT 1 ( T P ) TH ~ ~rTl( 1 - + p) 1 T H P
........ (17) ~ 2 The second term of the right side of the equation (17 is a negative component.
Due to the fact that there exists the negative com-ponent expressed by the above equation, the output level L
is forced to be reduced and consequently, the frequency char-acteristics are deteriorated.
Now, the deterioration of frequency characteristics will be described briefly. Since the sample-hold circuit is a kind of modulating circuit, the sampling pulse P2(x) for the sample-hold circuit at the rear stage is modulated by the respective frequency components (p, l/TH - p, l/TH + p, .... ) of the first held output Sl(x) (refer to Figures 5A and 5B).
The modulating signal contains positive and negative components, so that the composite modulated components thereby are affected by the negative modulating signal, then the output level thereof is made constant and the frequency characteristics are deter-mined. Though the frequency characteristics are determined by the next holding effect, the characteristics are such that the high frequency band component is deteriorated, for which improvement is desired.
When the sample-hold circuit at the rear stage is a mere sampling circuit, the holding effect is determined by the first sample-hold circuit and a phenomenon similar to that set forth above is accompanied.
The present invention, in consideration of the fore-going, is to improve the frequency characteristics and in theory to remove the component expressed by the second term of the right side of the equation (17). That is, with the present invention, as shown in Figure 6 which shows the fundamental construction thereof, a filter 13 having the low pass characteristic is interposed between a sample-hold circuit 11 and a circuit 12 having the sampling function which are connected in cascade.
The cut-off frequency of the filter 13 is selected to be 1/TH
in theoretical point of view, but possible somewhat higher or lower than 1ITH in accordance with a purpose of the circuit with no problem. By this, the negative component of the right side of the equation (17) can be removed completely, so that the output level in the case where the filter 13 is interposed is increased as compared with the case where the filter 13 is not used. Thus, the frequency characteristics, especially those lower than the intermediate frequency band can be improved much.
Figure 7 shows the frequency characteristics of the invention. A curve 16 in Figure 7A represents the case where the circuit 12 at the rear stage is a sampling circuit 12S, i.e., the frequency characteristic 2(f) at the time when the sampling is achieved by the ideal sampling pulse, from which it will be apparent that, especially the intermediate frequency band is boosted and hence the effect of the filter 13 is performed sufficiently. When the circuit 12 is made of a sample-hold circuit 12H, its frequency characteristics vary in accordance with the width T2 of the hold pulse. In Figure 7B, a curve 17a represents the frequency characteristic in the case where the width T2 is selected as -~- TH and a curve 17b represents the case where the width T2 is selected as -~- TH, respectively.
Accordingly, the frequency characteristics of the whole circuit in which the sample-hold circuit 12H is used become as shown in Figure 7C. In Figure 7C, a curve 18a repre-sents the frequency characteristic in the case of T2 = 2 TH, a curve 18b represents that in the case of T2 = -~- TH, and the frequency characteristics in the case where the filter 13 is not.
used, are the same as those shown in Figure 7B.
As may be apparent from the comparison of the invention _ _ . .. ...... . ....
~ 2~
with the prior art, when the filter 13 is interposed, the response in the intermediate frequency band and higher are boosted. As a result, the frequency characteristics are im-proved much.
An example of the present invention, which is applied to the signal transmission system of a solid state television camera, will be described.
Figure 8 is a systematic block diagram showing an example of the solid state television camera, in which the fundamental circuit of the invention is applied to each of the transmission paths of the picked-up signals S~, SG and SB, respectively, and in which the elements or parts corresponding to those of Figure 3 are marked with the same reference numerals and letters, respectively.
Now, the transmission path or line for the picked-up signal SR will be described. As shown in Figure 8, a low pass filter 13R and a sampling circuit 12SR are connected in this order to the output terminal of the second sample-hold circuit 7R or at the rear stage thereof to form together with the first sample-hold circuit 5R a transmission circuit or a response compensating circuit lOR in this example. When the filter 13 for improving the frequency characteristics is interposedJ the time delay caused by a former stage sample-hold circuit llR, consisting of the sample-hold circuits 5R and 7R, disappears, so that the phase ~11 in view of time of the sampling pulse fed to the sampling circuit 12SR at the rear stage is compensated for by a phase shifter 9R in the delay caused by the sample-hold circuit 7R and filter 13R from the spatial phase all.
The other signal transmission systems are similarly formed, so that their detailed description will be omitted.
The respective picked-up outputs SR, SG and SB, which are response-compensated for, are applied to a matrix circuit 15 which then produces a luminance signal Y and a pair of color difference signals (R-Y) and (B-Y) (in which R = SR and B = SB).
The luminance signal Y is fed through a low pass filter 16 and a delay circuit 17 to an adding circuit 18, while the pair of color difference signals (R-Y~ and (B-Y) are fed through low pass filters 19A and l9B to an encoder 20 to be converted into a desired carrier chrominance signal. The carrier chrominance signal is fed to the adding circuit 18 which is also supplied with the luminance signal Y and a composite synchronizing signal, so that a desired video signal is provided by the adding circuit 18 and then d~livered to its terminal 18a.
Since the response compensation is the necessary process for the luminance signal system, if the circuit of the invention is applied to the transmission path for the luminance signal Y developed by the matrix circuit 15, the same effect is obtained.
Figure 9 is a block diagram showing an example of this case. In this example, the response compensation is achieved by the first and second sample-hold circuits 5, 7 and the low pass filter 13 and the sampling circuit 12SR which are provided at the rear stage of the matrix circuit 15. In Figure 9, 22 designates a phase shifter and 23 designates a pulse width control circuit~ respectively.
The phase relation among the respective picked-up outputs SR to SB, when they are read out from the imaging devices lR to lB with the spatial phase li of the imaging devices lG and lB being kept, is shown in Figures lOA, lOB and lOC. Accordingly, if the alignment among the imaging devices lR to lB is perfect, the side band components are cancelled by composing the picked-up outputs SR to SB to eliminate influences ,, .................... . ... , ,, , .. ., .... . ., . ._~ .. . .. ... . . . . . .. .
on picture quality when a black and white object is picked up.
However, if any error exists in the alignment, the side band components are not cancelled and remain in a base band component YB with the result that the picture quality can not be improved and also the band width can not be expanded. In a practical problem, the imaging device is small and, if it requires several hundreds of picture elements in the horizontal direction, the mechanical displacement corresponding to the time -~- TH is difficult to be provided. This results in that the imaging device becomes low in yield and hence expensive.
For example, when the spatial phase al2 of the imaging device lB is displaced somewhat from the normal phase (which is shown in Figure lOB by the dotted line), the composite output SO corresponding to the luminance signal Y becomes as shown in Figure lOD. The phase 4of the side band component in Figure lOD shows the phase rotation after the side band component is composed.
The remaining of the side band component due to the alignment error can be easily removed by utilizing the above response compensator while the res~onse is compensated when the error is small. That is, the pulse width T2 and phase of the sampling pulse P2(x) applied to the sampling circuit 12S pro-vided at the rear stage of the filter 13 is sufficient to be selected as desired. The reason will be described with refer-ence to Figure 9.
An output signal Qli(x) from the sample-hold circuits 7R, 7G and 7B will be explained first. When i = 1, Qll(x) = SR.
Hence, Q12(X) = SG~ and Q13(x) = SB~ respectively- If it is assumed that the displacement from the reference phase in space .
is taken as ~ as set forth above, that in time is taken as ~ and the imaging device lR is taken as the reference as shown in Figure 1, the imaging device lG is displaced from the device lR
by 12 in space and ~12 in time. Similarly, the remaining imaging device lB is displaced from the reference device lR by 13 in space and ~l3 in time. Therefore, the output signal Qli(x) is expressed as follows:
Qli(x)=Ki[Ri(p)cos ~2~Px+(9li ~li)TH
+Ri(l/TH-p) cos ¦21r(l/TH-p)X~ (~li- Yli) THP Sli,~
+Ri(l/TH+p) cos ~ 2fr(l/TH+P)X+(~ li) THP Yli~
+Ri(2/TH-p) cos {21r(2/TH+p)x (~ li) T~p li}
+ ..... ] ........ (18) where Ki = - sin n T op TH ~ToP ...... (19) To corresponding to the opening width of the picture element, Li corresponding to the brightness of the object (the delay effect by the holding is neglected) and Ri() represents the total frequency characteris-tics of the transmission path.
In order to compensate for the alignment error in a desired frequency band, the spatial phase li and time phase ~li must be made coincident. Accordingly, the time phase ~12 for the device lB having an alignment error is varied in correspond-ence with a phase 912 including a displacement different from the normal phase and consequently defined as follows:
~ li .................................... ..(20) At this time, the output signal Qli(x) of the equation (18) becomes as follows:
Qli(x) = Ki(p)cos 2~px + Ki(l/lH-p)cos ~ 2~T(llTH-p)x- ~li~
+ Ki(l/lH+P)CS ~ 21r(l/TH+p)X_ ~li l5 + Ki(2/TH-p)COS ~2Tr(2lTH-p)x-2~
+ ................................ ........ (21) Accordingly, if the composite output signal Q(x) of the output signals Qll(x) to Q13(x) (the composite output signal corresponds to the luminance signal Y from the matrix circuit) is passed through the filter 13 having the cut-off frequency of l/TH to eliminate the components higher than l/TH, the com-posite output signal Q(x) can be expressed as follows:
Q(x) = K(p) cos 2~px + K(1/TH_P) [COS ~ 21T(1/TH-P)X ~
+ cos.~2~(1/TH-p)x- ~12 ~+ cos~ 2~(1/TH-p)x- ~l3} ]
........ (22) ..Q(x)=K(p) cos 2~px + K~ (1/TH-P)COS~2~(1/TH-P)X- ~ ~
........ (23) Figure lOD shows the phase relation of the equation (23).
When the output signal Q(x) expressed by the equation (23) is applied to the sampling circuit 12S, it becomes as follows where, however, the pulse width of the sa~pling pulse P2(x) is taken as T2 and its displacement from the sampling pulse applied to the reference imaging device lP~ is taken as Q (time phase is 2~ Q~ .
sin ~
Qo(x) = 12 ~Q(x) ~ 2Q(x) I cos(2~ ~ ~ )3 H
TH [K(p)cos2~px+K (l/TH-p)cos ~2~(1/TH-p)x~
sin ~
+ K (l/lH-p) IH cos (2~px+ ~-2~ T~ ) sln lr + K(p) I cos ~2~(1/TH-p)x-2~ 1 } ]
~r --........ (24) In the equation (24), the components of the first and second terms are the base band component of the final output signal Qo(x) and the components of the third and fourth terms are the side band component.
Accordingly, if the displacement ~ of the sampling pulse from the reference phase is selected by the following equation (25) 2~ ~ = y _ ~
IH ................................... (25) and the base and side band components at the frequency (l/TH-p) are selected by the following equation (26) sin ~ 12 K(p) T H = K'(l/lH-p) IH ................................... (26) the phase of the base band component and the phase of the side band components can be shown in Figures llA and llB, respect-ively. That is, the phases thereof are opposite at the frequency (l/lH-p) and hence in the output signal Qo(X) there is no fre-quency component of (l/TH-p), i.e., side band component. In -rs-__ , ,, _. _ _ . .. , . . .. , ,., _ , . . _ , _ ...
this case, however, the base band component is lowered some-what.
As set forth above, if the equations (20), (25) and (26) are satisfied, the alignment error can be compensated for.
Since during the sampling operation the ratio between the base band component and the side band component can be varied easily by changing the window width of the sampling pulse P2(x), the equation (26) can be satisfied only by select-ing the pulse width ~2. And, if the displacement phase ~ of the sampling pulse P2(x) is controlled in accordance with the phase ~ based upon the alignment error, the equation (25) is satisfied. A practical example, which will compensate for the alignment error, is substantially the same as that shown in Figure 9, in which if only the phase shifter 22 and the pulse width control circuit 23 are controlled in accordance with the alignment error, the desired purpose can be achieved.
Even if a balanced modulator is used as a circuit, which has the sampling function, in place of the sampling cir-cuit in the example of Figure 9, the band compensation and the alignment compensation can be similarly carried out. Figure 12 shows such an example. To achieve this purpose, it is enough to compose the modulated output and modulating output.
In Figure 12, 25 designates an adding or composing circuit, 26 designates a low pass filter, and 27 designates a de ~y circuit, respectively.
Since the modulating output is the equation (14) itself, a modulated output Mo(x) becomes as follows:
Mo(x) = A[K(l/TH-p) cos (2~px-2~ ~ + ~) + K(p) cos ~2~ H-p)x-2~ ~ ~ ]
........ (27) When the equation (25) is satisfied similar to the above and the level is selected equal, the phase relation of the modulating output becomes the same as that shown in Figure llA, and the phase relation of the modulated output becomes the same as that shown in Figure llB. Accordingly, when both the modulating and modulated outputs are composed, the side band component due to the alignment error can be positively removed.
Even with the example of Figure 8, the alignment error can be compensated for by suitably selecting the phase and pulse width of the sampling pulse P2(x). In this case, the circuit 12 which has the sampling function may be a sample-hold circuit.
As described in the foregoing, according to the con-struction of the present invention, the frequency band higher than the intermediate frequency band is boosted and the band com-pensation becomes possible, so that the present invention can be applied to various transmission systems in addition to the above examples.
When the present invention is constructed such as shown in Figures 10 and 12, the alignment error can be compensated for electrically. As a result, the yield of the imaging device can be increased much.
It will be apparent that many modifications and varia-tions could be effected by one skilled in the art without de-parting from the spirit or scope of the novel concepts of the present invention, so that the scope or spirits of the invention should be determined by the appended claims only.
Claims (12)
1. A signal transmission circuit comprising:
a) a first sample-hold circuit for receiving an incoming signal and supplying sample-hold output signal;
b) a first sampling pulse supplied to said first sample-hold circuit;
c) a low pass filter circuit to be supplied with said sample-hold output signal from said first sample-hold circuit;
d) a second sampling circuit connected to said low pass filter for re-sampling an output from said low pass filter circuit; and e) a second sampling pulse supplied to said second sampling circuit.
a) a first sample-hold circuit for receiving an incoming signal and supplying sample-hold output signal;
b) a first sampling pulse supplied to said first sample-hold circuit;
c) a low pass filter circuit to be supplied with said sample-hold output signal from said first sample-hold circuit;
d) a second sampling circuit connected to said low pass filter for re-sampling an output from said low pass filter circuit; and e) a second sampling pulse supplied to said second sampling circuit.
2. A signal transmission circuit as claimed in claim 1, wherein said second sampling circuit in-cludes a signal-hold circuit.
3. A signal transmission circuit as claimed in claim 1, wherein a frequency f1 of said first sampling pulse is equal to a frequency f2 of said second sampling pulse, and a cut-off frequency fcut of said low pass filter is selected to be about said frequency.
4. A signal transmission circuit as claimed in claim 3, wherein said second sampling circuit includes a signal-hold circuit, and the phase of the second sampling pulse is adjustable relative to said first sampling pulse.
5. A signal transmission circuit as claimed in claim 4, wherein said second sampling pulse is generated by passing said first sampling pulse through a vari-able phase shift circuit.
6. A signal transmission circuit as claimed in claim 4, wherein a holding time at said signal-hold circuit in the second sampling circuit is made to be adjustable.
7. A signal transmission circuit as claimed in claim 1, wherein said second sampling circuit is comprised of a sampling gate circuit.
8. A signal transmission circuit as claimed in claim 1, wherein said second sampling circuit is comprised of a balanced modulator and said second sampling pulse is supplied to the balanced modulator as a carrier signal.
9. A signal transmission circuit as claimed in claim 1, where said signal transmission circuit is applied to a signal derived from a sampling-type signal transmitting device.
10. A signal transmission circuit as claimed in claim 4, where said signal transmission circuit is applied to a solid state image pick-up system having a plurality of image sensing chips.
11. A signal transmission circuit as claimed in claim 10, wherein said plurality of image sensing chips are displaced each other relative to a focused object image, and said signal transmission circuit is interposed in respective outputs of said sensor chips.
12. A signal transmission circuit as claimed in claim 11, wherein said respective outputs after said signal transmission circuit are mixed for cancelling side band components for improving the resolution of a reproduced image.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP12891377A JPS5461822A (en) | 1977-10-27 | 1977-10-27 | Transmitter circuit |
| JP128913/77 | 1977-10-27 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1106924A true CA1106924A (en) | 1981-08-11 |
Family
ID=14996455
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA314,249A Expired CA1106924A (en) | 1977-10-27 | 1978-10-25 | Signal transmission circuit |
Country Status (7)
| Country | Link |
|---|---|
| US (1) | US4220977A (en) |
| JP (1) | JPS5461822A (en) |
| CA (1) | CA1106924A (en) |
| DE (1) | DE2846869A1 (en) |
| FR (1) | FR2407628B1 (en) |
| GB (1) | GB2007057B (en) |
| NL (1) | NL7810701A (en) |
Families Citing this family (46)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPS5819798A (en) * | 1981-07-28 | 1983-02-04 | Sony Corp | Sample holding circuit |
| US4524390A (en) * | 1983-03-07 | 1985-06-18 | Eastman Kodak Company | Imaging apparatus |
| JPS59174074A (en) * | 1983-03-23 | 1984-10-02 | Toshiba Corp | Reproducing circuit for output signal of solid state image pickup device |
| DE3418787A1 (en) * | 1984-05-19 | 1985-11-21 | Robert Bosch Gmbh, 7000 Stuttgart | METHOD FOR INCREASING THE RESOLUTION OF COLOR TELEVISION CAMERAS |
| FR2606242B1 (en) * | 1986-10-31 | 1989-02-24 | Thomson Csf | VIDEO MIXER CIRCUIT, USEFUL FOR HIGH RESOLUTION IMAGING WITH CAMERAS WITH SOLID MATRIX SENSORS |
| JP2919110B2 (en) * | 1990-08-28 | 1999-07-12 | 池上通信機株式会社 | Output signal processing circuit of solid-state imaging device |
| KR940004433B1 (en) * | 1991-02-26 | 1994-05-25 | 삼성전자 주식회사 | Method and apparatus for moving picture element by using sample and holding method |
| US7515896B1 (en) | 1998-10-21 | 2009-04-07 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same, and aperture relationships |
| US6061551A (en) | 1998-10-21 | 2000-05-09 | Parkervision, Inc. | Method and system for down-converting electromagnetic signals |
| US6694128B1 (en) * | 1998-08-18 | 2004-02-17 | Parkervision, Inc. | Frequency synthesizer using universal frequency translation technology |
| US6061555A (en) | 1998-10-21 | 2000-05-09 | Parkervision, Inc. | Method and system for ensuring reception of a communications signal |
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| US7236754B2 (en) | 1999-08-23 | 2007-06-26 | Parkervision, Inc. | Method and system for frequency up-conversion |
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| US6879817B1 (en) | 1999-04-16 | 2005-04-12 | Parkervision, Inc. | DC offset, re-radiation, and I/Q solutions using universal frequency translation technology |
| US6853690B1 (en) | 1999-04-16 | 2005-02-08 | Parkervision, Inc. | Method, system and apparatus for balanced frequency up-conversion of a baseband signal and 4-phase receiver and transceiver embodiments |
| US6873836B1 (en) | 1999-03-03 | 2005-03-29 | Parkervision, Inc. | Universal platform module and methods and apparatuses relating thereto enabled by universal frequency translation technology |
| US7110435B1 (en) | 1999-03-15 | 2006-09-19 | Parkervision, Inc. | Spread spectrum applications of universal frequency translation |
| US7693230B2 (en) | 1999-04-16 | 2010-04-06 | Parkervision, Inc. | Apparatus and method of differential IQ frequency up-conversion |
| US7110444B1 (en) | 1999-08-04 | 2006-09-19 | Parkervision, Inc. | Wireless local area network (WLAN) using universal frequency translation technology including multi-phase embodiments and circuit implementations |
| US7065162B1 (en) | 1999-04-16 | 2006-06-20 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same |
| US7054296B1 (en) | 1999-08-04 | 2006-05-30 | Parkervision, Inc. | Wireless local area network (WLAN) technology and applications including techniques of universal frequency translation |
| US8295406B1 (en) | 1999-08-04 | 2012-10-23 | Parkervision, Inc. | Universal platform module for a plurality of communication protocols |
| US7082171B1 (en) | 1999-11-24 | 2006-07-25 | Parkervision, Inc. | Phase shifting applications of universal frequency translation |
| US6963734B2 (en) | 1999-12-22 | 2005-11-08 | Parkervision, Inc. | Differential frequency down-conversion using techniques of universal frequency translation technology |
| US7292835B2 (en) | 2000-01-28 | 2007-11-06 | Parkervision, Inc. | Wireless and wired cable modem applications of universal frequency translation technology |
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| US7554508B2 (en) | 2000-06-09 | 2009-06-30 | Parker Vision, Inc. | Phased array antenna applications on universal frequency translation |
| US7454453B2 (en) | 2000-11-14 | 2008-11-18 | Parkervision, Inc. | Methods, systems, and computer program products for parallel correlation and applications thereof |
| US7010559B2 (en) | 2000-11-14 | 2006-03-07 | Parkervision, Inc. | Method and apparatus for a parallel correlator and applications thereof |
| US7085335B2 (en) | 2001-11-09 | 2006-08-01 | Parkervision, Inc. | Method and apparatus for reducing DC offsets in a communication system |
| US7072427B2 (en) | 2001-11-09 | 2006-07-04 | Parkervision, Inc. | Method and apparatus for reducing DC offsets in a communication system |
| JP4166466B2 (en) * | 2001-12-19 | 2008-10-15 | ソニー株式会社 | Wireless communication system and wireless communication method, wireless communication apparatus and control method thereof, and computer program |
| US6975848B2 (en) | 2002-06-04 | 2005-12-13 | Parkervision, Inc. | Method and apparatus for DC offset removal in a radio frequency communication channel |
| US7321640B2 (en) | 2002-06-07 | 2008-01-22 | Parkervision, Inc. | Active polyphase inverter filter for quadrature signal generation |
| US7379883B2 (en) | 2002-07-18 | 2008-05-27 | Parkervision, Inc. | Networking methods and systems |
| US7460584B2 (en) | 2002-07-18 | 2008-12-02 | Parkervision, Inc. | Networking methods and systems |
| JP5452973B2 (en) * | 2009-04-28 | 2014-03-26 | 富士機械製造株式会社 | IMAGING DEVICE AND CUTTING MACHINE HAVING THE IMAGING DEVICE |
Family Cites Families (6)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB928730A (en) * | 1960-12-22 | 1963-06-12 | British Broadcasting Corpor At | Improvements in and relating to the conversion of television signals from one standard to another |
| DE1774121A1 (en) * | 1968-04-11 | 1971-06-24 | Telefunken Patent | Summing sample and hold element (delay element) |
| US4071853A (en) * | 1974-03-29 | 1978-01-31 | Sony Corporation | Solid state television camera |
| JPS51132719A (en) * | 1975-05-13 | 1976-11-18 | Sony Corp | Solid-image pickup device |
| US4038690A (en) * | 1976-01-21 | 1977-07-26 | Fairchild Camera And Instrument Corporation | Charge-coupled device video-signal-generating system |
| US4153915A (en) * | 1978-03-06 | 1979-05-08 | Us Navy | Optical image modifier for distortion or other compensation |
-
1977
- 1977-10-27 JP JP12891377A patent/JPS5461822A/en active Granted
-
1978
- 1978-10-24 US US05/954,099 patent/US4220977A/en not_active Expired - Lifetime
- 1978-10-25 CA CA314,249A patent/CA1106924A/en not_active Expired
- 1978-10-26 NL NL7810701A patent/NL7810701A/en not_active Application Discontinuation
- 1978-10-27 GB GB7842184A patent/GB2007057B/en not_active Expired
- 1978-10-27 FR FR7830681A patent/FR2407628B1/en not_active Expired
- 1978-10-27 DE DE19782846869 patent/DE2846869A1/en not_active Ceased
Also Published As
| Publication number | Publication date |
|---|---|
| FR2407628B1 (en) | 1985-11-29 |
| JPS6145433B2 (en) | 1986-10-08 |
| US4220977A (en) | 1980-09-02 |
| GB2007057B (en) | 1982-04-07 |
| FR2407628A1 (en) | 1979-05-25 |
| GB2007057A (en) | 1979-05-10 |
| JPS5461822A (en) | 1979-05-18 |
| DE2846869A1 (en) | 1979-05-10 |
| NL7810701A (en) | 1979-05-02 |
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