CA1188758A - Electronic arrangement for generating an amplitude and phase-modulated carrier signal - Google Patents
Electronic arrangement for generating an amplitude and phase-modulated carrier signalInfo
- Publication number
- CA1188758A CA1188758A CA000397592A CA397592A CA1188758A CA 1188758 A CA1188758 A CA 1188758A CA 000397592 A CA000397592 A CA 000397592A CA 397592 A CA397592 A CA 397592A CA 1188758 A CA1188758 A CA 1188758A
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- CA
- Canada
- Prior art keywords
- signal
- phase
- signals
- arrangement
- output
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
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Classifications
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03C—MODULATION
- H03C5/00—Amplitude modulation and angle modulation produced simultaneously or at will by the same modulating signal
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/68—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission for wholly or partially suppressing the carrier or one side band
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/18—Phase-modulated carrier systems, i.e. using phase-shift keying
- H04L27/20—Modulator circuits; Transmitter circuits
- H04L27/2032—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner
- H04L27/2053—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases
- H04L27/206—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers
- H04L27/2067—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states
- H04L27/2078—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states in which the phase change per symbol period is constrained
- H04L27/2082—Modulator circuits; Transmitter circuits for discrete phase modulation, e.g. in which the phase of the carrier is modulated in a nominally instantaneous manner using more than one carrier, e.g. carriers with different phases using a pair of orthogonal carriers, e.g. quadrature carriers with more than two phase states in which the phase change per symbol period is constrained for offset or staggered quadrature phase shift keying
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
- H04L27/00—Modulated-carrier systems
- H04L27/32—Carrier systems characterised by combinations of two or more of the types covered by groups H04L27/02, H04L27/10, H04L27/18 or H04L27/26
- H04L27/34—Amplitude- and phase-modulated carrier systems, e.g. quadrature-amplitude modulated carrier systems
- H04L27/36—Modulator circuits; Transmitter circuits
- H04L27/362—Modulation using more than one carrier, e.g. with quadrature carriers, separately amplitude modulated
Abstract
ABSTRACT:
When an amplitude and phase-modulated carrier signal is generated by an electronic arrangement, two phase-modulated auxiliary signals are first generated from two modulation signals with substantially the same amplitude in order to avoid non-linear distortion. There-after the amplitude and phase-modulated carrier signal is generated by summing the phase-modulated signals by addi-tion or subtraction. The phase of the auxiliary signals are chosen so that the amplitude modulation of the ampli-tude and phase-modulated signal depends on the difference between these phases and depends on the modules of the modulation signals represented in a system of polar co-ordinates, the phase modulation of the amplitude and phase-modulated signal depending on the sum of the phases and on that argument of the phase-modulated signals represented in the system of polar coordinates.
When an amplitude and phase-modulated carrier signal is generated by an electronic arrangement, two phase-modulated auxiliary signals are first generated from two modulation signals with substantially the same amplitude in order to avoid non-linear distortion. There-after the amplitude and phase-modulated carrier signal is generated by summing the phase-modulated signals by addi-tion or subtraction. The phase of the auxiliary signals are chosen so that the amplitude modulation of the ampli-tude and phase-modulated signal depends on the difference between these phases and depends on the modules of the modulation signals represented in a system of polar co-ordinates, the phase modulation of the amplitude and phase-modulated signal depending on the sum of the phases and on that argument of the phase-modulated signals represented in the system of polar coordinates.
Description
7~ ~
PHN 9972 1 2~-8-1981 "Electronic arrangement for generating an amplitude and phase-modulated carrier signal."
The invention relates to an electronic arrange~
ment for genera-ting an amplitude and phase-modulated car-rier signal.
In radio transmission a great diversity of ampli-tude and phase-modulated carrier signals are employed, such as single-sideband signals, so-called "O~fset Quadra-ture Phase Shift KeyingU signals, abbrevia-ted to OQPSK-signals, and so on.
The most important advantage of signals modula-ted in this manner, compared to signals o~ which only -the am-plitude, the phase or the frequency have been modula-ted resides in the fact that -the bandwidth required for -the transmission of the same quanti-ty of information in ampli tude and phase modulated signals is less than for signals of which only the amplitude, the phase or the frequency is modulated. So, a carrier signal whose ampli-tude and phase are modulated by an information signal, for example ; a single-sideband signal and a L~-phase modulated signal with limited bandwidth respectively, or a OQPSK-modulated signal has half the bandwidth of a carrier signal -whose amplitude is modulated by the same informa-tion signal, such as a double-sideband signal and a narrower bandwidth9 respectively -than a carrier signal wi-thout band limitation which i5 L~-phase modulated by -the same information signal, such as what are commonly referred to as a "Fast Frequency Shift Keying"-signal (FFSK) or a "Minimum Shif-t Keying"-signal (MSK).
Exis-ting sys-tems often employ only phase or ~requency modula-tion, as -the case may be, as the distor-tion of non-~linear, efficient power amplifiers is then only located around multiples of the carrier ~requencies.
This dis-tor-tion can be eliminated in a simple manner sub-sequently by means of fil-tering~ If, however, amplitude ~88~7S~
modula-tion :is also presen-t -then distortion is produced in the in~ormation band itself. A low dis-tortion level is then realized by means of extremely linear amplifiers. These amplifiers are expensive and 'have a low efficiency.
It is an o'bjec-t of -the invention to provide an electronic circuit arrangement for an amplitude and phase-modulated signal whi~h obvia-tes the drawbacks mentioned in the foregoing to a very high 0xtend while maintaining the narrow-band character and which in addi-tion rnakes it pos-sible to realize the modulation at a low-signal level by means of in-tegrated circuits and to form -there~rom a high-power output signal in a very simple and e~icient manner.
According -to the invention there is provided an electronic arrangement for generating an amplitude and 15 phase-modulated carrier signal, the arrangement comprising a phase modulation s-tage having two inputs for receiving two modulation signals, which modulation stage under the control of the modulation signals generates in use two different phase-modulated signals having the same carrier 20 :frequency and subs-tantially the same amplitude, and an ou-tput stage to which the phase-modulated signals are applied for assembling by means of summa-tion of the phase-modulated signals, the ampli-tude and phase-modulated car-rier signal whose ampli-tude is modulated in dependence on 25 the phase difference of the phase-modulated signals and whose phase is modulated in dependence on the sum of the phases of the phase-modulated signals.
The use of -two auxiliary signals in the form of exclusively phase-modulated signals has the advantage -tha-t 30 it is possible to employ non-linear amplification for each of the phase-modulated signals because of the :~act that the information is located in the zero-crossings of the signal the location o:~ which is not adversely affected by non-linear amplification, so that a very high e~ficien-35 cy can be obtained on ampli~ication. In addition, it hasthe advantage that the suppression of the higher order signals generated during the amplification requires only a simple low-pass fil-ter. Furthermore, auxiliary signals 7~3 PHN 997-l 3 25-8-1981 having discrete values can be used, so that the electronic circuit can predominantly be realized in integrated form.
The use o~ substantially equal amplitudes f'or the phase-modulated signals has the advan-tage that not only the phase-modulated signals themselves but also the equip-ment f'or generating -these signals and -the assembly of' said signals to f'orm -the amplitude and phase-modulated signal can be greatly simplified.
Owing to the f'act -tha-t the amplitude and phase-modulated signal is obtained by adding or subtractingthese auxiliary signals having discrete values, but for a low-pass filtering operation, it is furthermore possible, in combina-tion with the use of non-linear amplification to implement a very simple power output stage.
I~ desired, the phase modulation stage may con-vert a first modula-tion signal x1(-t) and a second modula-tion signal x2(-t) into two phase-modulated signals Z1(t) = a cos (wot +~(-t)) and Z2(t) = a cos(wO-t + ~ (t)) the phase ~ (t) of the first phase-modulated signal Z1(t) 20 being equal to y(t) + e arccos (r(-t)/2a) + m2 ~ and wherein the phase ~ (t) of the second phase-modulated signal Z2(t) is equal to ~ (t) - arccos (r(t)/2a) +
n2~ + 2 '~ and for which it holds that ~ = +1 when adding Z1(t) and Z2~t) together to f'orm S(t) and ~2 = -1 25 when sub-tracting Z2(t) from Z1(t) to form S'(-t) and the modula-tion signals x1(t) and x2(t) represen-t -the coor-dinates of a vector in a system of coordina-tes, which vector, transformed in a system of polar coordina-tes by the quantities r(t) and ~ (t), wherein e = + 1, m and n 30 being integers and the amplitude and phase-modulated signal S(t) and S'(-t), respectively being represented by r(t) cos (wot ~ ~ (t)).
The present inven-tion will now be described~
by way of' example, with ref'erence -to the accompanying 35 drawings in which corresponding elements have been given -the same reference numerals, and in which Fig. 1 is a block schematic circuit diagram of' the electronic arrangement in accordance with the invention;
37~8 PHN 9972 l~ 24-8-1981 Fig. 2 shows an embodiment of the phase-modulation stage of the block diagram shown in Fig. 1;
Fig. 3 shows a further embodiment Or the phase modulation stage of the block diagram shown in Fig. 1 with dynamic modulator adaptation;
Fig. 4 shows an embodiment of the output stage of the b]ock diagram shown in Fig. 1 suitable for ana-logue signals;
Fig. 5a shows a diagram in which the ampli-tude and phase-modulated output signal of the block diagram shown in Fig. 1 is represen-ted as a three-level signal;
Fig. 5b is a diagram showing the ~requency spec-trum of the signal shown in Fig. 5a;
Fig. 5c is a diagram of the signal obtained after filtration by means of a low-pass fi]ter of the signal shown in Fig. 5a;
Fig. 5d is a diagram of the frequency spectrum of the signal shown in Fig. 5c;
Fig~ 6 shows an embodiment of the output stage of the block diagram shown in Fig. 1 by means of a vol-tage source suitable for generating the signal shown in Fig.
5a;
Fig. 7 shows an embodiment of the output s-tage of the block diagram shown in Fig. 1 which is a dual arrangement of the embodiment shown in Fig. 6;
Fig. 8 shows a further embodiment of the output stage of the block diagram shown in Fig. 1 by means of current sources suitable for generating the signal shown in Fig. 5a;
Figs. 9a to 9i, inclusive show a number of dia-grams of signals which may occur in the embodimen-ts shown in the Figs. 10 and 11.
Fig. 10 shows a fur-ther embodiment o~ the phase-modula-tion s-tage of -the block diagram shown in Fig. 1 35 suitable for signals having a hard defined level;
Fig. 11 shows a further embodiment of the signal con~ersion arrangement of the embodiment shown in Fig. 10;
~8~3~75~3 PHN 9972 5 25-8-1g81 Fig. 12 shows a furt'her embodiment of the phase modulator and c,~rrier generator of the embodimen-t shown :in Fig. 10; and Fig. 13 is a diagram of the characteristic of the phase-folding arrangement of the embodimen-t shown in Fig. 10.
The block schema-tic circuit diagram of an elec-; tronic arrangement in accordance with the inven-tion, shown in Fig. 1, comprises a phase-modulation s-tage 1 and an output stage 2 connected thereto for genera-ting an ampli-tude and phase-modulated outpu-t signal, For that purpose a firs-t modulation signal x1(t) is applied to a first input 3 of the phase modulation stage and a second modulation signal x2(t) to a second in-l5 put 4 of the modulation stage 1. In the modulation stage 1two phase-modulated signa]s Z,l(t) = a cos(w t + ~ (t)) (1) and Z2(t) - b cos(w t ~ ~ (t)) (2) having the same carrier frequencies are generated by means of the modulation sig-nals in a manner -to be described hereafter. The amplitudes 20 of these phase-modulated signals have here been chosen differently to explain one of the advantages of the ar-rangement in accordance with -the invention. Said phase-modulated signals are applied to the output stage 2 via the outputs 5 and 6 of the modulation stage 1. This output 25 stage comprises a first summing arrangement 7 which as-sembles the phase-modulated signals Z1(t) and Z2(t) by addition or subtraction. The sum signal S(t) = Z1(t) ~
~ Z2(t), wherein ~2 = ~ 1 thus obtained is applied to an output termlnal 8. This sum signal S(t) is an amplitude 30 and phase-modulated signal having the same carrier fre-quency as the phase-modulated signals and is represented by S(t) = r(t) cos( ~ ot ~ ~ (t)) (3) From the signal S( )~-t) = Z1(t) ~ Z2(t) obtained by addi-tion or subtraction, as the case may be, it follows that:
(t) = y(t) ~ ~ arctan ~ (a~b~ - r2(t~
arctan a+b ~ (2 ~ (k-1) ~ (~) ~8~7~3 PHN 9972 6 25-~-1981 and ( ) = y(t) - ~ arc-tan ~¦ la~b) -r2(t) ~ _ ~ V r (t)-(a-b) 5 0 arctan a b ~ (a+b)2-r2(t) 1~ 1 wherein ~ = ~1 for S(t) = Z1(t) + Z2(t) and ~ = -1 for S'(t) = Z1(t) - Z2(t) and wherein ~
is a degree of freedom and may consequently be chosen at op-tion (_l)k~1 = sign r(t) and k and l^are integers. The expressions (4) and (5) are very complex. By choosing in accordance with one of the measures of the invention -the ampli-tudes of the phasemodulated signals Z1(t) and Z2(t) to be equal to each other, for example equal t~
lS and r(t) >~ O, the expressions (1~) and(5) can be simplified to 4 (t) = y (t) ~ 0 arccos (r(t)/2a) + m2/~ (6) and X (t) = ~(t) - O arccos (r(t)/2a) ~ n2~ (7) wherein m and n are integersO
Similar expressions can likewise be derived Por ~ = -1. In 20 the -following description this possibility is no-t worked out in detail~ but can be effected in a simple manner in an identical way as for ~= +1 by a person s~illed in the art.
From the foregoing i-t follows than an amplitude and phase modulated carrier signal with a predetermined 25 carrier frequency in a very simple electronic arrangement may be obtained by summing two phase-modulated carrier sig-nals having the same amplitude and the same carrier frequen-cies. Subtracting the expression (7) from expression (6) shows that -the amplituds modulated of the amplitude and 30 phase-modulated carrier signal is a function of the rela-tive phase difference ( ~(t) - ~ (t)) of the phase-modulated signals and adding the expression (7) -to expression (6) shows that the phase modulated of -the ampli-tude and phase-modula-ted carrier signal is a function of the sum of the 35 phases ( ~(t) + ~(t)) of the phase-modulated signals.
Fig. 2 shows an embodiment wi-th reference to which the generation of -the phase-modulated signals Z1(t) and Z2(t) by means of the modula-tion signals x1(-t) and x2(-t) 375~
PHN 9972 7 2L~ 8-1981 will be explained in greater detai].O
As mo~u:Lation signals a signal r(t) is chosen for the signal x1(-t) and a signal ~ (t) for the si.gnal x2(t), wherein r(-t) and. ~ (t) represen-t the coordinates of a vector in a sys-tem of polar coordinates. The phase modulation stage 1 comprises a signal transforming ar~
rangement 1-1 and a phase moclula-tion arrangement 1-2. ~ia inpu-t -terminal 3 the modulati.on signal r(t) is applied to an arccosine generator 9 of -the signal transforming arrangement 1-1 to form an output signal ~ /2 ~ ~ arccos r(t). Such a generator for generating arcsine r(-t) is known from, for example, -the book "Elec-tronic Analog and Hybrid Computers" by G.A. Korn and T.M.~orn~ more speci-fically from Fig. 8-29(b), arccos r(t) is derived there-from by adding ~r/2 to it. The ou-tput signal of generator 9 and the signal ~ (t) are both applied. to a second summing arrangemen-t 11 and a third summing arrangement 12 for the determination in the second summing arrangement 11 of the sum signal:
~ (t) = ~ (t) ~ ~ arccos (r(-t)) and for the determination in the third sumrning arrangement 12 of the difference signal:
~ (t) = y (t) - ~ arccos (r(t)) The output signals of the signal transforming arrangement 1-1 are applied to the phase modulation arrangement 1-2.
More particularly, the output signal of the summing ar-rangement 11 and 12, respectively, is applied to a phase modulator 13 and lL~, respectively, to which phase-rnodulators a carrier generator 15 is connected for forming the phase-30 modula-ted signals Zl(-t) and Z2(-t). The phase modulators 13 and 1~ together may form, for example, a ~uadrature modula-tor which delivers its output signals from the out-put terminals 5 and 6.
So far the description has been based on a 35 modula-tion s:ignal which forms the vector compon.ents of a vector in a system of polar coordinates. Let it be assumed that the modulation signals x1(-t) and x2(t) are expressed as the signa:L components of a signal vector in a Carthesian PHN 9972 8 2ll-8~1981 coordinate system then there is required between -the input -terminals 3 and 4 and the inputs of the generator 9 and the arrangement 12 a coordinate transforming arrangemen-t~
not shown, which in known manner de-termines on the one hand the modulation envelope r(t) = ~x21(t) ~ x2~(t) of the modulation signals and applies said envelope -to th.e arccosi.ne genera-tor 9~ and on the other hand determines the argument ~(t) = arctg (x2(t)/x1(t)) and applie9 said argument to -the arrangement l2.
An example of an amplitude and phase-modulated carrier signal to be generated by the electronic circuit is a single-sideband signal.
Such a signal may be represented by:
Re (a(t) ~ j â(t)) e i c (8) wherein ei w ot represents the carrier signal and a(t) is the baseband signal, and â(t) -the Hilbert transform of the baseband signal a(-t). Be-tween a signal a(t) and a Hilbert transformed â(t) there is the following relationship:
r+~
20â(t) = rS J ~ dZ (9) --C:D
wherein S = tl for the upper sideband signal and S = -1 for the lower sideband signal, wherein ~ represents the angular frequency of the carrier 25 signal.
The sum signal S(t) shown in expression (3) may be written as:
Re [r(t) ei ~(t) . ei c ~ (10) Comparing the expression (10) with the expression (8) 30 gives that r(t) = ~a (t) ~ â (t) (11) and y (t) = arc-tan (â(t)/a(t)) (12) From the expressions (11) and (12) it follows -that apply-35 ing the base band signal a(t) -to inpu-t terminal 3 as the first modulation signal x1(-t) and applying the lIilbert transform â(t) of -the baseband signal a(t) to input ter-minal 4 as the second modulation signal x2(t) of the 75~
PHN 9972 9 2~-8-l981 embodiment shown in Fig. 2, the saicl coordinate trans-forming arrangement being provicded on -the one hand between -the input -terminals 3 and 4 and on the other hand be-tween the generator 9 and -the arrangement 12, the single-side-band-modula-ted signal is obtained by summing in -the output stage 2 the phase-modulated signals present on -the ter-minals 5 and 6.
It should be no-ted that the IIilber-t transform a(t) of the baseband signal a(-t) can be ob-tained in a pre-stage by shifting -the phase of every frequency component of the baseband signal a(t) over 9O, as is known ~ se from the left-hand parts of the Figs. 4-3 (a) and (b) of the book "Single sideband principles and circuits" by Pappenfus et al, 1964.
lS The phase modulators 13 and 14 generate phase-modulated signals having a spectrum which is wider than the spectrum of the single-sideband signal ob-tained by summation. The spec-tra of the biphase-modulated signals, which spectra are located outside the band, must eliminate each other to an acurate extent, which imposes severe requirements on the modulation stage 1 shown in Fig. 2.
Figo 3 shows a modulation stage embodiment which is very suitable for assembling a single-sideband signal.
Said embodiment is based on a signal shown in Cartesian coordinates with x1(t) = x(t) and x2(t) = y(t)- The signal transforming arrangement 1-1 of this modulation stage 1 comprises, connected between the input terminals 3 and 4, a func-tion generator 16 which assembles from the modulation signals x(t) and y(t) -the signal E(t) = 1/r(t) ~1-(r(t))2 30 wherein r(-t) = ~/ x2(t) ~ y2(-t). Said signal is applied to two mul-tiplying arrangements 17 and 18. The first modula-tion signal x1(t) = x(t) is applied to the multiplying ar-rangement 17 for forming therein the product signal O x(t)E(t) and the second modulation signal x2(t) = y(t) 35 is applied to the second multiplying arrangement 18 for forming therein the product signal O y(t)E(-t).
The modulation signal x(t) is applied together with the product signal ~ y(-t)E(t) to a fourth summing 37~i~
PHN 9972 10 2L~-8-1981 arrangement 21 which ~orms -the di~ference signal 1/2 (x(t) ~ ~ y(t)E(t)) = cos ~ (t) (13) In addition, -the modulation signal y(t) is applied to-gether with the product signal Ox(t)E(t) to a ~i~th summing arrangement 22, which ~orms the sum signal 1/2(y(-t) ~ ex (t)E(t)) = sin ~ (t) (1L~) Likewise the modulation signal x(-t) is applied together with the product signal O~(t)E(t) to a sixth summing arrangement 23 which ~orms -the sum signal 1/2(x(t) * ~ y(t)~(-t)) = cos X (-t) (15) and the modulation signal y(t) is applied -together wi-th the product signal Ox(t)E(t) to a seventh summing arrange-ment 24, which f`orms thedif~erence signal:
l/2(y(t) - ~ x(t)E(t)) = sin (t) (16) The signals cos ~ (t), sin ~(t) and cos X (t) and sin X (t) ~ormed by the summing arrangements 21 to 2L~, inclusive, are applied as output signals o~ the signal -trans~orming arrangement 1-1 to separate phase quadrature modulators 25 and 26 o~ -the phase modula-tion arrangement 1-2. The quadra-ture modulator 25 comprises two multipliers 27 and 28 towhich the respective signals cos ~ (t), sin ~(t) and the carrier signal cos ~ t, generated by a carrier genera-tor 29, and the carrier signal sin ~ t obtained via a 9O
phase-shifting network 3O are applied to ~orm the product 25 signals: cos~'(t). cos ~ t and sin ~ (t). sin ~ t. These signals are subtracted in an eighth summing arrangement 31, which results in the phase-modula-ted signal Z1(t) = 2 COS ( ~ot ~ ~ (t)) at ou-tput terminal 5.
~ilcewise, the quadrature modulator 26 comprises 30 two multipliers 32 and 33 to which -the respective signals cos ~ (t), sinX (t) and cos ~ t and sin ~ t are applied to form the product signals cos X (t)~cos~ -t and sin ~ (t).sin ~ t. These product signals are applied to a ninth summing arrangement 34 in which the phase-modulated 35 signal Z2(t) = ~cos( ~ot ~ (-t)) is ~ormed and supplied from output terminal 6.
In order to achieve that the spec-tra o~ the two phase-modulated signals Z1(t) and Z2(t), which spectra are 75~
PHN 9972 11 2Lr~8~1981 located outside -the band9 accurately elimina-te each o-ther, use is made of--~he fact that in the expressions (4) and (5) ~ may be chosen plus one or minus one and that none of -the branches of -the modulation stage 1 has a memory function. To achieve this, a pulse signal genera-tor 35 having a pulse repe-tition rate which is at least twice as high as the highest frequency of the modulation sig nals is connected to the signal transform arrangement 1-1 of the modulation s-tage 1. Said generator 35 is connected in particular to a third signa] input of all mul-tiplying arrangements 17 and 18. Under the control o* the pulse signal produced by the generator 35 -the value o* the quan-tity is alterna-tively +l and -1 in the rhythm of the pulse repeti~,ion ra-te.
lS This means that the ou-tput signals o* the multi-plying arrangements 17 and 18 change their sign in the rhythm of the pulse repetition rate. This can be achieved by means of, for example, inverting amplifiers which under the con-trol of the pulse signal are alternately 20 switched into and out of signal output circui-ts, not shown, of the multipliers 17 and 18.
As follows from the expressions (13), (14), (15) and (16)~ the output signals of the adding arrange-ments 21 to 2LI, inclusive, ~h~ their positions in 25 response to the alternation o* the value of the signal between the values ~1 ~nd -1.
Thus it *ollows from the formulae (13) and (15) that the output signal of summing arrangement 21 for ~ = +1 is the same as the output signal of summing arrange-30 ment 23 for ~ = -1 and vice versa.
Likewise it *ollows from the formulae (14) and (16) that -the ou-tput signal o* summing arrangement 22 for ~ = ~1 is tne same as the output signal of summing ar-rangement 24 for ~ = -1 and vice versa.
This results in that the quadra-ture modulators 25 and 26 each produce the two phase-modulated signals Zl(t) and Z2(-t), namely alternately in the rhythm of -the pulse rate of generator 35, with the proviso that if the PHN 9972 12 2l~-8-1981 modulator 2L~ produces -the signal Z1(-t)~ the modulator 2 produces the signal Z2(-t) and vice versa. Owing -to the dynamic modulator adapta-tion obtained in this way -the un-wanted spectra produced by the modulators 2L~ and 25 cancel each other for the most part by means of subtraction in the output stage connec-ted to the modulation stage 1.
Fig. 4 shows an outpu-t stage 2 with power ampli-fication which is advantageous for the electronic arrange-ment for analog signals.
The outpu-t stage 2 comprises two power amplifiers 36 and 37 and a hybrid circuit 38 connected thereto to which on the one hand the output load in the form of an aerial 39 and on -the o-ther hand a ma-tching impedance 4O
equal to the aerial impedance are connec-ted.
Power amplification of the phase-modulated sig-nals Z1(t) and Z2(t) is performed in the amplifiers 36 and 37. Owing to the fact that said signals have a constant amplitude, the non-linear amplification obtained during the amplification (higher harmonics of the carrier fre-20 quency) falls outside the band.
In the hybrid circuit the amplified signals Z1(t) and Z2(t) are assembled to form the amplitude and phase-modulated signal S(t) = r(t).cos( ~ t ~ ~(t)), wherein r(t) is the first modulation signal x1(t) and ~ (t) the second modula-tion signal x2(t). The signal S(t) is applied in i-ts -totality to the aerial 39. When matched properly the hybrîd ensures that the non-linear amplifiers do not see each other. This pre-vents cross~modulation of the two phase-modulated signals.
It is alternatively possible -to use,instead of the arrangements 16 to 24, inclusive so-called "look up tables" which comprise memories in which -the values of the output signals of the adder arrangements 21 to 2LI, inclusive are formed ~or a large number of discrete values 35 of the input signals x1(t) and x2(t). The values of these input signals form the addresses for the output signals which are associated wi-th said input signals, of the adder arrangements 21 to 24, inclusive. The memory arrangemen-t 7S~
further comprises an addressing arrangement which under the control of'-the modulation signals reads the signals corresponding therewi-th and which are proportional to -the phase signals ~ (t) and ~ (t). This may result in a saving in equipment.
It should be noted that in addition to SSB other forms of amplitude and phase-modula-ted signals, such as VSB, may alternatively be generated in the above-described manner. The only differen-t for VSB compared with SSB is -that slightly different fil-ters must be used at the base-band level for the generation of the -two modulation signals required for VSB.
~ s mentioned in the foregoing, the generation o*
an amplitude and phase-modulated signal from two modulation signals by means of two auxiliary signals in the form of phase-modula-ted signals furnishes -the considerable advantage that non-linear distortions in -the power amplifiers have no influence on -the loca-tion of the zero-crossings of the phase-modulated signals. This makes it possible to 20 use any type of amplifier.
If two phase-modulated signals Z1(t) and Z2(t) are passed -through circuits which are commonly referred to as hard limiter circuits then there are obtained from said signals square-wave signals sign Z1(t) and sign Z2(t) 25 which in this example are assembled in the outpu-t stage to form a difference signal sign S'(t) = sign (cos ~ t + ~(t) ~ ~ arccos (r(-t)))) ~
+ sign (cos W ot ~(t) - ~ arccos (r(t)))) (17) in accordance with -the expression r cos(~ ot + ~(t)) = 2 COS (w ot ~ ~(t)) ~ 2 COS (~ t~ X(t)), which has a different construction in order to generate a phase and ampli-tude-modulated signal from two phase-modula-ted signals.
Said signal sign S(t) is a three-level signal~
35 owing to the hard lirnitation of the sub-signals, as is shown in Fig. 5a.
The original amplitude and phase-modulated signal can be recovered therefrom by means of a low-pass filter, 5~3 PHN 9972 1L~ 24-8-1981 which will be demons-trated in the following calculation.
It now holds that:
sign (cos ~ot ~ ~ (t) + 0 arccos r(t))) =
k~co 5 4 ~ ( 1)(k-1)/2 cos k(~ o-t ~ ~ (t) ~ H arccos r(t)) IJ ~ . _ ~
k=1,3,5--- k and (18) sign (cos ~ t ~ y (t) - ~ arccos r(t))) =
= 4 ~ ( 1)(k-1)/2 cos k(~ ot ~ ~'(t)- ~ arccos r(t)) k=1,3,5,....................................... ~19) In the output stage the signal represented by expression (19) is, for example, added to the signal represented by expression ~18') which results in -tha-t ( ) ~ ~ ( 1)(k-1)/2 CS(k arCcOs r(t k=1,3~5~o-cos k ( ~ ot ~ ~(t)) (20) 20 1~riting this sum in individual terms givesS(t) =~ r(-t) cos( ~ot ~ ~(t)) - ~ ~
cos ~( ~ ot ~ ~(t)) + -- (21) Fig. 5b shows the frequency spectrum of this signal.
The use of a low-pass filter then results in that S(t) = r(t) cos( ~ t -~ ~(t)), which signal and asso-cia-ted frequency spectrum are shown in the Figs. 5c and 30 5d.
It is assumed that ~ 0 is sufficiently high to prevent the occurrence of what is commonly referred to as fold-over.
So it is possible to use class "d" amplifiers 35 in cornbination with a low-pass filter in -the aerial lead to provide a very high efficiency.
very ad-vantageous output stage with power amplification is shown in Fig. 6. The square-wave signal PHN 9972 15 Z4~
sign Z1(t) ob-tained by means of hard limiting o~ the signal Z1(t) is-~pplied to a control terminal 41 and -the square-wave signal sign Z2(t) obtained by means of hard limiting of the signal Z2(t) is applied to a control ter-minal 43 o~ change-over switches 42 and 44, respectively.
0~ each change-over switch 42 ancl 44, respectively one o~
the change-over contacts 42-1 and 44~2a respectively :is connected to a first terminal ~E o~ a voltage source having a value of 2E volts and the other change-over con-tact 42-2 and 44-1, respectively is connected to a second terminal -E of the vol-tage source. The swi-tching arms 42-3 and 44-3 o~ the change-over switches 22 and 44 are in-ter-connected via a primary winding 45-1 of an ou-tput trans~
~ormer 45. An aerial may be connected to terminals 46 and 47 o~ a secondary winding 45-2 of the trans~ormer 45.
Said output stage 2 opera-tes as follows.
If a value ~'1" is applied to the control ter-minals 41 and 43 then the two change-over switches 42 and 44 are in the positions not shown and a current ~lows ~rom 20 +E -to -E via contact 42-1, switching arm 42-3, the primary winding 45-1, switching arm 44-3 and contact 44-1. This current induces a voltage in the secondary winding 45-2 which can be taken of~ between the terminals 46 and 47.
I~ the value ~'0" is applied to the two control 25 terminals 41 and 43 then the change-over swi-tches are in the posi-tions shown7 causing a current to ~low from +E
to -E via contact 44-2, switching arm 44-3, the primary winding 45-1, switching arm 42-3 and contact 42-2. Con sequently, the sign o~ the voltage be-tween the -terminals 30 46 and 47 is reversed. I~ the value "1" is applied to one of the con-trol terminals 42 or 43 and the value "0"
to the other terminal or vice versa then the two swit-.. .. . ..
ching arms 42-3 and 44-3 are connected to ei-ther the terminal +E or the terminal -E and there is no voltage 35 across the primar-y winding. The voltage between the ter-minals 46 and 47 is then zero Volts.
Fig. 7 shows a dual outpu-t stage in which a current source 10 is used instead of a voltage source. At -this output 7~3 P~IN 9972 16 24-8-1981 stage one end of -the primary winding of trans~ormer 45 is connected to a-contact of the change-over switch L~2 as well as to a contac-t o~ the change-over swi-tch 44, the other end of the primary winding is connected to the other contact of the change-over switches 42 and 44 and the current source 10 is connected be-tween the switching arms of the change-over switches 42 and 44. This ou-tput stage operates in a similar manner as -the embodiment shown in Fig. 6, with the proviso that the current flowing through the primary winding of the transformer 45 is produced by the current source 10 instead of by the voltage source.
An advantageous embodiment of the change-over switches 42 and 44 of the output stages shown in the Figs. 6 and 7 is realized with the aid of, for example, MES or MOSFETS.
Fig. 8 shows an other output stage realized by means of current sources. The hard limited phase-modulated signal sign Z1(t) and sign Z2~t) applied by -the modulation stage 1 to the output stage are applied to three logic gate circuits, namely a first AND-gate circuit 48y an 20 exclusive-NOR-gate circuit 49 and a second AND-gate cir-cuit 50 having two inverting inpu-ts, which generate the respective signals sign Z1(t).sign Z2(t); the signal 1 g 2(t) ~ sign Z1(t).sign Z2(t) and the signal sign Z1(b). ~ -These signals are applied to a three-stage flip-flop circuit 60 as set signals on the positive edges~
which circuit comprises current sources 61 and 62 con nected to outputs 60-1 and 60-2. These curren-t sources 61 and 62 are connected to an aerial 63. Under the control 30 Of the output signal produced by the logic circui-t 48 the current sources 61 and 62 are s~itched on on a posi-tive edge o~ said ou-tput signal b~- the -three-s-tate flip-~lop circuit 60, which current sources then apply a current having a value 2 I to the aerial 63.
Under the control of the ou-tpu-t signal produced by the logic circuit 49 only current source 61 and 62 is switched on on a positive edge of said output signal by the three-state flip-flop circuit 60 in response to which 75~
only a current having a value I is applied to the aerial 63. Finally under the control of the output signal produced by the logic circui-t 50 the -two curren-t sources 61 and 62 are switched off on a positive edge of said signal by the three-state flip-flop circuit 60~ so that no current is applied to the aerial.
The aerial cable itself may have a low-pass characteristic so that in that even-t a discrete low-pass filter is no-t required~ as is shown in the Figs. 6, 7 and 8.
For -those outpu-t s-tages 2, which are controlled by phase-modulated signals having discrete values Figs.
10 and 11 show very advantageous modulation stages 1. For a more detailed description of these modulation stages it will first be demonstrated with reference to Fig. 9 that there is a relationship between the product of the hard limited phase-modulated signals sign Z1(t) and sign Z2(t) and the product of two pulse-duration modulated signals P1(t) and P2(t) with natural sampling.
The first hard limited phase-modulated signal is sign Z1(t) = sign (cos ~ot ~ (t) ~ 0 arccos r(t))) If, for example 0 is chosen equal to +1 then it may be written that:
sign Z1(t) = sign (cos ~ot ~ ~ (t~ ~ arcsin r(t) + 2 )) 25 = -sign (sin ~ t ~ ~ (t) ~ arcsin r(-t))) which signal is shown in Fig. 9d.
The second hard limited phase-modulated signal is sign Z2(t? = sign (cos ~Ot + ~ (-t) - 0 arccos r(t))) 30 = sign (C09 C~o-t ~ y(t) - arcsin r(t) ~ ~2 )) = sign (sin w t + y(t) - arcsin r(t))) which signal is shown in Fig. 9e.
The product of -the hard limited phase-modulated signals is consequently equal to:
-sign (sin( ~ Ot~ ~(t)~ arcsin r(t))).sign(sin(~ t~ ~(t) --arcsin r(t))) This may be written as:
1~387~
PIIN 9972 18 2L~-8-1981 -sign(sirl( ~ t+ ~(t)~arcsin r(t)).Sill(~ ot~ ~(t)-arcsin r(t))) which correspon~s to:
-sign(cos(2 arcsin r(t))-cos 2(~ t ~ y (t))) =
-sign(1-2 sin2(arcsin2r(t))-1+2 sin2(~ t ~ ~(t)) =
5 -sign(sin2(~ t~ Y(t)) - r2(t)) (22) Expression (22) may be written as:
sign(r(t)+sin( ~ t~ ~(t))).sign(r(t)-si~ t~ ~(t))) with which it is proved that sign (z1(t)).sign (Z2(t)) = P1(t) ' 2( ) (23) wherein Pl(t) = sign (r(t) ~ sin(~Ot + ~ (t))) and P2(t) = sign (r(t) - sin(W -t ~-y (t))) i5 .
From (23) it follows that -the zero-crossings of the product of the hard limited phase-modulated signals sign Z1(t) and sign Z2(t) are identical to -the zero-crossings of the 15 product of the pulse-width modulated P1(t).P2(t).
In Fig. 9a the function sin( ~ t +~(t)) (solid line) the function -~r(t) (dot dash line) and the function -r(t) (dashed line) are shown.
The sign of the function r(t)~sin( ~ t + ~(t)) 20 which represents -the pulse-modulated signal P1(t) is shown in Fig. 9b. The values for which sin( ~ t ~ y (t)) is equal to -r(t) form the transitions of the pulse-modulated signal P1(t) which points of intersection are designated by "b" in Fig. 9a. Likewise the poin-ts where the signal 25 +sin( ~ ot ~ ~'(-t)) intersects the signal +r(t) form the transitions of the pulse-modula-ted signal P2(t), -which signal P2(t) is shown in Fig. 9c, use being made of -the fact that r(t)-sin( ~ Ot~ ~ ~t)) = -(sin( ~ Ot ~ y (t)) - r(t)) 30 These points of intersection are designated "c" in Fig~ 9a.
As has been demonstrated in the foregoing~ the edges of these signals coincide with the edges of the pulse-modulated signals Pl(-t) and P2(t) 9 it appears how-ever that the ascending edges of the signal sign Z1(-t) 35 coincide with -the ascending edges of the signal P2(t) and the descending edges of -the signal Z1(-t) coincide with the ascending edges of the signal P1(t), which in Fig. 9a is shown by providing the relevant poin-ts of intersection 5~3 P~IN 9972 19 2ll-8-1981 wi-th a second designation "d1'. I-t likewise appears that the ascending edges of -the signal sign Z2(t) coincide with the descending edges of the signal P2(t) and the descending edges of the signal sign Z1(t) coincide with the descending edges of the signal P1(t)~ which is shown in Fig. 9a by providing the relevant points of in-tersection with a second designation "e".
The embodiments shown in Figs. 10 and 11 of` a modulation stage 1 utilize the above-mentioned property, Thus, Fig. 10 shows a modulate stage 1, in which a modula-tion signal ~(-t) applied to an input terminal 4 first se .~ modulates in a p~ modulator 64 -the phase of a carrier signal produced by a carrier generator 65. The signal sin( ~ t + ~(-t)) thus obtained is applied, together with a modula-tion signal r(t) applied to the input terminal 3, to terminals 66-2 and 66-1 of a signal converter arrange-ment 66. In arrangement 66 these signals are applied to signal inputs 67-1 and 67-2 of a first comparator circuit 67 and also to signal inputs 68-1 and 68 2 of a second comparator circuit 68. These comparator circuits 67 and 68 differ only in that the signal input 68-2 of circuit 68 is a signal-inverting input while -the corresponding input 61-2 of circuit 67 is a signal non-inverting input.
The output signals of -the comparator circuits 67 and 68 are binary so that comparator circuit 67 produces the pulse-modulated signal P1(t), shown in Fig. 9b and com-parator circuit 68 produces -the pulse modulated sigrnal P2(t), shown in Fig. 9c. The signal P1(t) is applied -to a divide-by-two circuit 69 and the signal P2(t) to a divide-30 by-two circuit 70. The output signal at the signal output a of the divide-by--two circuit 69 changes its value at every ascending edge of signal P1(t) as is shown in Fig.
9f and -the ou-tpu-t signal from the output b of said divide-by-two eireuit 69 changes its value at every deseending edge of signal P1(t) as shown in Fig. 9g.
Likewise the signal on signal output a of the divide-by-two eireuit 70 ehanges at every aseending edge of signal P2(t) as shown in Fig. 9h and the signal on signal output b of the divide-by-two circuit 70 changes at every desc-encl-ing edge of signal P2(t) as shown in Fig. 9i. The signal supplied from outpu-t a of the divide-by--two circui-t 69 and the signal supplied from outpu-t a of the divide-by-two circui-t 70 are applied to an exclu~
sive-OR~circuit which forms from the signal sign Z1(t) shown in Fig. 9d, which can be taken off -from output 5.
Likewise the signal f`rom output b of the divide-by-two circuit 70 and the signal from output b of the divide-by-two circuit 69 are applied to an exclusive /'OR"-circuit 72, which forms from these signals -the signal sign Z2(t) shown in Fig. 9e, which can be taken from output 6.
Owing to the fact that -the initial state of the divide-by-lS two circuits is not defined, said signal converter ar-rangement 66 comprising divide-by-two circuits 6~ and 70 creates an uncer-tain-ty as regards the sign of the ou-tput signal sign S(t) obtained by summation in the output s-tage
PHN 9972 1 2~-8-1981 "Electronic arrangement for generating an amplitude and phase-modulated carrier signal."
The invention relates to an electronic arrange~
ment for genera-ting an amplitude and phase-modulated car-rier signal.
In radio transmission a great diversity of ampli-tude and phase-modulated carrier signals are employed, such as single-sideband signals, so-called "O~fset Quadra-ture Phase Shift KeyingU signals, abbrevia-ted to OQPSK-signals, and so on.
The most important advantage of signals modula-ted in this manner, compared to signals o~ which only -the am-plitude, the phase or the frequency have been modula-ted resides in the fact that -the bandwidth required for -the transmission of the same quanti-ty of information in ampli tude and phase modulated signals is less than for signals of which only the amplitude, the phase or the frequency is modulated. So, a carrier signal whose ampli-tude and phase are modulated by an information signal, for example ; a single-sideband signal and a L~-phase modulated signal with limited bandwidth respectively, or a OQPSK-modulated signal has half the bandwidth of a carrier signal -whose amplitude is modulated by the same informa-tion signal, such as a double-sideband signal and a narrower bandwidth9 respectively -than a carrier signal wi-thout band limitation which i5 L~-phase modulated by -the same information signal, such as what are commonly referred to as a "Fast Frequency Shift Keying"-signal (FFSK) or a "Minimum Shif-t Keying"-signal (MSK).
Exis-ting sys-tems often employ only phase or ~requency modula-tion, as -the case may be, as the distor-tion of non-~linear, efficient power amplifiers is then only located around multiples of the carrier ~requencies.
This dis-tor-tion can be eliminated in a simple manner sub-sequently by means of fil-tering~ If, however, amplitude ~88~7S~
modula-tion :is also presen-t -then distortion is produced in the in~ormation band itself. A low dis-tortion level is then realized by means of extremely linear amplifiers. These amplifiers are expensive and 'have a low efficiency.
It is an o'bjec-t of -the invention to provide an electronic circuit arrangement for an amplitude and phase-modulated signal whi~h obvia-tes the drawbacks mentioned in the foregoing to a very high 0xtend while maintaining the narrow-band character and which in addi-tion rnakes it pos-sible to realize the modulation at a low-signal level by means of in-tegrated circuits and to form -there~rom a high-power output signal in a very simple and e~icient manner.
According -to the invention there is provided an electronic arrangement for generating an amplitude and 15 phase-modulated carrier signal, the arrangement comprising a phase modulation s-tage having two inputs for receiving two modulation signals, which modulation stage under the control of the modulation signals generates in use two different phase-modulated signals having the same carrier 20 :frequency and subs-tantially the same amplitude, and an ou-tput stage to which the phase-modulated signals are applied for assembling by means of summa-tion of the phase-modulated signals, the ampli-tude and phase-modulated car-rier signal whose ampli-tude is modulated in dependence on 25 the phase difference of the phase-modulated signals and whose phase is modulated in dependence on the sum of the phases of the phase-modulated signals.
The use of -two auxiliary signals in the form of exclusively phase-modulated signals has the advantage -tha-t 30 it is possible to employ non-linear amplification for each of the phase-modulated signals because of the :~act that the information is located in the zero-crossings of the signal the location o:~ which is not adversely affected by non-linear amplification, so that a very high e~ficien-35 cy can be obtained on ampli~ication. In addition, it hasthe advantage that the suppression of the higher order signals generated during the amplification requires only a simple low-pass fil-ter. Furthermore, auxiliary signals 7~3 PHN 997-l 3 25-8-1981 having discrete values can be used, so that the electronic circuit can predominantly be realized in integrated form.
The use o~ substantially equal amplitudes f'or the phase-modulated signals has the advan-tage that not only the phase-modulated signals themselves but also the equip-ment f'or generating -these signals and -the assembly of' said signals to f'orm -the amplitude and phase-modulated signal can be greatly simplified.
Owing to the f'act -tha-t the amplitude and phase-modulated signal is obtained by adding or subtractingthese auxiliary signals having discrete values, but for a low-pass filtering operation, it is furthermore possible, in combina-tion with the use of non-linear amplification to implement a very simple power output stage.
I~ desired, the phase modulation stage may con-vert a first modula-tion signal x1(-t) and a second modula-tion signal x2(-t) into two phase-modulated signals Z1(t) = a cos (wot +~(-t)) and Z2(t) = a cos(wO-t + ~ (t)) the phase ~ (t) of the first phase-modulated signal Z1(t) 20 being equal to y(t) + e arccos (r(-t)/2a) + m2 ~ and wherein the phase ~ (t) of the second phase-modulated signal Z2(t) is equal to ~ (t) - arccos (r(t)/2a) +
n2~ + 2 '~ and for which it holds that ~ = +1 when adding Z1(t) and Z2~t) together to f'orm S(t) and ~2 = -1 25 when sub-tracting Z2(t) from Z1(t) to form S'(-t) and the modula-tion signals x1(t) and x2(t) represen-t -the coor-dinates of a vector in a system of coordina-tes, which vector, transformed in a system of polar coordina-tes by the quantities r(t) and ~ (t), wherein e = + 1, m and n 30 being integers and the amplitude and phase-modulated signal S(t) and S'(-t), respectively being represented by r(t) cos (wot ~ ~ (t)).
The present inven-tion will now be described~
by way of' example, with ref'erence -to the accompanying 35 drawings in which corresponding elements have been given -the same reference numerals, and in which Fig. 1 is a block schematic circuit diagram of' the electronic arrangement in accordance with the invention;
37~8 PHN 9972 l~ 24-8-1981 Fig. 2 shows an embodiment of the phase-modulation stage of the block diagram shown in Fig. 1;
Fig. 3 shows a further embodiment Or the phase modulation stage of the block diagram shown in Fig. 1 with dynamic modulator adaptation;
Fig. 4 shows an embodiment of the output stage of the b]ock diagram shown in Fig. 1 suitable for ana-logue signals;
Fig. 5a shows a diagram in which the ampli-tude and phase-modulated output signal of the block diagram shown in Fig. 1 is represen-ted as a three-level signal;
Fig. 5b is a diagram showing the ~requency spec-trum of the signal shown in Fig. 5a;
Fig. 5c is a diagram of the signal obtained after filtration by means of a low-pass fi]ter of the signal shown in Fig. 5a;
Fig. 5d is a diagram of the frequency spectrum of the signal shown in Fig. 5c;
Fig~ 6 shows an embodiment of the output stage of the block diagram shown in Fig. 1 by means of a vol-tage source suitable for generating the signal shown in Fig.
5a;
Fig. 7 shows an embodiment of the output s-tage of the block diagram shown in Fig. 1 which is a dual arrangement of the embodiment shown in Fig. 6;
Fig. 8 shows a further embodiment of the output stage of the block diagram shown in Fig. 1 by means of current sources suitable for generating the signal shown in Fig. 5a;
Figs. 9a to 9i, inclusive show a number of dia-grams of signals which may occur in the embodimen-ts shown in the Figs. 10 and 11.
Fig. 10 shows a fur-ther embodiment o~ the phase-modula-tion s-tage of -the block diagram shown in Fig. 1 35 suitable for signals having a hard defined level;
Fig. 11 shows a further embodiment of the signal con~ersion arrangement of the embodiment shown in Fig. 10;
~8~3~75~3 PHN 9972 5 25-8-1g81 Fig. 12 shows a furt'her embodiment of the phase modulator and c,~rrier generator of the embodimen-t shown :in Fig. 10; and Fig. 13 is a diagram of the characteristic of the phase-folding arrangement of the embodimen-t shown in Fig. 10.
The block schema-tic circuit diagram of an elec-; tronic arrangement in accordance with the inven-tion, shown in Fig. 1, comprises a phase-modulation s-tage 1 and an output stage 2 connected thereto for genera-ting an ampli-tude and phase-modulated outpu-t signal, For that purpose a firs-t modulation signal x1(t) is applied to a first input 3 of the phase modulation stage and a second modulation signal x2(t) to a second in-l5 put 4 of the modulation stage 1. In the modulation stage 1two phase-modulated signa]s Z,l(t) = a cos(w t + ~ (t)) (1) and Z2(t) - b cos(w t ~ ~ (t)) (2) having the same carrier frequencies are generated by means of the modulation sig-nals in a manner -to be described hereafter. The amplitudes 20 of these phase-modulated signals have here been chosen differently to explain one of the advantages of the ar-rangement in accordance with -the invention. Said phase-modulated signals are applied to the output stage 2 via the outputs 5 and 6 of the modulation stage 1. This output 25 stage comprises a first summing arrangement 7 which as-sembles the phase-modulated signals Z1(t) and Z2(t) by addition or subtraction. The sum signal S(t) = Z1(t) ~
~ Z2(t), wherein ~2 = ~ 1 thus obtained is applied to an output termlnal 8. This sum signal S(t) is an amplitude 30 and phase-modulated signal having the same carrier fre-quency as the phase-modulated signals and is represented by S(t) = r(t) cos( ~ ot ~ ~ (t)) (3) From the signal S( )~-t) = Z1(t) ~ Z2(t) obtained by addi-tion or subtraction, as the case may be, it follows that:
(t) = y(t) ~ ~ arctan ~ (a~b~ - r2(t~
arctan a+b ~ (2 ~ (k-1) ~ (~) ~8~7~3 PHN 9972 6 25-~-1981 and ( ) = y(t) - ~ arc-tan ~¦ la~b) -r2(t) ~ _ ~ V r (t)-(a-b) 5 0 arctan a b ~ (a+b)2-r2(t) 1~ 1 wherein ~ = ~1 for S(t) = Z1(t) + Z2(t) and ~ = -1 for S'(t) = Z1(t) - Z2(t) and wherein ~
is a degree of freedom and may consequently be chosen at op-tion (_l)k~1 = sign r(t) and k and l^are integers. The expressions (4) and (5) are very complex. By choosing in accordance with one of the measures of the invention -the ampli-tudes of the phasemodulated signals Z1(t) and Z2(t) to be equal to each other, for example equal t~
lS and r(t) >~ O, the expressions (1~) and(5) can be simplified to 4 (t) = y (t) ~ 0 arccos (r(t)/2a) + m2/~ (6) and X (t) = ~(t) - O arccos (r(t)/2a) ~ n2~ (7) wherein m and n are integersO
Similar expressions can likewise be derived Por ~ = -1. In 20 the -following description this possibility is no-t worked out in detail~ but can be effected in a simple manner in an identical way as for ~= +1 by a person s~illed in the art.
From the foregoing i-t follows than an amplitude and phase modulated carrier signal with a predetermined 25 carrier frequency in a very simple electronic arrangement may be obtained by summing two phase-modulated carrier sig-nals having the same amplitude and the same carrier frequen-cies. Subtracting the expression (7) from expression (6) shows that -the amplituds modulated of the amplitude and 30 phase-modulated carrier signal is a function of the rela-tive phase difference ( ~(t) - ~ (t)) of the phase-modulated signals and adding the expression (7) -to expression (6) shows that the phase modulated of -the ampli-tude and phase-modula-ted carrier signal is a function of the sum of the 35 phases ( ~(t) + ~(t)) of the phase-modulated signals.
Fig. 2 shows an embodiment wi-th reference to which the generation of -the phase-modulated signals Z1(t) and Z2(t) by means of the modula-tion signals x1(-t) and x2(-t) 375~
PHN 9972 7 2L~ 8-1981 will be explained in greater detai].O
As mo~u:Lation signals a signal r(t) is chosen for the signal x1(-t) and a signal ~ (t) for the si.gnal x2(t), wherein r(-t) and. ~ (t) represen-t the coordinates of a vector in a sys-tem of polar coordinates. The phase modulation stage 1 comprises a signal transforming ar~
rangement 1-1 and a phase moclula-tion arrangement 1-2. ~ia inpu-t -terminal 3 the modulati.on signal r(t) is applied to an arccosine generator 9 of -the signal transforming arrangement 1-1 to form an output signal ~ /2 ~ ~ arccos r(t). Such a generator for generating arcsine r(-t) is known from, for example, -the book "Elec-tronic Analog and Hybrid Computers" by G.A. Korn and T.M.~orn~ more speci-fically from Fig. 8-29(b), arccos r(t) is derived there-from by adding ~r/2 to it. The ou-tput signal of generator 9 and the signal ~ (t) are both applied. to a second summing arrangemen-t 11 and a third summing arrangement 12 for the determination in the second summing arrangement 11 of the sum signal:
~ (t) = ~ (t) ~ ~ arccos (r(-t)) and for the determination in the third sumrning arrangement 12 of the difference signal:
~ (t) = y (t) - ~ arccos (r(t)) The output signals of the signal transforming arrangement 1-1 are applied to the phase modulation arrangement 1-2.
More particularly, the output signal of the summing ar-rangement 11 and 12, respectively, is applied to a phase modulator 13 and lL~, respectively, to which phase-rnodulators a carrier generator 15 is connected for forming the phase-30 modula-ted signals Zl(-t) and Z2(-t). The phase modulators 13 and 1~ together may form, for example, a ~uadrature modula-tor which delivers its output signals from the out-put terminals 5 and 6.
So far the description has been based on a 35 modula-tion s:ignal which forms the vector compon.ents of a vector in a system of polar coordinates. Let it be assumed that the modulation signals x1(-t) and x2(t) are expressed as the signa:L components of a signal vector in a Carthesian PHN 9972 8 2ll-8~1981 coordinate system then there is required between -the input -terminals 3 and 4 and the inputs of the generator 9 and the arrangement 12 a coordinate transforming arrangemen-t~
not shown, which in known manner de-termines on the one hand the modulation envelope r(t) = ~x21(t) ~ x2~(t) of the modulation signals and applies said envelope -to th.e arccosi.ne genera-tor 9~ and on the other hand determines the argument ~(t) = arctg (x2(t)/x1(t)) and applie9 said argument to -the arrangement l2.
An example of an amplitude and phase-modulated carrier signal to be generated by the electronic circuit is a single-sideband signal.
Such a signal may be represented by:
Re (a(t) ~ j â(t)) e i c (8) wherein ei w ot represents the carrier signal and a(t) is the baseband signal, and â(t) -the Hilbert transform of the baseband signal a(-t). Be-tween a signal a(t) and a Hilbert transformed â(t) there is the following relationship:
r+~
20â(t) = rS J ~ dZ (9) --C:D
wherein S = tl for the upper sideband signal and S = -1 for the lower sideband signal, wherein ~ represents the angular frequency of the carrier 25 signal.
The sum signal S(t) shown in expression (3) may be written as:
Re [r(t) ei ~(t) . ei c ~ (10) Comparing the expression (10) with the expression (8) 30 gives that r(t) = ~a (t) ~ â (t) (11) and y (t) = arc-tan (â(t)/a(t)) (12) From the expressions (11) and (12) it follows -that apply-35 ing the base band signal a(t) -to inpu-t terminal 3 as the first modulation signal x1(-t) and applying the lIilbert transform â(t) of -the baseband signal a(t) to input ter-minal 4 as the second modulation signal x2(t) of the 75~
PHN 9972 9 2~-8-l981 embodiment shown in Fig. 2, the saicl coordinate trans-forming arrangement being provicded on -the one hand between -the input -terminals 3 and 4 and on the other hand be-tween the generator 9 and -the arrangement 12, the single-side-band-modula-ted signal is obtained by summing in -the output stage 2 the phase-modulated signals present on -the ter-minals 5 and 6.
It should be no-ted that the IIilber-t transform a(t) of the baseband signal a(-t) can be ob-tained in a pre-stage by shifting -the phase of every frequency component of the baseband signal a(t) over 9O, as is known ~ se from the left-hand parts of the Figs. 4-3 (a) and (b) of the book "Single sideband principles and circuits" by Pappenfus et al, 1964.
lS The phase modulators 13 and 14 generate phase-modulated signals having a spectrum which is wider than the spectrum of the single-sideband signal ob-tained by summation. The spec-tra of the biphase-modulated signals, which spectra are located outside the band, must eliminate each other to an acurate extent, which imposes severe requirements on the modulation stage 1 shown in Fig. 2.
Figo 3 shows a modulation stage embodiment which is very suitable for assembling a single-sideband signal.
Said embodiment is based on a signal shown in Cartesian coordinates with x1(t) = x(t) and x2(t) = y(t)- The signal transforming arrangement 1-1 of this modulation stage 1 comprises, connected between the input terminals 3 and 4, a func-tion generator 16 which assembles from the modulation signals x(t) and y(t) -the signal E(t) = 1/r(t) ~1-(r(t))2 30 wherein r(-t) = ~/ x2(t) ~ y2(-t). Said signal is applied to two mul-tiplying arrangements 17 and 18. The first modula-tion signal x1(t) = x(t) is applied to the multiplying ar-rangement 17 for forming therein the product signal O x(t)E(t) and the second modulation signal x2(t) = y(t) 35 is applied to the second multiplying arrangement 18 for forming therein the product signal O y(t)E(-t).
The modulation signal x(t) is applied together with the product signal ~ y(-t)E(t) to a fourth summing 37~i~
PHN 9972 10 2L~-8-1981 arrangement 21 which ~orms -the di~ference signal 1/2 (x(t) ~ ~ y(t)E(t)) = cos ~ (t) (13) In addition, -the modulation signal y(t) is applied to-gether with the product signal Ox(t)E(t) to a ~i~th summing arrangement 22, which ~orms the sum signal 1/2(y(-t) ~ ex (t)E(t)) = sin ~ (t) (1L~) Likewise the modulation signal x(-t) is applied together with the product signal O~(t)E(t) to a sixth summing arrangement 23 which ~orms -the sum signal 1/2(x(t) * ~ y(t)~(-t)) = cos X (-t) (15) and the modulation signal y(t) is applied -together wi-th the product signal Ox(t)E(t) to a seventh summing arrange-ment 24, which f`orms thedif~erence signal:
l/2(y(t) - ~ x(t)E(t)) = sin (t) (16) The signals cos ~ (t), sin ~(t) and cos X (t) and sin X (t) ~ormed by the summing arrangements 21 to 2L~, inclusive, are applied as output signals o~ the signal -trans~orming arrangement 1-1 to separate phase quadrature modulators 25 and 26 o~ -the phase modula-tion arrangement 1-2. The quadra-ture modulator 25 comprises two multipliers 27 and 28 towhich the respective signals cos ~ (t), sin ~(t) and the carrier signal cos ~ t, generated by a carrier genera-tor 29, and the carrier signal sin ~ t obtained via a 9O
phase-shifting network 3O are applied to ~orm the product 25 signals: cos~'(t). cos ~ t and sin ~ (t). sin ~ t. These signals are subtracted in an eighth summing arrangement 31, which results in the phase-modula-ted signal Z1(t) = 2 COS ( ~ot ~ ~ (t)) at ou-tput terminal 5.
~ilcewise, the quadrature modulator 26 comprises 30 two multipliers 32 and 33 to which -the respective signals cos ~ (t), sinX (t) and cos ~ t and sin ~ t are applied to form the product signals cos X (t)~cos~ -t and sin ~ (t).sin ~ t. These product signals are applied to a ninth summing arrangement 34 in which the phase-modulated 35 signal Z2(t) = ~cos( ~ot ~ (-t)) is ~ormed and supplied from output terminal 6.
In order to achieve that the spec-tra o~ the two phase-modulated signals Z1(t) and Z2(t), which spectra are 75~
PHN 9972 11 2Lr~8~1981 located outside -the band9 accurately elimina-te each o-ther, use is made of--~he fact that in the expressions (4) and (5) ~ may be chosen plus one or minus one and that none of -the branches of -the modulation stage 1 has a memory function. To achieve this, a pulse signal genera-tor 35 having a pulse repe-tition rate which is at least twice as high as the highest frequency of the modulation sig nals is connected to the signal transform arrangement 1-1 of the modulation s-tage 1. Said generator 35 is connected in particular to a third signa] input of all mul-tiplying arrangements 17 and 18. Under the control o* the pulse signal produced by the generator 35 -the value o* the quan-tity is alterna-tively +l and -1 in the rhythm of the pulse repeti~,ion ra-te.
lS This means that the ou-tput signals o* the multi-plying arrangements 17 and 18 change their sign in the rhythm of the pulse repetition rate. This can be achieved by means of, for example, inverting amplifiers which under the con-trol of the pulse signal are alternately 20 switched into and out of signal output circui-ts, not shown, of the multipliers 17 and 18.
As follows from the expressions (13), (14), (15) and (16)~ the output signals of the adding arrange-ments 21 to 2LI, inclusive, ~h~ their positions in 25 response to the alternation o* the value of the signal between the values ~1 ~nd -1.
Thus it *ollows from the formulae (13) and (15) that the output signal of summing arrangement 21 for ~ = +1 is the same as the output signal of summing arrange-30 ment 23 for ~ = -1 and vice versa.
Likewise it *ollows from the formulae (14) and (16) that -the ou-tput signal o* summing arrangement 22 for ~ = ~1 is tne same as the output signal of summing ar-rangement 24 for ~ = -1 and vice versa.
This results in that the quadra-ture modulators 25 and 26 each produce the two phase-modulated signals Zl(t) and Z2(-t), namely alternately in the rhythm of -the pulse rate of generator 35, with the proviso that if the PHN 9972 12 2l~-8-1981 modulator 2L~ produces -the signal Z1(-t)~ the modulator 2 produces the signal Z2(-t) and vice versa. Owing -to the dynamic modulator adapta-tion obtained in this way -the un-wanted spectra produced by the modulators 2L~ and 25 cancel each other for the most part by means of subtraction in the output stage connec-ted to the modulation stage 1.
Fig. 4 shows an outpu-t stage 2 with power ampli-fication which is advantageous for the electronic arrange-ment for analog signals.
The outpu-t stage 2 comprises two power amplifiers 36 and 37 and a hybrid circuit 38 connected thereto to which on the one hand the output load in the form of an aerial 39 and on -the o-ther hand a ma-tching impedance 4O
equal to the aerial impedance are connec-ted.
Power amplification of the phase-modulated sig-nals Z1(t) and Z2(t) is performed in the amplifiers 36 and 37. Owing to the fact that said signals have a constant amplitude, the non-linear amplification obtained during the amplification (higher harmonics of the carrier fre-20 quency) falls outside the band.
In the hybrid circuit the amplified signals Z1(t) and Z2(t) are assembled to form the amplitude and phase-modulated signal S(t) = r(t).cos( ~ t ~ ~(t)), wherein r(t) is the first modulation signal x1(t) and ~ (t) the second modula-tion signal x2(t). The signal S(t) is applied in i-ts -totality to the aerial 39. When matched properly the hybrîd ensures that the non-linear amplifiers do not see each other. This pre-vents cross~modulation of the two phase-modulated signals.
It is alternatively possible -to use,instead of the arrangements 16 to 24, inclusive so-called "look up tables" which comprise memories in which -the values of the output signals of the adder arrangements 21 to 2LI, inclusive are formed ~or a large number of discrete values 35 of the input signals x1(t) and x2(t). The values of these input signals form the addresses for the output signals which are associated wi-th said input signals, of the adder arrangements 21 to 24, inclusive. The memory arrangemen-t 7S~
further comprises an addressing arrangement which under the control of'-the modulation signals reads the signals corresponding therewi-th and which are proportional to -the phase signals ~ (t) and ~ (t). This may result in a saving in equipment.
It should be noted that in addition to SSB other forms of amplitude and phase-modula-ted signals, such as VSB, may alternatively be generated in the above-described manner. The only differen-t for VSB compared with SSB is -that slightly different fil-ters must be used at the base-band level for the generation of the -two modulation signals required for VSB.
~ s mentioned in the foregoing, the generation o*
an amplitude and phase-modulated signal from two modulation signals by means of two auxiliary signals in the form of phase-modula-ted signals furnishes -the considerable advantage that non-linear distortions in -the power amplifiers have no influence on -the loca-tion of the zero-crossings of the phase-modulated signals. This makes it possible to 20 use any type of amplifier.
If two phase-modulated signals Z1(t) and Z2(t) are passed -through circuits which are commonly referred to as hard limiter circuits then there are obtained from said signals square-wave signals sign Z1(t) and sign Z2(t) 25 which in this example are assembled in the outpu-t stage to form a difference signal sign S'(t) = sign (cos ~ t + ~(t) ~ ~ arccos (r(-t)))) ~
+ sign (cos W ot ~(t) - ~ arccos (r(t)))) (17) in accordance with -the expression r cos(~ ot + ~(t)) = 2 COS (w ot ~ ~(t)) ~ 2 COS (~ t~ X(t)), which has a different construction in order to generate a phase and ampli-tude-modulated signal from two phase-modula-ted signals.
Said signal sign S(t) is a three-level signal~
35 owing to the hard lirnitation of the sub-signals, as is shown in Fig. 5a.
The original amplitude and phase-modulated signal can be recovered therefrom by means of a low-pass filter, 5~3 PHN 9972 1L~ 24-8-1981 which will be demons-trated in the following calculation.
It now holds that:
sign (cos ~ot ~ ~ (t) + 0 arccos r(t))) =
k~co 5 4 ~ ( 1)(k-1)/2 cos k(~ o-t ~ ~ (t) ~ H arccos r(t)) IJ ~ . _ ~
k=1,3,5--- k and (18) sign (cos ~ t ~ y (t) - ~ arccos r(t))) =
= 4 ~ ( 1)(k-1)/2 cos k(~ ot ~ ~'(t)- ~ arccos r(t)) k=1,3,5,....................................... ~19) In the output stage the signal represented by expression (19) is, for example, added to the signal represented by expression ~18') which results in -tha-t ( ) ~ ~ ( 1)(k-1)/2 CS(k arCcOs r(t k=1,3~5~o-cos k ( ~ ot ~ ~(t)) (20) 20 1~riting this sum in individual terms givesS(t) =~ r(-t) cos( ~ot ~ ~(t)) - ~ ~
cos ~( ~ ot ~ ~(t)) + -- (21) Fig. 5b shows the frequency spectrum of this signal.
The use of a low-pass filter then results in that S(t) = r(t) cos( ~ t -~ ~(t)), which signal and asso-cia-ted frequency spectrum are shown in the Figs. 5c and 30 5d.
It is assumed that ~ 0 is sufficiently high to prevent the occurrence of what is commonly referred to as fold-over.
So it is possible to use class "d" amplifiers 35 in cornbination with a low-pass filter in -the aerial lead to provide a very high efficiency.
very ad-vantageous output stage with power amplification is shown in Fig. 6. The square-wave signal PHN 9972 15 Z4~
sign Z1(t) ob-tained by means of hard limiting o~ the signal Z1(t) is-~pplied to a control terminal 41 and -the square-wave signal sign Z2(t) obtained by means of hard limiting of the signal Z2(t) is applied to a control ter-minal 43 o~ change-over switches 42 and 44, respectively.
0~ each change-over switch 42 ancl 44, respectively one o~
the change-over contacts 42-1 and 44~2a respectively :is connected to a first terminal ~E o~ a voltage source having a value of 2E volts and the other change-over con-tact 42-2 and 44-1, respectively is connected to a second terminal -E of the vol-tage source. The swi-tching arms 42-3 and 44-3 o~ the change-over switches 22 and 44 are in-ter-connected via a primary winding 45-1 of an ou-tput trans~
~ormer 45. An aerial may be connected to terminals 46 and 47 o~ a secondary winding 45-2 of the trans~ormer 45.
Said output stage 2 opera-tes as follows.
If a value ~'1" is applied to the control ter-minals 41 and 43 then the two change-over switches 42 and 44 are in the positions not shown and a current ~lows ~rom 20 +E -to -E via contact 42-1, switching arm 42-3, the primary winding 45-1, switching arm 44-3 and contact 44-1. This current induces a voltage in the secondary winding 45-2 which can be taken of~ between the terminals 46 and 47.
I~ the value ~'0" is applied to the two control 25 terminals 41 and 43 then the change-over swi-tches are in the posi-tions shown7 causing a current to ~low from +E
to -E via contact 44-2, switching arm 44-3, the primary winding 45-1, switching arm 42-3 and contact 42-2. Con sequently, the sign o~ the voltage be-tween the -terminals 30 46 and 47 is reversed. I~ the value "1" is applied to one of the con-trol terminals 42 or 43 and the value "0"
to the other terminal or vice versa then the two swit-.. .. . ..
ching arms 42-3 and 44-3 are connected to ei-ther the terminal +E or the terminal -E and there is no voltage 35 across the primar-y winding. The voltage between the ter-minals 46 and 47 is then zero Volts.
Fig. 7 shows a dual outpu-t stage in which a current source 10 is used instead of a voltage source. At -this output 7~3 P~IN 9972 16 24-8-1981 stage one end of -the primary winding of trans~ormer 45 is connected to a-contact of the change-over switch L~2 as well as to a contac-t o~ the change-over swi-tch 44, the other end of the primary winding is connected to the other contact of the change-over switches 42 and 44 and the current source 10 is connected be-tween the switching arms of the change-over switches 42 and 44. This ou-tput stage operates in a similar manner as -the embodiment shown in Fig. 6, with the proviso that the current flowing through the primary winding of the transformer 45 is produced by the current source 10 instead of by the voltage source.
An advantageous embodiment of the change-over switches 42 and 44 of the output stages shown in the Figs. 6 and 7 is realized with the aid of, for example, MES or MOSFETS.
Fig. 8 shows an other output stage realized by means of current sources. The hard limited phase-modulated signal sign Z1(t) and sign Z2~t) applied by -the modulation stage 1 to the output stage are applied to three logic gate circuits, namely a first AND-gate circuit 48y an 20 exclusive-NOR-gate circuit 49 and a second AND-gate cir-cuit 50 having two inverting inpu-ts, which generate the respective signals sign Z1(t).sign Z2(t); the signal 1 g 2(t) ~ sign Z1(t).sign Z2(t) and the signal sign Z1(b). ~ -These signals are applied to a three-stage flip-flop circuit 60 as set signals on the positive edges~
which circuit comprises current sources 61 and 62 con nected to outputs 60-1 and 60-2. These curren-t sources 61 and 62 are connected to an aerial 63. Under the control 30 Of the output signal produced by the logic circui-t 48 the current sources 61 and 62 are s~itched on on a posi-tive edge o~ said ou-tput signal b~- the -three-s-tate flip-~lop circuit 60, which current sources then apply a current having a value 2 I to the aerial 63.
Under the control of the ou-tpu-t signal produced by the logic circuit 49 only current source 61 and 62 is switched on on a positive edge of said output signal by the three-state flip-flop circuit 60 in response to which 75~
only a current having a value I is applied to the aerial 63. Finally under the control of the output signal produced by the logic circui-t 50 the -two curren-t sources 61 and 62 are switched off on a positive edge of said signal by the three-state flip-flop circuit 60~ so that no current is applied to the aerial.
The aerial cable itself may have a low-pass characteristic so that in that even-t a discrete low-pass filter is no-t required~ as is shown in the Figs. 6, 7 and 8.
For -those outpu-t s-tages 2, which are controlled by phase-modulated signals having discrete values Figs.
10 and 11 show very advantageous modulation stages 1. For a more detailed description of these modulation stages it will first be demonstrated with reference to Fig. 9 that there is a relationship between the product of the hard limited phase-modulated signals sign Z1(t) and sign Z2(t) and the product of two pulse-duration modulated signals P1(t) and P2(t) with natural sampling.
The first hard limited phase-modulated signal is sign Z1(t) = sign (cos ~ot ~ (t) ~ 0 arccos r(t))) If, for example 0 is chosen equal to +1 then it may be written that:
sign Z1(t) = sign (cos ~ot ~ ~ (t~ ~ arcsin r(t) + 2 )) 25 = -sign (sin ~ t ~ ~ (t) ~ arcsin r(-t))) which signal is shown in Fig. 9d.
The second hard limited phase-modulated signal is sign Z2(t? = sign (cos ~Ot + ~ (-t) - 0 arccos r(t))) 30 = sign (C09 C~o-t ~ y(t) - arcsin r(t) ~ ~2 )) = sign (sin w t + y(t) - arcsin r(t))) which signal is shown in Fig. 9e.
The product of -the hard limited phase-modulated signals is consequently equal to:
-sign (sin( ~ Ot~ ~(t)~ arcsin r(t))).sign(sin(~ t~ ~(t) --arcsin r(t))) This may be written as:
1~387~
PIIN 9972 18 2L~-8-1981 -sign(sirl( ~ t+ ~(t)~arcsin r(t)).Sill(~ ot~ ~(t)-arcsin r(t))) which correspon~s to:
-sign(cos(2 arcsin r(t))-cos 2(~ t ~ y (t))) =
-sign(1-2 sin2(arcsin2r(t))-1+2 sin2(~ t ~ ~(t)) =
5 -sign(sin2(~ t~ Y(t)) - r2(t)) (22) Expression (22) may be written as:
sign(r(t)+sin( ~ t~ ~(t))).sign(r(t)-si~ t~ ~(t))) with which it is proved that sign (z1(t)).sign (Z2(t)) = P1(t) ' 2( ) (23) wherein Pl(t) = sign (r(t) ~ sin(~Ot + ~ (t))) and P2(t) = sign (r(t) - sin(W -t ~-y (t))) i5 .
From (23) it follows that -the zero-crossings of the product of the hard limited phase-modulated signals sign Z1(t) and sign Z2(t) are identical to -the zero-crossings of the 15 product of the pulse-width modulated P1(t).P2(t).
In Fig. 9a the function sin( ~ t +~(t)) (solid line) the function -~r(t) (dot dash line) and the function -r(t) (dashed line) are shown.
The sign of the function r(t)~sin( ~ t + ~(t)) 20 which represents -the pulse-modulated signal P1(t) is shown in Fig. 9b. The values for which sin( ~ t ~ y (t)) is equal to -r(t) form the transitions of the pulse-modulated signal P1(t) which points of intersection are designated by "b" in Fig. 9a. Likewise the poin-ts where the signal 25 +sin( ~ ot ~ ~'(-t)) intersects the signal +r(t) form the transitions of the pulse-modula-ted signal P2(t), -which signal P2(t) is shown in Fig. 9c, use being made of -the fact that r(t)-sin( ~ Ot~ ~ ~t)) = -(sin( ~ Ot ~ y (t)) - r(t)) 30 These points of intersection are designated "c" in Fig~ 9a.
As has been demonstrated in the foregoing~ the edges of these signals coincide with the edges of the pulse-modulated signals Pl(-t) and P2(t) 9 it appears how-ever that the ascending edges of the signal sign Z1(-t) 35 coincide with -the ascending edges of the signal P2(t) and the descending edges of -the signal Z1(-t) coincide with the ascending edges of the signal P1(t), which in Fig. 9a is shown by providing the relevant poin-ts of intersection 5~3 P~IN 9972 19 2ll-8-1981 wi-th a second designation "d1'. I-t likewise appears that the ascending edges of -the signal sign Z2(t) coincide with the descending edges of the signal P2(t) and the descending edges of the signal sign Z1(t) coincide with the descending edges of the signal P1(t)~ which is shown in Fig. 9a by providing the relevant points of in-tersection with a second designation "e".
The embodiments shown in Figs. 10 and 11 of` a modulation stage 1 utilize the above-mentioned property, Thus, Fig. 10 shows a modulate stage 1, in which a modula-tion signal ~(-t) applied to an input terminal 4 first se .~ modulates in a p~ modulator 64 -the phase of a carrier signal produced by a carrier generator 65. The signal sin( ~ t + ~(-t)) thus obtained is applied, together with a modula-tion signal r(t) applied to the input terminal 3, to terminals 66-2 and 66-1 of a signal converter arrange-ment 66. In arrangement 66 these signals are applied to signal inputs 67-1 and 67-2 of a first comparator circuit 67 and also to signal inputs 68-1 and 68 2 of a second comparator circuit 68. These comparator circuits 67 and 68 differ only in that the signal input 68-2 of circuit 68 is a signal-inverting input while -the corresponding input 61-2 of circuit 67 is a signal non-inverting input.
The output signals of -the comparator circuits 67 and 68 are binary so that comparator circuit 67 produces the pulse-modulated signal P1(t), shown in Fig. 9b and com-parator circuit 68 produces -the pulse modulated sigrnal P2(t), shown in Fig. 9c. The signal P1(t) is applied -to a divide-by-two circuit 69 and the signal P2(t) to a divide-30 by-two circuit 70. The output signal at the signal output a of the divide-by--two circuit 69 changes its value at every ascending edge of signal P1(t) as is shown in Fig.
9f and -the ou-tpu-t signal from the output b of said divide-by-two eireuit 69 changes its value at every deseending edge of signal P1(t) as shown in Fig. 9g.
Likewise the signal on signal output a of the divide-by-two eireuit 70 ehanges at every aseending edge of signal P2(t) as shown in Fig. 9h and the signal on signal output b of the divide-by-two circuit 70 changes at every desc-encl-ing edge of signal P2(t) as shown in Fig. 9i. The signal supplied from outpu-t a of the divide-by--two circui-t 69 and the signal supplied from outpu-t a of the divide-by-two circui-t 70 are applied to an exclu~
sive-OR~circuit which forms from the signal sign Z1(t) shown in Fig. 9d, which can be taken off -from output 5.
Likewise the signal f`rom output b of the divide-by-two circuit 70 and the signal from output b of the divide-by-two circuit 69 are applied to an exclusive /'OR"-circuit 72, which forms from these signals -the signal sign Z2(t) shown in Fig. 9e, which can be taken from output 6.
Owing to the fact that -the initial state of the divide-by-lS two circuits is not defined, said signal converter ar-rangement 66 comprising divide-by-two circuits 6~ and 70 creates an uncer-tain-ty as regards the sign of the ou-tput signal sign S(t) obtained by summation in the output s-tage
2.
To avoid this, Fig. 11 shows a differen-t embodi-ment of a signal converter 66 for use in a modulation stage 1 as shown in Fig. 9. This arrangement 66 also em-ploys pulse-duration modulated signals for generating the hard limited phase-modulated signals, but differs in tha-t 25 it comprises a logic circuit which only responds to posi-tive signal changes.
The signals r(t) and sin( wot ~y(t)) applies to the input -terminals 66-1 and 66-2 are applied to two further comparator arrangemen-ts 73 and 74, respec-tively, 30 which only differ from the compara-tor arrangements 68 and 67, respectively, shown in Fig. 10~ in that in addition to signal outputs 73-1 and 7L'-1 they also have signal inverting outputs 73-2 and 74-2.
Conse~uently, the outpu-t signal of output 73-1 35 is equal to -the signal P2(-t) shown in Fig. 9c and -the output signal at output 73-2 is the inverse of said signal.
Likewise the output signal at output 74-1 is the signal Pl~t) shown in Fig. 9b and the output signal at ~8~S~
P~IN 9972 21 24~8~'l981 output 74-2 is the inverse of said signal.
A set-reset flip-flop circuit 51 whic'h is onl~
responsive to positive edges is provided between the out-puts 73-1 and 74-2.
The operation will be described in greater de-tail with reference to Figs. 9b to 9e, inclusive.
At the instant t2 there occurs in the output signal 9c at output 73-1 a positive edge which se-ts the flip-~lop circuit 51 to the initial state and the signal output q delivers a high signal (Fig. 9d) from output 5.
At the instant tL~ there occurs in the output signal 9b a-t output 74-1 a positive edge which resets the flip-flop circuit 51 and the signal output q supplies a low signal (Fig. 9d), etc. Consequently, the hard-limited first phase-modulated signal sign Z1(-t) as shown in Fig. 9d appears at output terminal 5.
A flip-flop circuit 52 is provided between the outpu-ts 73-2 and 74-2.
In a similar manner as was demons-trated for flip-flop circuit 51 it can be demonstrated from the inverse version of the signals shown in the Figs. gb and 9c~ which are supplied by the outputs 73-2 and 74-2 that the signal output q of said f]ip-flop circuit 52 applies -the hard-limited second phase-modulated signal sign Z2(t) to out-25 put terminal 6.
It is further obvious tha-t if the flip-flop circuit 52 is only responsive to negative edges -the input of said ~lip-flop circuit must then also be connected be-tween the outpu-t terminal 73-1 and 74-1 in order to obtain 30 the signal Z2(t) at output -terminal 6, e-tc.
Fig. 12 shows an embodiment which is suitable for the phase modulator 64 of Fig, 10.
The modulation signal ~(t) applied to the input terminal 4 is applied to a phase-folding arrangement 77, 35 which converts the signal ~ (t) into a signal sin 0 (t) in a manner still to be described. Said signal sin 0(t) is applied to a multiplier 78 where the signal is multi-plied br a signal sign (cos ~ t), wherein ~ represents
To avoid this, Fig. 11 shows a differen-t embodi-ment of a signal converter 66 for use in a modulation stage 1 as shown in Fig. 9. This arrangement 66 also em-ploys pulse-duration modulated signals for generating the hard limited phase-modulated signals, but differs in tha-t 25 it comprises a logic circuit which only responds to posi-tive signal changes.
The signals r(t) and sin( wot ~y(t)) applies to the input -terminals 66-1 and 66-2 are applied to two further comparator arrangemen-ts 73 and 74, respec-tively, 30 which only differ from the compara-tor arrangements 68 and 67, respectively, shown in Fig. 10~ in that in addition to signal outputs 73-1 and 7L'-1 they also have signal inverting outputs 73-2 and 74-2.
Conse~uently, the outpu-t signal of output 73-1 35 is equal to -the signal P2(-t) shown in Fig. 9c and -the output signal at output 73-2 is the inverse of said signal.
Likewise the output signal at output 74-1 is the signal Pl~t) shown in Fig. 9b and the output signal at ~8~S~
P~IN 9972 21 24~8~'l981 output 74-2 is the inverse of said signal.
A set-reset flip-flop circuit 51 whic'h is onl~
responsive to positive edges is provided between the out-puts 73-1 and 74-2.
The operation will be described in greater de-tail with reference to Figs. 9b to 9e, inclusive.
At the instant t2 there occurs in the output signal 9c at output 73-1 a positive edge which se-ts the flip-~lop circuit 51 to the initial state and the signal output q delivers a high signal (Fig. 9d) from output 5.
At the instant tL~ there occurs in the output signal 9b a-t output 74-1 a positive edge which resets the flip-flop circuit 51 and the signal output q supplies a low signal (Fig. 9d), etc. Consequently, the hard-limited first phase-modulated signal sign Z1(-t) as shown in Fig. 9d appears at output terminal 5.
A flip-flop circuit 52 is provided between the outpu-ts 73-2 and 74-2.
In a similar manner as was demons-trated for flip-flop circuit 51 it can be demonstrated from the inverse version of the signals shown in the Figs. gb and 9c~ which are supplied by the outputs 73-2 and 74-2 that the signal output q of said f]ip-flop circuit 52 applies -the hard-limited second phase-modulated signal sign Z2(t) to out-25 put terminal 6.
It is further obvious tha-t if the flip-flop circuit 52 is only responsive to negative edges -the input of said ~lip-flop circuit must then also be connected be-tween the outpu-t terminal 73-1 and 74-1 in order to obtain 30 the signal Z2(t) at output -terminal 6, e-tc.
Fig. 12 shows an embodiment which is suitable for the phase modulator 64 of Fig, 10.
The modulation signal ~(t) applied to the input terminal 4 is applied to a phase-folding arrangement 77, 35 which converts the signal ~ (t) into a signal sin 0 (t) in a manner still to be described. Said signal sin 0(t) is applied to a multiplier 78 where the signal is multi-plied br a signal sign (cos ~ t), wherein ~ represents
3'7S~
PHN 9972 22 2l~-8-1981 the angular frequency of the carrier signal. Said signal sign cos ( ~ t~ is supplied by an output 65-2 of the car-rier generator 65. For that purpose a hard-Limiting cir-cuit, not shol~l, is provided in the output circuit of -the carrier generator. In the mul-tiplier 78 the sign of -the signal sin 0 (t) is switched between plus and minus by the signal sign cos (~ t) at the rate of -the carrier frequency. The output signal sign (cos ~ t).sin 0(t) of the mul-tiplier 78 is applied to an input 79-2 of a further comparator circuit 79 where a carrier signal sin t supplied by the output 65-1 of the carrier generator 65 is applied to an input 79-1. Said additional comparator circuit 79 generates from these inpu-t signals the signal sign(sin ~ ot ~ sign (cos ~ot) sin 0 (t)) which signal is iden-tical to the signal sign sin( ~ t + ~(t)) as will be demonstrated hereinafter.
For the zero-crossings of the signal produced by the comparator circuit 79 i-t holds that sin ~o(t) = sign (cos~ ot) sin 0 (-t) For consecutive zero-crossings i-t therefore applies that sin ~ot = - sin 0 (t) sin wot = ~ sin 0 (t) sin ~Ot = - sin 0 (t) etc.
25 ~his means ~hat:
~ t ~ 0 (t) = K'~ wherein K = (...,0,1,2,3,...) This rela-tion represents the zero-crossings of the signal sign (sin ~ -t ~ 0(t)), so that the output signal of the further comparator circuit 79 is equal to sign sin( ~ ot +
30 0 (t)). Now the signal sin ( ~ t / 0 (t)) is identical to t'he signal sin ( ~ t ~ y(t)) if 0(t) is chosen so that i-t holds that -~ /2 ~ O(t) C ~ ~ /2, so is limited, while ~ (t) is unlimited. The phase-folding arrangement 77 must consequently ensure that the signal ~ (t) in the signal 35 O(t) is converted, 0(t) within the above-mentioned value 'being limited in order to apply a signal having a finite value to the multiplier 78, for example at a continuously increasing value of the signal ~(t). For that purpose the ~3875~3 PHN ~972 23 2L~-8-1981 phase-folding arrangement has, for example, a charac-teris-tic as shown in Fig. 13.
This Fig. 13 shows that associa-ted with each value of the inpu-t signal ~ ~t) there is a predetermined value of the output signal 0(-t) located within the limits - ~/2 and ~ /2. Associated with the sawtooth-shaped characteris-tic, shown in Fig. 13, for the conversion the signal ~(t) into ~(t), which contains the signal sin ~(t) as -the fundamental wave -there is a likewise sawtooth-shaped signal as the carrier signal. This is illustrated in Fig. 12 for -the output signal from the output 65-1 of the carrier generator 65. It is, however, possible to use sinusoidal signals or other approximate signals instead oP the sinusoidal signals represented by the sawtooth-shaped signals.
From the foregoing it has been found that the relation-ship be-tween the signals ~ (t) and ~(t~ is unambiguous.
Said folding arrangement may consequently be represented by means of a table which is commonly re~erred to as a "look-up table", which comprises a memory in which for each value of the input signal ~(t) the associated value of the output signal ~(t) is stored and for which the input signal ~ (t) is employed as the address signal of the associa-ted signal ~(t).
Via a low-pass fil-ter 80 the output signal of the further comparator circuitis applied to a terminal 62-2 where the phase-modulated carrier signal sin ( ~ t -~
(t)) becomes available for fur-ther processing.
The embodiments of the modulation stage 1 as 30 shown in the Figs. 10 and 11 have the advantage -that they may be realize in in-tegrated form and may be connected to, for example, the output stage 2 shown in Figs. 6, 7 or 80 The electronic arrangement for generating, for example, a high power single-sideband signal is then in 35 the form of an integrated modulation stage 1 connected to an output stage 2 forrned by change-over swi-tches in the form of~ for example, ~IESFET--transistors, and an outpu-t transformer.
1~8751~
PEIN 9972 2l~ 24-8-1981 From -the f'oregoing description it will be obvious -that any type-of-signalg such as for example any -two data signals may be chosen as the modulation signals x1(t) and ~2(t). The electronic arrangement is particularly suitable to modulate one single incoming bit stream of a data signal, coded in accordance with the four-phase method with reduced bandwidth~ on a carrier sig~al, it being possible to use the x and y components of the phase poin-ts in the phase plane as the modulation signals xl(-t) and 1D x2(t) 7 or the x componen-t ,~nd the y component shifted over half a period of the phase points. These last-mentioned modulation signals then result in a OQPSI~ signal.
PHN 9972 22 2l~-8-1981 the angular frequency of the carrier signal. Said signal sign cos ( ~ t~ is supplied by an output 65-2 of the car-rier generator 65. For that purpose a hard-Limiting cir-cuit, not shol~l, is provided in the output circuit of -the carrier generator. In the mul-tiplier 78 the sign of -the signal sin 0 (t) is switched between plus and minus by the signal sign cos (~ t) at the rate of -the carrier frequency. The output signal sign (cos ~ t).sin 0(t) of the mul-tiplier 78 is applied to an input 79-2 of a further comparator circuit 79 where a carrier signal sin t supplied by the output 65-1 of the carrier generator 65 is applied to an input 79-1. Said additional comparator circuit 79 generates from these inpu-t signals the signal sign(sin ~ ot ~ sign (cos ~ot) sin 0 (t)) which signal is iden-tical to the signal sign sin( ~ t + ~(t)) as will be demonstrated hereinafter.
For the zero-crossings of the signal produced by the comparator circuit 79 i-t holds that sin ~o(t) = sign (cos~ ot) sin 0 (-t) For consecutive zero-crossings i-t therefore applies that sin ~ot = - sin 0 (t) sin wot = ~ sin 0 (t) sin ~Ot = - sin 0 (t) etc.
25 ~his means ~hat:
~ t ~ 0 (t) = K'~ wherein K = (...,0,1,2,3,...) This rela-tion represents the zero-crossings of the signal sign (sin ~ -t ~ 0(t)), so that the output signal of the further comparator circuit 79 is equal to sign sin( ~ ot +
30 0 (t)). Now the signal sin ( ~ t / 0 (t)) is identical to t'he signal sin ( ~ t ~ y(t)) if 0(t) is chosen so that i-t holds that -~ /2 ~ O(t) C ~ ~ /2, so is limited, while ~ (t) is unlimited. The phase-folding arrangement 77 must consequently ensure that the signal ~ (t) in the signal 35 O(t) is converted, 0(t) within the above-mentioned value 'being limited in order to apply a signal having a finite value to the multiplier 78, for example at a continuously increasing value of the signal ~(t). For that purpose the ~3875~3 PHN ~972 23 2L~-8-1981 phase-folding arrangement has, for example, a charac-teris-tic as shown in Fig. 13.
This Fig. 13 shows that associa-ted with each value of the inpu-t signal ~ ~t) there is a predetermined value of the output signal 0(-t) located within the limits - ~/2 and ~ /2. Associated with the sawtooth-shaped characteris-tic, shown in Fig. 13, for the conversion the signal ~(t) into ~(t), which contains the signal sin ~(t) as -the fundamental wave -there is a likewise sawtooth-shaped signal as the carrier signal. This is illustrated in Fig. 12 for -the output signal from the output 65-1 of the carrier generator 65. It is, however, possible to use sinusoidal signals or other approximate signals instead oP the sinusoidal signals represented by the sawtooth-shaped signals.
From the foregoing it has been found that the relation-ship be-tween the signals ~ (t) and ~(t~ is unambiguous.
Said folding arrangement may consequently be represented by means of a table which is commonly re~erred to as a "look-up table", which comprises a memory in which for each value of the input signal ~(t) the associated value of the output signal ~(t) is stored and for which the input signal ~ (t) is employed as the address signal of the associa-ted signal ~(t).
Via a low-pass fil-ter 80 the output signal of the further comparator circuitis applied to a terminal 62-2 where the phase-modulated carrier signal sin ( ~ t -~
(t)) becomes available for fur-ther processing.
The embodiments of the modulation stage 1 as 30 shown in the Figs. 10 and 11 have the advantage -that they may be realize in in-tegrated form and may be connected to, for example, the output stage 2 shown in Figs. 6, 7 or 80 The electronic arrangement for generating, for example, a high power single-sideband signal is then in 35 the form of an integrated modulation stage 1 connected to an output stage 2 forrned by change-over swi-tches in the form of~ for example, ~IESFET--transistors, and an outpu-t transformer.
1~8751~
PEIN 9972 2l~ 24-8-1981 From -the f'oregoing description it will be obvious -that any type-of-signalg such as for example any -two data signals may be chosen as the modulation signals x1(t) and ~2(t). The electronic arrangement is particularly suitable to modulate one single incoming bit stream of a data signal, coded in accordance with the four-phase method with reduced bandwidth~ on a carrier sig~al, it being possible to use the x and y components of the phase poin-ts in the phase plane as the modulation signals xl(-t) and 1D x2(t) 7 or the x componen-t ,~nd the y component shifted over half a period of the phase points. These last-mentioned modulation signals then result in a OQPSI~ signal.
Claims (16)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. An electronic arrangement for generating an am-plitude and phase-modulated carrier signal, the arrange-ment comprising a phase modulation stage having two inputs for receiving two modulation signals, which modulation stage under the control of the modulation signals generates in use two different phase-modulated signals having the same carrier frequency and substantially the same amplitude and an output stage to which the phase modulated signals are applied for assembling by means of summation of the phase-modulated signals, the amplitude and phase-modulated carrier signal whose amplitude is modulated in dependence on the phase difference of the phase-modulated signals and whose phase is modulated in dependence on the sum of the phases of the phase-modulated signals.
2. An electronic arrangement as claimed in Claim 1, wherein the phase modulation stage converts a first modu-lation signal x1(t) and the second modulation signal x2(t) into two phase-modulated signals Z1(t) = a cos(.omega.ot +? (t)) and Z2(t) = a cos (.omega.ot + ? (t)), the phase ? (t) of the first phase-modulated signal Z1(t) being equal to ?(t) + .THETA. arccos (r(t)/2a) + m . 2 .pi. and the phase ?(t) of the second phase-modulated signal Z2(t) being equal to ?(t) - .THETA. arccos (r(t)/2a) + n.2 .pi. + wherein ? = ?1, it holding that ? = + 1 when adding Z1(t) + Z2(t) to form the summation signal S(t) and ? = -1 when subtracting Z2(t) from Z1(t) to form the difference signal S'(t), and the modulation signals x1(t) and x2(t) representing the coordinates of a vector in a system of coordinates, which vector, when transformed in a system of polar coordinates, is represented by the quantities r(t) and ?(t), wherein .THETA. = ? 1, m and n being integers and the amplitude and phase-modulated signal S(t) and S'(t) respectively being represented by r(t)cos(.omega.ot + ? (t)).
3. An electronic arrangement as claimed in Claim 2, wherein the phase modulation stage comprises a signal transforming arrangement and a phase modulation arrange-ment connected thereto, the signal transforming arrange-ment is arranged for generating from the modulation signals x1(t) and x2(t) signals which are proportional to the phases ? (t) and ? (t) and the phase modulation stage com-prises phase modulators for generating phase-modulated signals Z1(t) and Z2(t) from the signals which are pro-portional to the phases ? (t) and ? (t).
4. An electronic arrangement as claimed in Claim 3, wherein the signal transforming arrangement comprises an arccosine generator for generating the signal .THETA. arccos (r(t)/2a) from the modulation signals, and second and third summing arrangements to which the output signal of the arccosine generator and the signal ? (t) are applied for assembling in the second summing arrangement the phase signal ?(t) = ?(t) + .THETA. arccos (r(t)/2a) by means of addition and for assembling the phase signal ?(t) = ?(t) -.THETA. arccos r(t)/2a in the third summing arrangement by means of subtraction.
5. An electronic arrangement as claimed in Claim 2, wherein the signal transforming arrangement of the phase modulation stage comprises a function generator connected to the input terminals of the phase modulation stage for generating the signal E(t) = 1/r(t) from the modulation signals x1(t) and x2(t), two multiplying arrange-ments a first one of which is connected to one of the input terminals and a second one to the second input terminal and which are both connected to the function generator, wherein a pulse signal generator connected to the two multiplying arrangements is provided, said generator having a pulse repetition rate which is at least twice as high as the highest signal frequency of the modulation signals, said pulse signal representing a signal .THETA., for forming in the first multiplying arrangement the product signal .THETA..x1(t) . E (t) and for forming in the second multiplying arrangement the product signal .THETA..x2(t) . E(t), a fourth fifth, sixth and seventh summing arrangement, the fourth arrangement being connected to the first input terminal and to the first multiplying arrangement for forming the difference signal ?(x1(t) - .THETA.x2(t).E(t)) = cos?(t), the fifth being connected to the second input terminal and the second multiplying arrangement for forming the sum signal ?(.THETA.x1(t).E(t)+x2(t)) = sin?(t), the sixth being connected to the first input terminal and to the second multiplying arrangement for forming the sum signal ?(x1(t) + .THETA.x2(t).E(t) = cos?(t), and the seventh multi-plying arrangement being connected to the second input terminal and to the second multiplying arrangement for forming the difference signal ?(-.THETA.x1(t).E(t) + x2(t)) =
sin X (t), and wherein the modulation arrangement com-prises two quadrature modulators a first of which is con-nected to the fourth and the fifth summing arrangement for forming the signal ? cos (.omega.ot + ?(t)) = Z1(t) and a second is connected to the sixth and the seventh summing arrangement for forming the signal ? cos(.omega.ot + ?(t)) =
Z2(t).
sin X (t), and wherein the modulation arrangement com-prises two quadrature modulators a first of which is con-nected to the fourth and the fifth summing arrangement for forming the signal ? cos (.omega.ot + ?(t)) = Z1(t) and a second is connected to the sixth and the seventh summing arrangement for forming the signal ? cos(.omega.ot + ?(t)) =
Z2(t).
6. An electronic arrangement as claimed in Claim 3 wherein the signal transforming arrangement comprises a memory to which samples of the modulation signals x1(t) and x2(t) are applied as addressing signals and each memory address there are stored the output signals which are proportional to the phase signals ? (t) and ? (t) and associated with each combination of the signals x1(t) and x2(t) and wherein the memory arrangement comprises an addressing arrangement for reading out under the control of the address signal the output signals associated with each address.
7. An electronic arrangement as claimed in Claim 2, wherein the phase modulation stage comprises a phase modulator connected to one of the input terminals and hav-ing a carrier generator connected thereto for modulating the phase of the carrier signal with the modulation signal present on said input terminal, the phase modulation stage comprises a first comparator circuit having two signal inputs a first input of which is connected to the phase modulator and the second input to the other input terminal for generating from the phase-modulated signal and the other modulation signal a first pulse-duration modulated signal with natural sampling, a second comparator circuit having a signal input and a signal inverting input, the signal inverting input of which is connected to the phase modulator and the signal input to the said other input terminal for generating from the phase-modulated signal and the said other modulation signal a second pulse-duration modulated signal with natural sampling and connected to the output of the comparator circuit, a logic signal converter arrangement for generating the two phase-modulated signals Z1(t) and Z2(t) in the hard-limited form from the pulse-duration modulated signals.
8. An electronic arrangement as claimed in Claim 7, wherein the logic signal converting arrangement comprises a divide-by-two circuit connected to each output of the first and second comparator circuits and each divide-by-two circuit having two outputs for producing at a first output one of the two possible divided signals obtained by divi-sion by two of the pulse-duration modulated signals ap-plied thereto and for producing at the second output the other divided signal obtained by means of division by two, a first exclusive-"OR"-gate circuit connected to the first output of a first of the two divider circuits and to the first output of the second divide-by-two circuit and a second exclusive "OR"-gate circuit connected to the second output of the second divide-by-two circuit and to the second output of the first divide-by-two circuit, and wherein the outputs of the exclusive OR-gate circuits are connected to the outputs of the modulation stage.
9. An electronic arrangement as claimed in Claim 7, wherein the comparator circuits each have a signal output and an inversed signal output, the logic signal arrange-ment comprises two flip-flop circuits each being only responsive to edges of one polarity, and wherein the inputs of one of the two flip-flop circuits are connected between corresponding outputs of the two comparator circuits and the inputs of the other of the two flip-flop circuits is also connected between the corresponding outputs of the two comparator circuits the outputs of the flip-flop circuits being connected to the output terminals of the modulation stage.
10. An electronic arrangement as claimed in Claim 1, wherein the output stage comprises an amplifying summing circuit for amplifying and summing the phase-modulated signals Z1(t) and Z2(t) applied to the amplifier stage and for assembling said signals to form the amplitude and phase-modulated output signal.
11. An electronic arrangement as claimed in Claim 10, wherein the amplifying summing arrangement comprises two amplifiers for individually amplifying the phase-modulated signals Z1(t) and Z2(t), a hybrid circuit is connected to the outputs of the amplifier for assembling the phase-modulated signal supplied by the amplifier to form the amplitude and phase-modulated signal and wherein the output of the hybrid circuit is connected to the output of the output stage.
12. An electronic arrangement as claimed in Claim 10, wherein the amplifying summing circuit comprises a first logic "AND"-gate circuit and a second logic "AND"-gate circuit having two signal-inverting inputs, and a logic exclusive-NOR-gate circuit, one of the signal inputs of the first AND-gate circuit, one of the inputs of the exclusive NOR-gate circuit and one of the inverting signal inputs of the second AND-gate circuit being connected to one input of the output stage, the other signal input of the first AND-gate circuit, the other signal input of the exclusive NOR-gate circuit and the other signal-inverting input of the second AND-gate circuit being connected to the other input of the output stage, a three-state trigger circuit connected to the logic gate circuits and two current sources having control inputs which are connected to respective signal outputs of the three-state trigger circuit, the out-puts of the current sources being connected to the output of the output stage.
13. An electronic arrangement as claimed in Claim 10, wherein the amplifying summing arrangement comprises a transformer and two change-over switches having control inputs, the control inputs of the change-over switches are each connected to one associated input of the two inputs of the output stage, one of the contacts of the two change-over switches is connected to a first terminal of a voltage source, the other contact of the two change-over switches is connected to a second terminal of the voltage source with a voltage different from the first terminal, the switching arms of the change-over switches are intercon-nected via a primary winding of the transformer and wherein a secondary winding of the transformer is connected between the output terminals of the output of the output stage.
14. An electronic arrangement as claimed in Claim 10, wherein the amplifying summing arrangement comprises a transformer, a first current source and two change-over switches having control inputs, one of the contacts of the two change-over switches is connected to one end of the primary winding of the transformer, the other contact of the two change-over switches is connected to the other end of the primary winding of the transformer, the current source is connected between the switching arms of the change-over switches and wherein the control inputs of the change-over switches are each connected to an associated input of the two inputs of the output stage.
15. An electronic circuit as claimed in Claim 13 or 14, wherein each change-over switch is realized by means of two FET-transistors.
16. An electronic circuit as claimed in Claim 7, wherein the phase modulator comprises a phase-folding arrangement connected to an input terminal for limiting the modulation signal present on said input terminal within the limits - .pi./2 and + .pi./2 the carrier generator has two out-puts for supplying two carrier signals whose phases are shifted 90° relative to each other, a further multiplier is provided which is connected to the phase-folding circuit and via a hard-limiting circuit to one of the outputs of the carrier generator, a comparator circuit is provided which is connected to the multiplier and to the other out-put of the carrier generator and wherein a low-pass filter is provided which is connected to the output of the further comparator circuit.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| NL8101109 | 1981-03-09 | ||
| NL8101109A NL8101109A (en) | 1981-03-09 | 1981-03-09 | ELECTRONIC DEVICE FOR GENERATING AN AMPLITUDE AND PHASE MODULATED CARRIER SIGNAL. |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1188758A true CA1188758A (en) | 1985-06-11 |
Family
ID=19837123
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000397592A Expired CA1188758A (en) | 1981-03-09 | 1982-03-04 | Electronic arrangement for generating an amplitude and phase-modulated carrier signal |
Country Status (8)
| Country | Link |
|---|---|
| US (1) | US4485357A (en) |
| JP (1) | JPS57159156A (en) |
| CA (1) | CA1188758A (en) |
| DE (1) | DE3207786A1 (en) |
| FR (1) | FR2501441A1 (en) |
| GB (1) | GB2095492B (en) |
| IT (1) | IT1198356B (en) |
| NL (1) | NL8101109A (en) |
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| US8315336B2 (en) | 2007-05-18 | 2012-11-20 | Parkervision, Inc. | Systems and methods of RF power transmission, modulation, and amplification, including a switching stage embodiment |
| US7620129B2 (en) | 2007-01-16 | 2009-11-17 | Parkervision, Inc. | RF power transmission, modulation, and amplification, including embodiments for generating vector modulation control signals |
| WO2008156800A1 (en) | 2007-06-19 | 2008-12-24 | Parkervision, Inc. | Combiner-less multiple input single output (miso) amplification with blended control |
| WO2009145887A1 (en) | 2008-05-27 | 2009-12-03 | Parkervision, Inc. | Systems and methods of rf power transmission, modulation, and amplification |
| WO2012139126A1 (en) | 2011-04-08 | 2012-10-11 | Parkervision, Inc. | Systems and methods of rf power transmission, modulation, and amplification |
| WO2012167111A2 (en) | 2011-06-02 | 2012-12-06 | Parkervision, Inc. | Antenna control |
| US20130082756A1 (en) * | 2011-10-04 | 2013-04-04 | Electronics And Telecommunications Research Institute | Signal input device of digital-rf converter |
| KR20160058855A (en) | 2013-09-17 | 2016-05-25 | 파커비전, 인크. | Method, apparatus and system for rendering an information bearing function of time |
| RU169213U1 (en) * | 2016-06-14 | 2017-03-09 | Акционерное общество "Научно-исследовательский институт Приборостроения имени В.В. Тихомирова" | MULTI-CHANNEL COMPOSER OF MULTI-FREQUENCY SIGNALS |
Family Cites Families (5)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US3778718A (en) * | 1972-04-28 | 1973-12-11 | Avco Corp | Modulation system |
| US3896395A (en) * | 1974-07-18 | 1975-07-22 | Bell Telephone Labor Inc | Linear amplification using quantized envelope components to phase reverse modulate quadrature reference signals |
| US4006418A (en) * | 1975-05-14 | 1977-02-01 | Raytheon Company | Quaternary phase-shift keying with time delayed channel |
| US4079204A (en) * | 1975-12-26 | 1978-03-14 | Sansui Electric Co., Ltd. | AM Stereophonic transmission system |
| NL8001903A (en) * | 1980-04-01 | 1981-11-02 | Philips Nv | DEVICE FOR AMPLIFYING A MODULATED CARRIER SIGNAL. |
-
1981
- 1981-03-09 NL NL8101109A patent/NL8101109A/en not_active Application Discontinuation
-
1982
- 1982-02-08 US US06/346,703 patent/US4485357A/en not_active Expired - Lifetime
- 1982-03-04 CA CA000397592A patent/CA1188758A/en not_active Expired
- 1982-03-04 DE DE19823207786 patent/DE3207786A1/en active Granted
- 1982-03-05 GB GB8206494A patent/GB2095492B/en not_active Expired
- 1982-03-05 FR FR8203717A patent/FR2501441A1/en active Granted
- 1982-03-05 IT IT20004/82A patent/IT1198356B/en active
- 1982-03-09 JP JP57035948A patent/JPS57159156A/en active Granted
Also Published As
| Publication number | Publication date |
|---|---|
| IT8220004A0 (en) | 1982-03-05 |
| JPS57159156A (en) | 1982-10-01 |
| FR2501441B1 (en) | 1985-03-08 |
| DE3207786C2 (en) | 1992-04-16 |
| JPS6348464B2 (en) | 1988-09-29 |
| FR2501441A1 (en) | 1982-09-10 |
| US4485357A (en) | 1984-11-27 |
| NL8101109A (en) | 1982-10-01 |
| GB2095492B (en) | 1984-09-05 |
| DE3207786A1 (en) | 1982-11-04 |
| IT1198356B (en) | 1988-12-21 |
| GB2095492A (en) | 1982-09-29 |
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