CA1199058A - Multi-quadrant brushless dc motor drive - Google Patents
Multi-quadrant brushless dc motor driveInfo
- Publication number
- CA1199058A CA1199058A CA000422263A CA422263A CA1199058A CA 1199058 A CA1199058 A CA 1199058A CA 000422263 A CA000422263 A CA 000422263A CA 422263 A CA422263 A CA 422263A CA 1199058 A CA1199058 A CA 1199058A
- Authority
- CA
- Canada
- Prior art keywords
- current
- motor
- switches
- signal
- level dependent
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
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Classifications
-
- H—ELECTRICITY
- H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
- H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
- H02P6/00—Arrangements for controlling synchronous motors or other dynamo-electric motors using electronic commutation dependent on the rotor position; Electronic commutators therefor
- H02P6/06—Arrangements for speed regulation of a single motor wherein the motor speed is measured and compared with a given physical value so as to adjust the motor speed
Abstract
Abstract of the Disclosure A control circuit for a brushless DC motor providing positive control in all four quadrants of motor operation.
The circuit provides control of motor current, thereby controlling motor torque, in response to an applied current command signal, relying solely on feedback signals of motor current and rotor angle. The heart of the system is a level dependent logic stage which selectively controls the states of the switches applying power to the motor windings in accordance with a current error signal developed by reference to the feedback current signal. The control circuit causes the motor current to correspond to any applied current command and works at any motor speed and torque for rotation in either direction. In a preferred embodiment, the baste control system is combined with additional feedback loops developing servo information regarding rate and actuator position to provide an overall position control system, such as is particularly useful in place of the hydraulic actuator systems now employed in aircraft.
The circuit provides control of motor current, thereby controlling motor torque, in response to an applied current command signal, relying solely on feedback signals of motor current and rotor angle. The heart of the system is a level dependent logic stage which selectively controls the states of the switches applying power to the motor windings in accordance with a current error signal developed by reference to the feedback current signal. The control circuit causes the motor current to correspond to any applied current command and works at any motor speed and torque for rotation in either direction. In a preferred embodiment, the baste control system is combined with additional feedback loops developing servo information regarding rate and actuator position to provide an overall position control system, such as is particularly useful in place of the hydraulic actuator systems now employed in aircraft.
Description
l~ackqround of the ~nven-tion __ 1. Field of the Invention The present invention is related generally to electro-mechanical actuation systems and, more partlcularly, -to a multi-quadrant brushless mo-tor drive system for controlling the mag-nitude and direction of the torque output of a DC motor.
2. Description of the Prior Art Numerous applications reguire an actua-tion system for positioning a specified load as a function of a position command signal. Ac-tuation systems used to control aircraft flight are of particular importance in the design of safe, energy-efficient aircraft. The design of an electromechanical actuation sys-tem or an aircraft flight control system is complicated by the necessity of satisfying numerous, and often conflicting, requirements such as steady-state and dynamic performance, duty cycle, weight, envelope, installation, reliability and cost.
A variety of motor control systems are known, none of which completely satisfies the needs met by arrangements in accord-ance with the present invention. For example, the Masse et al patent 3,683,254 discloses a servo system which contains provision for protecting against instabilities and system per-fo~nance due to system noise. The system provides for auto-matic noise cancellation and acts, throuyh velocity control in the servo system driving circuitry, to limit the drive when-ever the system is subjected to noise. ~owever, the disclosed servo system is directed to the control of a stepping motor, and therefore is not pertinent to the present system for con-trolling a brushless DC motor.
The Dixon e-t al patent 4,019,107 discloses an arranyement for 1 1~ 9C3l38 , ..- .
1 controlling a shunt DC motor, rather than the permanent magnet DC motor 2 to which the present syste~ is directed. Moreover, this system utilizes
A variety of motor control systems are known, none of which completely satisfies the needs met by arrangements in accord-ance with the present invention. For example, the Masse et al patent 3,683,254 discloses a servo system which contains provision for protecting against instabilities and system per-fo~nance due to system noise. The system provides for auto-matic noise cancellation and acts, throuyh velocity control in the servo system driving circuitry, to limit the drive when-ever the system is subjected to noise. ~owever, the disclosed servo system is directed to the control of a stepping motor, and therefore is not pertinent to the present system for con-trolling a brushless DC motor.
The Dixon e-t al patent 4,019,107 discloses an arranyement for 1 1~ 9C3l38 , ..- .
1 controlling a shunt DC motor, rather than the permanent magnet DC motor 2 to which the present syste~ is directed. Moreover, this system utilizes
3 a tachometer readout to develop a motor speed signal which is required
4 in the control system.
~ Im~mura patent 4,155,169 discloses a system desisned to control 6 the velocity of a motor by comparing a velocity signal with a velocity 7 command signal to vary the motor drive. This differs from the present 8 system which is capable of developing control without resort to motor 9 shaft velocity.
The Liska et al patent 4,167,693 discloses a commutation circuit 1~ for controlling a brushless DC motor. However, the contro~ circuitry of 12 this patent i~ principally directed to reducing the sensitivity of the -13 drive circuits to self-induced current spikes.
14 Smooth motor operation is essential in providing the degree of safety and control required in aircraft actuator systems. Such prior 16 art DC motor systems as are known fail to provide suffieiently smooth 17 motor operation for electromechanical actuators used in aircraft flight 18 control systems. A typical prior art DC motor drive system controls the 19 operation of the motor only in the first ~uadrant, where both the ~¦ voltage and current are positive~ and in the third quadrant where both 21¦ the current and voltage are negative. A DC motor drive system which 22 provides driven operation of a motor only in the first and third 23¦ quadrants ignores the transients which some switching actions develop in 241 the second and fourth quadrants. The failure of prior art systems to 25 provide smooth motor operation during voltage and current transients 26 throughout the operating cycle result5 from discontinuities in the logîc 27 which controls the switching actions. The difficulties associated with 2~ the prior art motor drive circuits are,particularly acute in 2g electromechanical actuators such as may be usëd in aircraft flight 30 control systems because the motors in such systems are often operated at 31~ low speed where the effects of logic discontinuities have the greatest ~ ~2-1 effect.
2 There are prior art DC motor drive systems which provide 3 four-quadrant operation; however such systems generally require an input 4 signal indicative of the direction of rotation of the rotor. Known prior art four-quadrant motor drive systems are complex and fail to 6 provide the smoothness of motor operation required for aircraft flight 7 control systems.
8 A particular DC motor control system i~ needed which can provide 9 smootht precisely controllable motor operation for aircraft flight control systems which are suitable for replacing hydraulic actuation 11 systems presently in use.
12 Summary of the Invention 13 In brief, arrangements in accordance with the present invèntion 14 comprise contr~l circuitry for driving a brushless DC motor. This 15 control circuitry controls the application of power to the field 16 windings of the motor and is effective in all four quadrants of motor 17 operation; that is, when -the motor current and voltage are both:of.the 18 same polarity (quadrants I and III) and when the motor current voltage 19 are of opposite polarity with respect to each other (quadrants II and 20 IV~. .
21 In one particular arrangement of a basic csntrol circuit in ~2 accordance with the invention, a current command signal, deriYed as an 23 lerror signal frem a comparison of signals representing shaft position 24 ¦and desired position, is compared with a feedback si~nal corresponding 25 !to current in the switching circuit. The difference between the current 26 ~command signal and the feedback signal is integrated and processed in a 27 ¦dynamic compenSation network of a mathematical form comprising an 28 ¦integration and 2 zero and a pole. The zeAro and the pole form a 2~1 lead-lag network which controls the modulation frequency of the 30 switching control loop~ The th~s-processed current error signal is 31 applied to a level dependent logic stage which comprises a plurality of _3_ ~9~g~5~ `
. .
1 drive swi~ches, each designed to change state between sat~rated on and 2 saturated off condltions at different levels of the control current.
3 The o~tputs of these switches are applied ~o a commutation logic stage 4 where, in combination with signals indicating motor rotor position, the determination is made as to which of the power switches are to be turned 6 on or off to control the application of power to the respective motor 7 windings~
8 Each of the power switches has connected in parallel with it a 9 reversely poled diode to provide a path for motor current when the associated transistor switch i5 turned off. Those switching transistors 11 co~pled to one side of the power supply are tied in parallel to a common 12 series resistor which serves as a first sensing resistor for feedback -13 current. The associated diodes are tied in common to a second series 14 resistor which serves as a secsnd sensing resistor for feedback current.
The feedback current signal is developed from a comparison of the 16 transistor current and diode current, and the absolute value of the - - 17 higher-of-the two sensing current signals is used for- the feedback 18 current.
19 As the error current signal sweeps across the control range of the level dependent switches, the switches respond to select the motor 21 Iwindings which are to be energized, relative to instantaneous rotor 22¦ position. Quasi-steady state conditions may develop in the level 23 dependent log1c in which motor current approaches the level of the 24 Icommand current in a given mode. ~owever, typically, motor current 251 overshoots the current command level. Th~s is sensed through the 26 Ifeedback current, which results in a change in the error current so that 27 Ithe motor current a~ain a~proaches th command level, but from the other 28 ~side with subsequent overshoot back and forth about the command level.
2g¦ This occurs as the error current input to the ~evel dependent logic 3~1 dithers back and forth about the switching point of one or another of 31 the level dependent logic switches. This ditherlng occurs at a rapid i _g_ !
l~ o,r;~
1 ¦rate, compared with the electromechanical response characteristics of 2 the motor circuitry and components--typically in the range of 10 3 kilohert~--with a corresponding high repetition rate for the application 4 of pulses of power to the motor windings. Operation of the level dependent logic switches in the manner described permits pulse width 6 modulation control of motor current. The parameters of the networks in 7 the control loop dynamic compensation network are dependent up~n the 8 characteristics of the motor to be driven and on the desired ripple 9 frequency. A principal function of this ioop is to allow a high frequency oscillation in the error current at a low amplitude. This 11 oscillation allows the motor to be driven under pulse width mod~lation 12 control. Variation of the duty cycle in the pulse width modulation -13 control scheme is effected by selecting thè extent of the hysteresis in 14 the level dependent logic switches (that is, the extent of the potential 15 ¦ difference between the switch-on voltaye and the switch-off-voltage) and-16 ¦ selectin~ the gain of the dynamic compensation network which processes 17 ~-the:error current for application to the level dependent-logic circ~it.
18 ¦ Another dynamic compensation network is provided ahead of the 19 ¦point at which the current command and the feedback current signal are 201 compared. The two dynamic compensation networks serve to prevent 21¦ erroneous switchings in the control circuitry due to command signal ~21 noise. This additional dynamic compensation network opera~es as single 231 pole filter to reject sharp transients, such 25 noise, in the command 2~1 signal. The`input of this dynamic compensation network also includes a 251 current limiting circuit which arbitrarily limits the command si~nal to 26 a level compatible with maximum permissible current levels in the motor 27 windings.
28¦ The level dependent lo~ic allows the appropriate transistor 29 switches to ~e turned on without having to know ~hi h transistors are are required because of variations in operating modes. Ihis operation 31 is possible by virt~e of the feedback current which is switched in
~ Im~mura patent 4,155,169 discloses a system desisned to control 6 the velocity of a motor by comparing a velocity signal with a velocity 7 command signal to vary the motor drive. This differs from the present 8 system which is capable of developing control without resort to motor 9 shaft velocity.
The Liska et al patent 4,167,693 discloses a commutation circuit 1~ for controlling a brushless DC motor. However, the contro~ circuitry of 12 this patent i~ principally directed to reducing the sensitivity of the -13 drive circuits to self-induced current spikes.
14 Smooth motor operation is essential in providing the degree of safety and control required in aircraft actuator systems. Such prior 16 art DC motor systems as are known fail to provide suffieiently smooth 17 motor operation for electromechanical actuators used in aircraft flight 18 control systems. A typical prior art DC motor drive system controls the 19 operation of the motor only in the first ~uadrant, where both the ~¦ voltage and current are positive~ and in the third quadrant where both 21¦ the current and voltage are negative. A DC motor drive system which 22 provides driven operation of a motor only in the first and third 23¦ quadrants ignores the transients which some switching actions develop in 241 the second and fourth quadrants. The failure of prior art systems to 25 provide smooth motor operation during voltage and current transients 26 throughout the operating cycle result5 from discontinuities in the logîc 27 which controls the switching actions. The difficulties associated with 2~ the prior art motor drive circuits are,particularly acute in 2g electromechanical actuators such as may be usëd in aircraft flight 30 control systems because the motors in such systems are often operated at 31~ low speed where the effects of logic discontinuities have the greatest ~ ~2-1 effect.
2 There are prior art DC motor drive systems which provide 3 four-quadrant operation; however such systems generally require an input 4 signal indicative of the direction of rotation of the rotor. Known prior art four-quadrant motor drive systems are complex and fail to 6 provide the smoothness of motor operation required for aircraft flight 7 control systems.
8 A particular DC motor control system i~ needed which can provide 9 smootht precisely controllable motor operation for aircraft flight control systems which are suitable for replacing hydraulic actuation 11 systems presently in use.
12 Summary of the Invention 13 In brief, arrangements in accordance with the present invèntion 14 comprise contr~l circuitry for driving a brushless DC motor. This 15 control circuitry controls the application of power to the field 16 windings of the motor and is effective in all four quadrants of motor 17 operation; that is, when -the motor current and voltage are both:of.the 18 same polarity (quadrants I and III) and when the motor current voltage 19 are of opposite polarity with respect to each other (quadrants II and 20 IV~. .
21 In one particular arrangement of a basic csntrol circuit in ~2 accordance with the invention, a current command signal, deriYed as an 23 lerror signal frem a comparison of signals representing shaft position 24 ¦and desired position, is compared with a feedback si~nal corresponding 25 !to current in the switching circuit. The difference between the current 26 ~command signal and the feedback signal is integrated and processed in a 27 ¦dynamic compenSation network of a mathematical form comprising an 28 ¦integration and 2 zero and a pole. The zeAro and the pole form a 2~1 lead-lag network which controls the modulation frequency of the 30 switching control loop~ The th~s-processed current error signal is 31 applied to a level dependent logic stage which comprises a plurality of _3_ ~9~g~5~ `
. .
1 drive swi~ches, each designed to change state between sat~rated on and 2 saturated off condltions at different levels of the control current.
3 The o~tputs of these switches are applied ~o a commutation logic stage 4 where, in combination with signals indicating motor rotor position, the determination is made as to which of the power switches are to be turned 6 on or off to control the application of power to the respective motor 7 windings~
8 Each of the power switches has connected in parallel with it a 9 reversely poled diode to provide a path for motor current when the associated transistor switch i5 turned off. Those switching transistors 11 co~pled to one side of the power supply are tied in parallel to a common 12 series resistor which serves as a first sensing resistor for feedback -13 current. The associated diodes are tied in common to a second series 14 resistor which serves as a secsnd sensing resistor for feedback current.
The feedback current signal is developed from a comparison of the 16 transistor current and diode current, and the absolute value of the - - 17 higher-of-the two sensing current signals is used for- the feedback 18 current.
19 As the error current signal sweeps across the control range of the level dependent switches, the switches respond to select the motor 21 Iwindings which are to be energized, relative to instantaneous rotor 22¦ position. Quasi-steady state conditions may develop in the level 23 dependent log1c in which motor current approaches the level of the 24 Icommand current in a given mode. ~owever, typically, motor current 251 overshoots the current command level. Th~s is sensed through the 26 Ifeedback current, which results in a change in the error current so that 27 Ithe motor current a~ain a~proaches th command level, but from the other 28 ~side with subsequent overshoot back and forth about the command level.
2g¦ This occurs as the error current input to the ~evel dependent logic 3~1 dithers back and forth about the switching point of one or another of 31 the level dependent logic switches. This ditherlng occurs at a rapid i _g_ !
l~ o,r;~
1 ¦rate, compared with the electromechanical response characteristics of 2 the motor circuitry and components--typically in the range of 10 3 kilohert~--with a corresponding high repetition rate for the application 4 of pulses of power to the motor windings. Operation of the level dependent logic switches in the manner described permits pulse width 6 modulation control of motor current. The parameters of the networks in 7 the control loop dynamic compensation network are dependent up~n the 8 characteristics of the motor to be driven and on the desired ripple 9 frequency. A principal function of this ioop is to allow a high frequency oscillation in the error current at a low amplitude. This 11 oscillation allows the motor to be driven under pulse width mod~lation 12 control. Variation of the duty cycle in the pulse width modulation -13 control scheme is effected by selecting thè extent of the hysteresis in 14 the level dependent logic switches (that is, the extent of the potential 15 ¦ difference between the switch-on voltaye and the switch-off-voltage) and-16 ¦ selectin~ the gain of the dynamic compensation network which processes 17 ~-the:error current for application to the level dependent-logic circ~it.
18 ¦ Another dynamic compensation network is provided ahead of the 19 ¦point at which the current command and the feedback current signal are 201 compared. The two dynamic compensation networks serve to prevent 21¦ erroneous switchings in the control circuitry due to command signal ~21 noise. This additional dynamic compensation network opera~es as single 231 pole filter to reject sharp transients, such 25 noise, in the command 2~1 signal. The`input of this dynamic compensation network also includes a 251 current limiting circuit which arbitrarily limits the command si~nal to 26 a level compatible with maximum permissible current levels in the motor 27 windings.
28¦ The level dependent lo~ic allows the appropriate transistor 29 switches to ~e turned on without having to know ~hi h transistors are are required because of variations in operating modes. Ihis operation 31 is possible by virt~e of the feedback current which is switched in
-5-1 larity after deri~ation o: tle absolu value ot the highest sensed 2 current, depending upon whether or not the motor is to be driven in the 3 counterclockwise direction. Thus, in this current loop as described, it 4 is not necessary that a tachometer signal be provided to indicate motor speed or direction.
6 In one particular application of the motor control system of the
7 invention described above, a tachometer 700p and actuator position
8 sensor loop are added, together with appropriate comparison circuitry, to develop t~e current command signal. In that control circuit, a position command, indicative of the desired position of an actuator 11 driven by the motor, is compared with a feedback signal indicating true 12 actuator position. The difference is applied as an error signal to a 13 further comparision stage for combination with a rate feedback signal 14 from a tachometer coupled to the motor. A tachometer signal is useful in determining the level of current for the current command signal.
16 Rate feedback in this manner allows the use of higher loop gains for 17 improved performance of the overall control circuit, and helps to 18 mitigate instabilit~ du~ to non-linearities and large inertial loads.
lg Brief Description of the Drawing 2b A better understanding of the present invention may be had from 21 a consideration of the following detailed description, taken in 22 conjunction with the accompanying drawing in which:
23 ¦ Fig. 1 is a graph ilIustrating the switching parameters of motor 24 Icontrol switches employed in arrangements embodying the invention;
251 Fig. 2 i5 a circuit diagram provided for purposes of 26 ¦illustration in explaining the operation of the invention, in 271 conjunction with Fig. l;
28¦ Fig~ 3 is a graph showing particular ~aveforms developed in the 29j operation of a motor ccntrol circuit of the presen`t invention;
301 Fig~ 4 is a block diagram of a control circuit in accordance 31 wi h the invention;
~ 05~
1 Flg. 5 is a simplified circ~it diagram of the motor switching 2 circuit of Fig 4, 3 Fig. 6, comprising parts 6a and 6b, is a schematic diagram 4 representing portions of the control circuitry of Fig. 4;
Fig. 7 is a graphic representation, similar to Fig. 1, showing 6 the conditions of various switching stages in the level dependent logic 7 as a function of control voltage; and 8 Fig. 8 is a partial diagram of the circuit of Fig. 5,
16 Rate feedback in this manner allows the use of higher loop gains for 17 improved performance of the overall control circuit, and helps to 18 mitigate instabilit~ du~ to non-linearities and large inertial loads.
lg Brief Description of the Drawing 2b A better understanding of the present invention may be had from 21 a consideration of the following detailed description, taken in 22 conjunction with the accompanying drawing in which:
23 ¦ Fig. 1 is a graph ilIustrating the switching parameters of motor 24 Icontrol switches employed in arrangements embodying the invention;
251 Fig. 2 i5 a circuit diagram provided for purposes of 26 ¦illustration in explaining the operation of the invention, in 271 conjunction with Fig. l;
28¦ Fig~ 3 is a graph showing particular ~aveforms developed in the 29j operation of a motor ccntrol circuit of the presen`t invention;
301 Fig~ 4 is a block diagram of a control circuit in accordance 31 wi h the invention;
~ 05~
1 Flg. 5 is a simplified circ~it diagram of the motor switching 2 circuit of Fig 4, 3 Fig. 6, comprising parts 6a and 6b, is a schematic diagram 4 representing portions of the control circuitry of Fig. 4;
Fig. 7 is a graphic representation, similar to Fig. 1, showing 6 the conditions of various switching stages in the level dependent logic 7 as a function of control voltage; and 8 Fig. 8 is a partial diagram of the circuit of Fig. 5,
9 illustrating a modification thereof.
Description of the Preferred Embodiment 17 It might be helpful to an understanding of the concept of the 12 present invention to briefly discuss a simplified control circuit for an 13 electric motor. Fig. 1 i~ a graphical representation, without scale 14 designation, of the four power control switches TSl-TS4 in Fig. 2.
These switches are shown in Fig. 2 as connecting a motor between plus 16 and minus line voltage. Each of the switches is connected in parallel 17 with a diode (Dl to D4) poled for conducting current in the opposite 18 direction to its associated switch. The switches have been given 19 designations indicative of their position in the circuit and their 20 driving effect on the motor M. Thus, switch TSl is designated CWP (for 21 ¦"clockwise positiven). Switch TS2 is designated CCWP (counterclockwise 22¦ positive), TS3 is designated CCWM (counterclockwise minus~ and switch 231 TS4 is designated CWM (clockwise minus).
241 The graphical repre~entation of Fig. 1 indicates the states of their respective switches TSl-TS4 as a function of an applied bias 26 ¦voltage (corresponding to a variable control level). As the voltage 27 ¦applied to the respective control terminals of the respective switches 28¦ is varied between a value of -V, at which all switches are in the 2~1 condition indicated at the left side of the figure, to +V, at which all 301 switches are in the condition indicated at the right-hand side of the 31¦ figure, the switches change state along paths indicated by the arrows.
l -7-~ ~ 9 ~tj~
In changing state between saturated off and saturated on conditions ! each of the swi-tches changes state from off to on at a slightly different bias potential from that at which it changed state from on to off, thus exhibiting the hysteresis loops shown in Fig. 1. The arrows indicate the direction of the change of state versus con-trol voltage. Moreover, -the switching circuits are designed so that each switch exhibits its changes of s-tate at a different level of control voltage.
There are three cases to consider in discussing the operation of the simplified control circuit of Fig. 2: motor current negative, motor curren-t zero, and motor current positive.
Case 1: Negative Motor Current With the control voltage at -V (see Eig. 1) both counter~
clockwise switches TS2 and TS3 are on, while the clockwise switches TSl and TS4 are off. This results in a motor voltage of -VL across the motor M. As the control voltage becomes more positive, switch TS2 turns off first, resulting in zero voltage being applied to the motor. The back EMF of the motor now drives current through the conducting switch TS3, then through the transistor current sense resistor RSl and back through the diode current sense resistor RS2 and diode D4. It will be noted that when a shunting diode conducts as the result of turning ofr one of the control switches, current in the diode sensing resistor is opposite in direction, relative to line potentia~, to current in the transistor sensing resistor. The resulting current sensing levels are picked off from the circuit of Fig. 2 as feedback currents ISl and IS2, for tran-sistor current sensing and diode current sensing, respectively.
As the control voltage is increased further, the switch TS3 turns off and the energy stored in the inductance of the motor drives current to and from the line through diodes Dl and D4, respec-tively. The voltage applied across the motor is t-VL.
5~
Case 2: Zero Motor Curren-t Again starting with the most negative input (-V of Fig. 1) to the level dependent logic, this condition results in -VL
being applied to -the motor, as before. ~s the control voltage is increased, switch TS2 turns off and the application of line voltage VL is removed, so that the voltage across the motor becomes equal to its back EMF. This is the voltage seen across the motor until the applied contro] voltage reaches the highest level, at which swi-tch TSl is turned on. ~t this point voltage +VL is applied across the motor.
Case 3: Positive Motor Current ~ tarting again with the most negative applied control voltage, -VL is applied to the motor as before, although the direction of motor current is opposite to that of Case 1, corresponding to a deceleration of clockwise rotation. As the level of control voltages increases, the voltage across the motor remains -VL until the switch TS4 is turned on, result-ing in zero volts being applied across the motor. Thereafter, when the switch TSl turns on, the voltage becomes +VL across the motor.
The following table may be constructed to illustrate the different levels of motor voltage for the different switch conditions.
-V TABLE I +V
CWP off off off off on) CWM off off off on on) C~WM on on off off off) Switch condition CCWP on off off off off I~O -VL O +VL +VL +VL) I=O -VL VBEMF VBEMF VBEMF +VL) Motor voltage I > O -VL -VL -VL O +VL ) Upon examination of Table I, it can be seen tha-t in every case as the input level increases to the control switches (level dependent logic) the motor voltage increases. The motor current affects where the changes to the motor voltage take place. Since the difference between the desired or commanded motor current and the actual motor current is _9_ ~ 9~s~
1 ~integrated, the time average of the actual motor current will equal the 2 commanded motor current. The output of the integration (the control 3 error signal) will always be moving in the direction to bring the actual 4 motor voltage toward the desired motor voltage (and cur~ent). Since the motor voltage switches from one discrete level to another, it is always 6 too low or too high and the feedback current changes accordingly, th~s 7 causing the error signal applied as control to the level dependent B switches to oscillate correspondingly.
9 Fig~ 3 illustrates waveforms showing motor operation for a brief 1~¦ interval during a specific control mode. The command current Ic is 11¦ shown rising to a steady state level. The motor current ~m follows and 12 ¦ overshoots, then oscillates back and forth about the command current 13 I level~ The error current Ie which is applied to the control switches 14 ¦ for the example of Fis. 1 dithers back and forth across the biasing 15 ¦ level at which a particular one of the switches changes state. -The 16 ¦ voltage to the motor (Vm) shifts correspondingly between +VL and zero.
17 ¦ Thus, when the motor voltage Vm is first switched off, the slope of the 18 I motor current Im changes sign and motor current decreases. This causes 19 ¦the feedback current to change, thus causing the error signal Ie to move 20 I positively until a particular switch affecting control turns on. Motor 21 Icurrent then increases until the switch is turned off and the cycle 22 I repeats. The time span for the segments of waveform illustrated in FigO
23 13 is approximately one millisecond. Thus it can be seen that the 24 ¦oscillation fre~uency is about 10 kilohertz.
25 I The block diayram of Fig. 4 shows a control circuit 10 for a 26 ¦ motor 11 coupled to drive an actuator 12, which may, for example, be a 27 I member coupled to control the position of a control surface of an 28 ~ aircraft, the position of which is to be set in response to an applied 2g I position co~,and signal.
30 I A position sensor 14 is coupled to the actuator 12 to develop a 31 I position signal which is applied to a summing stage 16 for comparison I! -10-with the applied position command signal. The difference between the two signals is amplified in amplifier 18 and applied to comparison stage ~0 for comparison with a rate feedback signal from a rate feedback loop including a tachome-ter 22 coupled to the motor 11 and providing a rate signal through a demodulator 24 and amplifier 26. The thus-modified position error signal is applied to a dynamic compensation stage 28 which has at its input a current limiting circuit 29. This current limiting circuit 29 may comprise an operational amplifier having a pair of Zener diodes for limiting the command signal between predetermined limits which correspond to the particular maximum currents to be permitted in the motor 11.
The portion of the circuit of Fig. 4 shown within the broken line lOA represents the basic control circuitry of my preferred embodiment, and it will be demonstrated hereinafter that this control circuit can be used to control a motor with-out the need for a rate feedback signal, as developed from the tachometer 22 and applied to the comparator 20 to modify or condition the applied position error signal. Thus, any suitable current comrnand signal can be applied as input to the current limiter 29 and first dynamic compensation stage 28 to drive the basic control circuit lOA. In the preferred embodi-ment, the dynamic compensation stage 28 includes a single pole filter providing a roll-off at 15,000 radians per second.
This has the LaPlace transform function l/(S/15,000 + 1). The resulting filtered current command signal from the dynamic compensation stage 28 is applied to a comparison stage 30 which also receives as its input a feedback current signal from a feedback loop which includes a current sensor 41 coupled to a power switching circuit 40 which drives the motor 11. As will be e~plained further, this current sensor 41 provides a signal equal to the absolute value of the larger of two currents which are sensed in -the switching circuit 40. This absolute 9 C~
value signal is -then multiplied in the stage 42 by a signal (designated NORMAL) from the level dependent logic stage 34 which has a value of ei-ther plus or minus 1. The output stage 42 will then be either posi-tive or negative, depending on the stage of the NORMAL signal from the stage 34. This feedback signal is then fed through a filter 44, preferably having a roll-off at approximately 314,000 radians per second, having the LaPlace transform l/(S0314~000 + 1). The result of the comparison in stage 30 of the corNnand current and the filtered feedback current becomes the error signal Ie which is applied to the second dynamic compensation stage 32. There the signal is amplified, integrated and filtered in accordance with a function having the ~aPlace transform KN/[S(S/l99,000 + 1)], wherein, in one preferred embodiment, the gain K equals 725, and N = 0.00005Q3S + 1. As noted above, the compensating functions of dynamic compensation stages 28 and 32 operate together to prevent erroneous switching due to command signal noise. The compensation provided in dynamic compensation stage 32 is used to control the modulation frequency of the switching.
The rest of the circuit of Fig. 4 comprises the level depend-ent logic stage 34, coupled to receive the compensated error signal Ie from stage 32 and provide signals to a following comrnutation logic stage 38 representing desired on and off states for the switches corresponding to those shown in Fig. 2 (CWP, CWM, CCWM, and CCWP, and their complements). These are combined with phase signals (01, 02 and 03 and the comple-ments thereof for the three phase winding shown in Fig. 5) as developed from the motor 11 by a rotor position sensor stage 3~. Circuitry in the commutation logic stage 38 develops control signals for the switching circuit 40 which drive the motor 11 frorn a power source.
Fig. 5 is an exemplary circuit corresponding to the switch-iny circuit 40 of Fig. 4 wherein the mo-tor 11 is represented schematically as having windings A, B and C (respectively numbered 50, 48 ~nd 46) connected in a wye configuration and coupled to switching transistors A-, B-, C-, A-~, B+ and C+.
The switching transistors shown in Fig. 5 are merely exemplary of solid stage switching devices. It will be . . . . . . .
-12a-.. ,.;. . I
1 ¦understood that the actual circuits corresponding to these individual 2 ¦ transistors may comprise conventional power switching circuits as are 3 ¦ known in the ar~. In the circuit of Fig. S, the negative transistor-~4 ¦ A-, B- and C- have their emitters coupled together and connected to 51 ground through a series sensing resistor RSl. The positive switching 61 transistor~ Af, B~ and C~ have their collectors coupled to~ether and 71 connected to a positiYe line voltage ter~ninal 52. The free end of each 81 motor winding is coupled between the two transistors of a pair. Thus, 9 for example, the pair of transistors A- and A+ which are connected to the A winding 50 have the capability of connecting the winding 50 either 11 to ground or to positive VL termin~l 52. The same is true for the 1~ remaining transiStor pairs and the corresponding motor windings 48 and -13 46.
14¦ Each of the switching transistors of Fig. 5 has an asso~iated 1~¦ diode 56, 58, 60, 62, 64 or 66, connected in parallel with it, but with 16¦ a polarity such as to permit current flow in the opposite direction to 17¦ that through the associated transistor. Diodes 56, 58 and 60 are 18¦ coupl~d together and connected to ~round through a diode current sensing 19¦ resistor RS2.
20¦ By compariSon with the circ~it of Fig. 2, the similarities 21 between the switching transistors of Fig. S and the switching ~ -22 transistors of Fig.~2 can be discerned, except that the circuit of Fig~
23 ¦ 2 is for a single phase motor, whereas the circuit of Fig. 5 i~ for a 241 three phase motor. As indicated in Fig. 2, the transistor current 251 sensing and diode current sensing signals ISl and IS2 are picked off the 26¦ sensing resistorS RSl and RS2 for application in the current sensor 27¦ stage 41 ~Fig. 4).
28¦ Fig. ~ is a schematic diagram illustratin~ particular details of 291 the circuitry employed in the level dependent Iogic stage 34 and the 301 commutation logic stage 3B for driving the switching circ~it 40 ~see 31¦ Figs. 4 and 53. As shown in Fig~ 6a, the integrated error si~nal Ie is L1~ ,$~
1 dropped across a voltage divider comprising resistors Rl and R2 and 2 filter capacitor Cl and applied to one input of each of four level 3 dependent switches ~lAI UlB, UlC and UlD. These are connected across 4 positive and negative power supplies as shown, and furthermore the two ~ switches UlA and UlB have second input terminals coupled through 6 resistors R9 and R10 to the ~5 volt supply while the switches UlC and 7 UlD have their second input terminals coupled through resistors R13 and 8 R14 to the -5 volt supply. The different connections and the different 9 values of the input cou~ling resistors develop different bias levels for
Description of the Preferred Embodiment 17 It might be helpful to an understanding of the concept of the 12 present invention to briefly discuss a simplified control circuit for an 13 electric motor. Fig. 1 i~ a graphical representation, without scale 14 designation, of the four power control switches TSl-TS4 in Fig. 2.
These switches are shown in Fig. 2 as connecting a motor between plus 16 and minus line voltage. Each of the switches is connected in parallel 17 with a diode (Dl to D4) poled for conducting current in the opposite 18 direction to its associated switch. The switches have been given 19 designations indicative of their position in the circuit and their 20 driving effect on the motor M. Thus, switch TSl is designated CWP (for 21 ¦"clockwise positiven). Switch TS2 is designated CCWP (counterclockwise 22¦ positive), TS3 is designated CCWM (counterclockwise minus~ and switch 231 TS4 is designated CWM (clockwise minus).
241 The graphical repre~entation of Fig. 1 indicates the states of their respective switches TSl-TS4 as a function of an applied bias 26 ¦voltage (corresponding to a variable control level). As the voltage 27 ¦applied to the respective control terminals of the respective switches 28¦ is varied between a value of -V, at which all switches are in the 2~1 condition indicated at the left side of the figure, to +V, at which all 301 switches are in the condition indicated at the right-hand side of the 31¦ figure, the switches change state along paths indicated by the arrows.
l -7-~ ~ 9 ~tj~
In changing state between saturated off and saturated on conditions ! each of the swi-tches changes state from off to on at a slightly different bias potential from that at which it changed state from on to off, thus exhibiting the hysteresis loops shown in Fig. 1. The arrows indicate the direction of the change of state versus con-trol voltage. Moreover, -the switching circuits are designed so that each switch exhibits its changes of s-tate at a different level of control voltage.
There are three cases to consider in discussing the operation of the simplified control circuit of Fig. 2: motor current negative, motor curren-t zero, and motor current positive.
Case 1: Negative Motor Current With the control voltage at -V (see Eig. 1) both counter~
clockwise switches TS2 and TS3 are on, while the clockwise switches TSl and TS4 are off. This results in a motor voltage of -VL across the motor M. As the control voltage becomes more positive, switch TS2 turns off first, resulting in zero voltage being applied to the motor. The back EMF of the motor now drives current through the conducting switch TS3, then through the transistor current sense resistor RSl and back through the diode current sense resistor RS2 and diode D4. It will be noted that when a shunting diode conducts as the result of turning ofr one of the control switches, current in the diode sensing resistor is opposite in direction, relative to line potentia~, to current in the transistor sensing resistor. The resulting current sensing levels are picked off from the circuit of Fig. 2 as feedback currents ISl and IS2, for tran-sistor current sensing and diode current sensing, respectively.
As the control voltage is increased further, the switch TS3 turns off and the energy stored in the inductance of the motor drives current to and from the line through diodes Dl and D4, respec-tively. The voltage applied across the motor is t-VL.
5~
Case 2: Zero Motor Curren-t Again starting with the most negative input (-V of Fig. 1) to the level dependent logic, this condition results in -VL
being applied to -the motor, as before. ~s the control voltage is increased, switch TS2 turns off and the application of line voltage VL is removed, so that the voltage across the motor becomes equal to its back EMF. This is the voltage seen across the motor until the applied contro] voltage reaches the highest level, at which swi-tch TSl is turned on. ~t this point voltage +VL is applied across the motor.
Case 3: Positive Motor Current ~ tarting again with the most negative applied control voltage, -VL is applied to the motor as before, although the direction of motor current is opposite to that of Case 1, corresponding to a deceleration of clockwise rotation. As the level of control voltages increases, the voltage across the motor remains -VL until the switch TS4 is turned on, result-ing in zero volts being applied across the motor. Thereafter, when the switch TSl turns on, the voltage becomes +VL across the motor.
The following table may be constructed to illustrate the different levels of motor voltage for the different switch conditions.
-V TABLE I +V
CWP off off off off on) CWM off off off on on) C~WM on on off off off) Switch condition CCWP on off off off off I~O -VL O +VL +VL +VL) I=O -VL VBEMF VBEMF VBEMF +VL) Motor voltage I > O -VL -VL -VL O +VL ) Upon examination of Table I, it can be seen tha-t in every case as the input level increases to the control switches (level dependent logic) the motor voltage increases. The motor current affects where the changes to the motor voltage take place. Since the difference between the desired or commanded motor current and the actual motor current is _9_ ~ 9~s~
1 ~integrated, the time average of the actual motor current will equal the 2 commanded motor current. The output of the integration (the control 3 error signal) will always be moving in the direction to bring the actual 4 motor voltage toward the desired motor voltage (and cur~ent). Since the motor voltage switches from one discrete level to another, it is always 6 too low or too high and the feedback current changes accordingly, th~s 7 causing the error signal applied as control to the level dependent B switches to oscillate correspondingly.
9 Fig~ 3 illustrates waveforms showing motor operation for a brief 1~¦ interval during a specific control mode. The command current Ic is 11¦ shown rising to a steady state level. The motor current ~m follows and 12 ¦ overshoots, then oscillates back and forth about the command current 13 I level~ The error current Ie which is applied to the control switches 14 ¦ for the example of Fis. 1 dithers back and forth across the biasing 15 ¦ level at which a particular one of the switches changes state. -The 16 ¦ voltage to the motor (Vm) shifts correspondingly between +VL and zero.
17 ¦ Thus, when the motor voltage Vm is first switched off, the slope of the 18 I motor current Im changes sign and motor current decreases. This causes 19 ¦the feedback current to change, thus causing the error signal Ie to move 20 I positively until a particular switch affecting control turns on. Motor 21 Icurrent then increases until the switch is turned off and the cycle 22 I repeats. The time span for the segments of waveform illustrated in FigO
23 13 is approximately one millisecond. Thus it can be seen that the 24 ¦oscillation fre~uency is about 10 kilohertz.
25 I The block diayram of Fig. 4 shows a control circuit 10 for a 26 ¦ motor 11 coupled to drive an actuator 12, which may, for example, be a 27 I member coupled to control the position of a control surface of an 28 ~ aircraft, the position of which is to be set in response to an applied 2g I position co~,and signal.
30 I A position sensor 14 is coupled to the actuator 12 to develop a 31 I position signal which is applied to a summing stage 16 for comparison I! -10-with the applied position command signal. The difference between the two signals is amplified in amplifier 18 and applied to comparison stage ~0 for comparison with a rate feedback signal from a rate feedback loop including a tachome-ter 22 coupled to the motor 11 and providing a rate signal through a demodulator 24 and amplifier 26. The thus-modified position error signal is applied to a dynamic compensation stage 28 which has at its input a current limiting circuit 29. This current limiting circuit 29 may comprise an operational amplifier having a pair of Zener diodes for limiting the command signal between predetermined limits which correspond to the particular maximum currents to be permitted in the motor 11.
The portion of the circuit of Fig. 4 shown within the broken line lOA represents the basic control circuitry of my preferred embodiment, and it will be demonstrated hereinafter that this control circuit can be used to control a motor with-out the need for a rate feedback signal, as developed from the tachometer 22 and applied to the comparator 20 to modify or condition the applied position error signal. Thus, any suitable current comrnand signal can be applied as input to the current limiter 29 and first dynamic compensation stage 28 to drive the basic control circuit lOA. In the preferred embodi-ment, the dynamic compensation stage 28 includes a single pole filter providing a roll-off at 15,000 radians per second.
This has the LaPlace transform function l/(S/15,000 + 1). The resulting filtered current command signal from the dynamic compensation stage 28 is applied to a comparison stage 30 which also receives as its input a feedback current signal from a feedback loop which includes a current sensor 41 coupled to a power switching circuit 40 which drives the motor 11. As will be e~plained further, this current sensor 41 provides a signal equal to the absolute value of the larger of two currents which are sensed in -the switching circuit 40. This absolute 9 C~
value signal is -then multiplied in the stage 42 by a signal (designated NORMAL) from the level dependent logic stage 34 which has a value of ei-ther plus or minus 1. The output stage 42 will then be either posi-tive or negative, depending on the stage of the NORMAL signal from the stage 34. This feedback signal is then fed through a filter 44, preferably having a roll-off at approximately 314,000 radians per second, having the LaPlace transform l/(S0314~000 + 1). The result of the comparison in stage 30 of the corNnand current and the filtered feedback current becomes the error signal Ie which is applied to the second dynamic compensation stage 32. There the signal is amplified, integrated and filtered in accordance with a function having the ~aPlace transform KN/[S(S/l99,000 + 1)], wherein, in one preferred embodiment, the gain K equals 725, and N = 0.00005Q3S + 1. As noted above, the compensating functions of dynamic compensation stages 28 and 32 operate together to prevent erroneous switching due to command signal noise. The compensation provided in dynamic compensation stage 32 is used to control the modulation frequency of the switching.
The rest of the circuit of Fig. 4 comprises the level depend-ent logic stage 34, coupled to receive the compensated error signal Ie from stage 32 and provide signals to a following comrnutation logic stage 38 representing desired on and off states for the switches corresponding to those shown in Fig. 2 (CWP, CWM, CCWM, and CCWP, and their complements). These are combined with phase signals (01, 02 and 03 and the comple-ments thereof for the three phase winding shown in Fig. 5) as developed from the motor 11 by a rotor position sensor stage 3~. Circuitry in the commutation logic stage 38 develops control signals for the switching circuit 40 which drive the motor 11 frorn a power source.
Fig. 5 is an exemplary circuit corresponding to the switch-iny circuit 40 of Fig. 4 wherein the mo-tor 11 is represented schematically as having windings A, B and C (respectively numbered 50, 48 ~nd 46) connected in a wye configuration and coupled to switching transistors A-, B-, C-, A-~, B+ and C+.
The switching transistors shown in Fig. 5 are merely exemplary of solid stage switching devices. It will be . . . . . . .
-12a-.. ,.;. . I
1 ¦understood that the actual circuits corresponding to these individual 2 ¦ transistors may comprise conventional power switching circuits as are 3 ¦ known in the ar~. In the circuit of Fig. S, the negative transistor-~4 ¦ A-, B- and C- have their emitters coupled together and connected to 51 ground through a series sensing resistor RSl. The positive switching 61 transistor~ Af, B~ and C~ have their collectors coupled to~ether and 71 connected to a positiYe line voltage ter~ninal 52. The free end of each 81 motor winding is coupled between the two transistors of a pair. Thus, 9 for example, the pair of transistors A- and A+ which are connected to the A winding 50 have the capability of connecting the winding 50 either 11 to ground or to positive VL termin~l 52. The same is true for the 1~ remaining transiStor pairs and the corresponding motor windings 48 and -13 46.
14¦ Each of the switching transistors of Fig. 5 has an asso~iated 1~¦ diode 56, 58, 60, 62, 64 or 66, connected in parallel with it, but with 16¦ a polarity such as to permit current flow in the opposite direction to 17¦ that through the associated transistor. Diodes 56, 58 and 60 are 18¦ coupl~d together and connected to ~round through a diode current sensing 19¦ resistor RS2.
20¦ By compariSon with the circ~it of Fig. 2, the similarities 21 between the switching transistors of Fig. S and the switching ~ -22 transistors of Fig.~2 can be discerned, except that the circuit of Fig~
23 ¦ 2 is for a single phase motor, whereas the circuit of Fig. 5 i~ for a 241 three phase motor. As indicated in Fig. 2, the transistor current 251 sensing and diode current sensing signals ISl and IS2 are picked off the 26¦ sensing resistorS RSl and RS2 for application in the current sensor 27¦ stage 41 ~Fig. 4).
28¦ Fig. ~ is a schematic diagram illustratin~ particular details of 291 the circuitry employed in the level dependent Iogic stage 34 and the 301 commutation logic stage 3B for driving the switching circ~it 40 ~see 31¦ Figs. 4 and 53. As shown in Fig~ 6a, the integrated error si~nal Ie is L1~ ,$~
1 dropped across a voltage divider comprising resistors Rl and R2 and 2 filter capacitor Cl and applied to one input of each of four level 3 dependent switches ~lAI UlB, UlC and UlD. These are connected across 4 positive and negative power supplies as shown, and furthermore the two ~ switches UlA and UlB have second input terminals coupled through 6 resistors R9 and R10 to the ~5 volt supply while the switches UlC and 7 UlD have their second input terminals coupled through resistors R13 and 8 R14 to the -5 volt supply. The different connections and the different 9 values of the input cou~ling resistors develop different bias levels for
10 the switches UlA-UlD so that the switches change state for different
11 levels of applie~ error signal Ie, applied at the terminal block 70.
12 Phase signals from the rotor position sensor 36 (Fig. 4) are also
13 applied at the terminal block 70 and fed directly through to the
14 commutation lGgic stage 38. These lines are connected respectively to the +5 volt power supply through resistors R3-R8. Plus 15 volts, -15 16 volts, and ground connections are provided to establish the +5 volt and 17 -5 volt power supplies. The ~5 volt power supply comprises a Zener 18 diode 76 and filter capacitors C2 and C3, connected to the ~15 volt line 19 72 through series resistor Rll. The plus voltage for the switches 20 UlA-UlD is taken from another 15 volt line through series resistor Rl9, 21 filtered by capacitors C6 and C7.
22 The -5 volt supply is developed from -15 volts supplied through 23 series resistor R12 and the Zener diode 78 and parallel capacitors C4 24 and CS. Each of the switches Ul~-UlD is provided with resistors R15-R18 251 and R20-R23, respectively, connected as shown, plus output coupling 26¦ resistors R24, R26, R28 and R30, respectively. These switch output 27 lines are further connected to the +5 volt line through respective 28 resistors R25, R27, R29 and R31. As noted in^ Fig. 6a immediately above 29 these resistors, these output lines correspond respectively to the switching signals representin~ CWP complement, CWM complement, CCWM and 31 CC~;P. These outp~t signals from the level dependent switches in level ! -14-I .' ~9~.5 '- .
1 dependent logic sta~e 34 are applied to comparators in the commutation 2 logic stage 38 for combination with the respective phase signals ~ 2, 3 ~3 and their respective complements which are combined in the manner 4 indicated in the circuit of the commutation logic stage 3B by means cf the NOR gates U2~-C~ U~A-C, U4A-D, U5A-C, U6A-C, and U7A-D as shown.
6 The six outputs of the commutation logic stage 38 are indicated as the 7 complements of signals A+, B+, C+, A-, B- and C- which correspond to th~
8 followin~ commutation equations by virtue of the operation of the 9 commutation logic on the variables indicated: .
12 A* =~1 ~3 CWP ~ ~1 03 CCWP
13 B+ =52 ~3 CWP + ~2 ~3 CCWP
1~ c~ 2 ~WP + ~1 ~2 CCWP
22 The -5 volt supply is developed from -15 volts supplied through 23 series resistor R12 and the Zener diode 78 and parallel capacitors C4 24 and CS. Each of the switches Ul~-UlD is provided with resistors R15-R18 251 and R20-R23, respectively, connected as shown, plus output coupling 26¦ resistors R24, R26, R28 and R30, respectively. These switch output 27 lines are further connected to the +5 volt line through respective 28 resistors R25, R27, R29 and R31. As noted in^ Fig. 6a immediately above 29 these resistors, these output lines correspond respectively to the switching signals representin~ CWP complement, CWM complement, CCWM and 31 CC~;P. These outp~t signals from the level dependent switches in level ! -14-I .' ~9~.5 '- .
1 dependent logic sta~e 34 are applied to comparators in the commutation 2 logic stage 38 for combination with the respective phase signals ~ 2, 3 ~3 and their respective complements which are combined in the manner 4 indicated in the circuit of the commutation logic stage 3B by means cf the NOR gates U2~-C~ U~A-C, U4A-D, U5A-C, U6A-C, and U7A-D as shown.
6 The six outputs of the commutation logic stage 38 are indicated as the 7 complements of signals A+, B+, C+, A-, B- and C- which correspond to th~
8 followin~ commutation equations by virtue of the operation of the 9 commutation logic on the variables indicated: .
12 A* =~1 ~3 CWP ~ ~1 03 CCWP
13 B+ =52 ~3 CWP + ~2 ~3 CCWP
1~ c~ 2 ~WP + ~1 ~2 CCWP
15 ¦ A~ 3 CWM + ~1 ~3 CCWM
16 ¦ . B- =02 03 C~M + ~2 ~3 CCWM
17 I C- = ~1 ~2 CWM + ~1 ~2 CCWM
19¦ The development of the commutation equations in the circuitry of 201 Fig~ 6 may be considered by an example with respect to the upper line 21 involving the NOR gates U2A, U2B and U4A. The inputs to NOR gate U2A
22 are ~1 and the complements of CWP a,nd ~3. The vutput of NOR gate U2A, 23 ¦ applied as one of the inputs to NOR gate U4A, will thus be true ~or 24 j representing the value 1~ only when all three of the inputs are false 2S ~or O~. This is e~uivalent to 01~3CWP.
26 ~ Similarly, the inputs to NOR gate U2B are ~3 and the complements 27 of ~1 and CCWP, the latter being developed by inversion through the NOR
28 gate U4D. The output of NOR gate U2B, applied as the other input to NO~
29 gate U4A, is 01~3CCWP. The output of NOR gat'e U~A thus becomes the complement of the co~nutation equation for A~, which becomes inverted in 31 ¦ the transistor stage ~4 ts develop the A+ signal as expressed in the 32 ~ ;
l -15-g0~
1 commutation equationsO
2 The remaining portion of Fig. 6, located within the broken lines 3 to the right of the figure and designated 40A, represents a driving 4 stage to the power switching transistors of the switching transistors designated 40B in FigO S. As shown in Fig. 6b the driving transistors 6 are designated by numbers 84-89, having respective resistors R33-R38 7 connected to the +15 volt side of a power supply which is center tapped 8 to ground and has filter capacitors C8 and C9 coupled to ~15 volts and 9 capacitors C10 and Cll coupled to -15 volts. -15 volts is fed out through the terminal block 103 as the minus side of the respective drive 11 line pairs carrying the signals corresponding to the commutation 12 equations. It wlll be understood that, although the transistor switches -13 of Fig. 5 are shown as single transistors for simplicity and ease of 14 understanding, these transistors are in fact, as indicated above, more complex power switching circuits to which dual drive lines are connected 16 for control~ The other one of each pair of the output control lines 17 from the terminal block 103 in Fig. 6 is coupled to the o~tput terminal
19¦ The development of the commutation equations in the circuitry of 201 Fig~ 6 may be considered by an example with respect to the upper line 21 involving the NOR gates U2A, U2B and U4A. The inputs to NOR gate U2A
22 are ~1 and the complements of CWP a,nd ~3. The vutput of NOR gate U2A, 23 ¦ applied as one of the inputs to NOR gate U4A, will thus be true ~or 24 j representing the value 1~ only when all three of the inputs are false 2S ~or O~. This is e~uivalent to 01~3CWP.
26 ~ Similarly, the inputs to NOR gate U2B are ~3 and the complements 27 of ~1 and CCWP, the latter being developed by inversion through the NOR
28 gate U4D. The output of NOR gate U2B, applied as the other input to NO~
29 gate U4A, is 01~3CCWP. The output of NOR gat'e U~A thus becomes the complement of the co~nutation equation for A~, which becomes inverted in 31 ¦ the transistor stage ~4 ts develop the A+ signal as expressed in the 32 ~ ;
l -15-g0~
1 commutation equationsO
2 The remaining portion of Fig. 6, located within the broken lines 3 to the right of the figure and designated 40A, represents a driving 4 stage to the power switching transistors of the switching transistors designated 40B in FigO S. As shown in Fig. 6b the driving transistors 6 are designated by numbers 84-89, having respective resistors R33-R38 7 connected to the +15 volt side of a power supply which is center tapped 8 to ground and has filter capacitors C8 and C9 coupled to ~15 volts and 9 capacitors C10 and Cll coupled to -15 volts. -15 volts is fed out through the terminal block 103 as the minus side of the respective drive 11 line pairs carrying the signals corresponding to the commutation 12 equations. It wlll be understood that, although the transistor switches -13 of Fig. 5 are shown as single transistors for simplicity and ease of 14 understanding, these transistors are in fact, as indicated above, more complex power switching circuits to which dual drive lines are connected 16 for control~ The other one of each pair of the output control lines 17 from the terminal block 103 in Fig. 6 is coupled to the o~tput terminal
18 ~collector) of a corresponding one of the inverting driver transistors
19 84-89. In series with each such output line is a correspondins one of a plurality of light emitting diodes (LEDs) 97-102 which are included in 21 the circuit for test purposes.
~2 Another switching stage, similar to the stages ~lA-UlD but not 23 shown in the circuitry of Fig. 6, is included in the level dependent 24 ¦ logic stage 34 of the cir`cuit of Fig. 4. This switching circuit is 25 ¦ biased to change state about the zero level of applied control voltage.
26 ¦ This provides the NORMAL output and its complement (+1/-1) which is 27 ¦ applied as the multiplier for the feedback current in the stage 42 so 28 ¦ that the feedback current can be made negative when counterclockwise 29 operation is desired~ The relationship of the NO~MAL switch relative to 30 ¦ the remaining switches in the level dependent logic is indicated in Fig.
31¦ 7. The changes of s.ate of the NORM~L switch occur as the applied error 32 !
I ~16-11 !
Il I
~95~ S8 1 signal ch nges pola-it~ ard a- the c~ntrol circuitry is progressing fr~m 2 ¦ counterclockwise operation to clockwise operation (considering the 3 ¦ applied control voltage progressing in the positive direction). With 4 ¦ the exception of the addition of the NORMAL switch states shown in Fig.
5 ¦ 7, Fig. 7 corresponds to Fig. 1 and the explanation is comparable 61 thereto.
7 In accordance with an aspect of the present invention, the level 8 dependent logic, as just 2escribed, allows for the proper switching 9 transistors to be turned on without having to have prior knowled~e as to 10 which transistors are needed because of variatio~s in the operatin~
11 modes. As described above, the current sensor 41 receives the feedback 12 current signals ISl and IS2 developed across the sensing resistors as 13 shown in Fig. 5 and provides an output signal correspondin~ to the 14 larger of the two. Since the currents through RSl and RS2 only give the 15 absolute value of the current flow in the motor, the resultant current 16 feedback signal must be inverted if the motor is being driven in the 17 counterclockwise direction. The current feedback is inverted in the 18 stage 42 (Fig. 4~ depending on which transistor switches are being 19 turned on next. The feedback sign may not agree with the current in the
~2 Another switching stage, similar to the stages ~lA-UlD but not 23 shown in the circuitry of Fig. 6, is included in the level dependent 24 ¦ logic stage 34 of the cir`cuit of Fig. 4. This switching circuit is 25 ¦ biased to change state about the zero level of applied control voltage.
26 ¦ This provides the NORMAL output and its complement (+1/-1) which is 27 ¦ applied as the multiplier for the feedback current in the stage 42 so 28 ¦ that the feedback current can be made negative when counterclockwise 29 operation is desired~ The relationship of the NO~MAL switch relative to 30 ¦ the remaining switches in the level dependent logic is indicated in Fig.
31¦ 7. The changes of s.ate of the NORM~L switch occur as the applied error 32 !
I ~16-11 !
Il I
~95~ S8 1 signal ch nges pola-it~ ard a- the c~ntrol circuitry is progressing fr~m 2 ¦ counterclockwise operation to clockwise operation (considering the 3 ¦ applied control voltage progressing in the positive direction). With 4 ¦ the exception of the addition of the NORMAL switch states shown in Fig.
5 ¦ 7, Fig. 7 corresponds to Fig. 1 and the explanation is comparable 61 thereto.
7 In accordance with an aspect of the present invention, the level 8 dependent logic, as just 2escribed, allows for the proper switching 9 transistors to be turned on without having to have prior knowled~e as to 10 which transistors are needed because of variatio~s in the operatin~
11 modes. As described above, the current sensor 41 receives the feedback 12 current signals ISl and IS2 developed across the sensing resistors as 13 shown in Fig. 5 and provides an output signal correspondin~ to the 14 larger of the two. Since the currents through RSl and RS2 only give the 15 absolute value of the current flow in the motor, the resultant current 16 feedback signal must be inverted if the motor is being driven in the 17 counterclockwise direction. The current feedback is inverted in the 18 stage 42 (Fig. 4~ depending on which transistor switches are being 19 turned on next. The feedback sign may not agree with the current in the
20 actual motor winding, but this is in accordance with the four-guadrant
21 control realiæed ~y the control circuit of the present invention.
22 The "highest wins" stage for comparing the absolute value
23 signals developed across RSi and RS2 of Fig. 5 may be eliminated and a
24 simple summing stage may be used if the circuit modification shown in
25 Fig. 8 is employed. Fig. 8 represents a portion of the circuit of Fig.
26¦ 5 showing the diodes 62, 64 and 66 coupled in association with switching 271 transistors A~, B+ and C~, si~ilar to the arrangement in Fig. 5 except ~¦ that the common connection to the cathodes o the diodes 62, 64, 66 is 29 connected in series with a resistor RS2' to the +VL terminal 52. With 30 Ithe switching circuit connected as shown in FigA 8, the resistor RS2 of 31 ~Fig. ~ would be eliminated, and the diode feedback signal IS2 is derived ;
l . . .
1 from the resistor RS2'. With this circuit, a simple summing stage can 2 be provided for comparing ISl and IS2. The method of driving the motor 3 is independent of the method of making the c~rrent measurement for the 4 feedback signalO
~ The use of control circuits in accordance with the present 6 invention as shown and described hereinabove for controlling the power 7 applied to an actuator motor i5 particularly advantageous in electrical 8 actuating syste~s for controlling the control surfaces and certain other g elements which are now controlled by hydra~lic systems in aircraft.
10¦ Indeed, control syst~ms in accordance with the invention may be used in 11¦ many applications where r~liable operation, simplification of hardware 12~ and reduction of weight are desireda It will be understood that for -13 ~ fail-safe reliability, as in an aircraft control system or the like, 14 ¦ dual channel redundancy is provided by duplicating the actuator motors, ~5 ¦ control circuits, and the like.
16¦ Although there have been described above specific arrangement~
17¦ of a multi-quadrant brushless DC motor drive in accordance with the 18¦ invention for the purpose of illustrating the manner in which the 19¦ invention may be used to advantage, it will be.appreciated that the 201 invention is not limited thereto. Accordingly, any and all 21 modifications, variations or equivalent arrang~ments which may occ~r to 22 ~ those skilled in the art should be considered to be within the scope of 23 ¦ the invention as defined in the annexed claims.
29 . .
l . . .
1 from the resistor RS2'. With this circuit, a simple summing stage can 2 be provided for comparing ISl and IS2. The method of driving the motor 3 is independent of the method of making the c~rrent measurement for the 4 feedback signalO
~ The use of control circuits in accordance with the present 6 invention as shown and described hereinabove for controlling the power 7 applied to an actuator motor i5 particularly advantageous in electrical 8 actuating syste~s for controlling the control surfaces and certain other g elements which are now controlled by hydra~lic systems in aircraft.
10¦ Indeed, control syst~ms in accordance with the invention may be used in 11¦ many applications where r~liable operation, simplification of hardware 12~ and reduction of weight are desireda It will be understood that for -13 ~ fail-safe reliability, as in an aircraft control system or the like, 14 ¦ dual channel redundancy is provided by duplicating the actuator motors, ~5 ¦ control circuits, and the like.
16¦ Although there have been described above specific arrangement~
17¦ of a multi-quadrant brushless DC motor drive in accordance with the 18¦ invention for the purpose of illustrating the manner in which the 19¦ invention may be used to advantage, it will be.appreciated that the 201 invention is not limited thereto. Accordingly, any and all 21 modifications, variations or equivalent arrang~ments which may occ~r to 22 ~ those skilled in the art should be considered to be within the scope of 23 ¦ the invention as defined in the annexed claims.
29 . .
Claims (37)
OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. Apparatus for controlling the current and torque of a brushless DC motor comprising:
a brushless DC motor having a plurality of windings;
a plurality of solid state power switches respectively connected to said windings for selectively applying power to develop current therein;
means for determining the states and order of switching between states of said solid state switches in response to a current error signal;
means for deriving a current feedback signal indicative of current in the motor windings; and means for combining the current feedback signal with an applied current command signal to develop the current error signal.
a brushless DC motor having a plurality of windings;
a plurality of solid state power switches respectively connected to said windings for selectively applying power to develop current therein;
means for determining the states and order of switching between states of said solid state switches in response to a current error signal;
means for deriving a current feedback signal indicative of current in the motor windings; and means for combining the current feedback signal with an applied current command signal to develop the current error signal.
2. The apparatus of claim 1 wherein the motor has at least three windings and further including rotor position sensing means for providing phase signals for application to said determining means.
3. The apparatus of claim 2 wherein the determining means comprises a plurality of level dependent switches coupled to be controlled by said current error signal, each of said switches being switchable between on and off states and exhibiting a hysteresis effect relative to the directions of change of state in response to the control voltage thereof.
4. The apparatus of claim 3 wherein each of said level dependent switches is biased to change state at a different control voltage level than the other switches.
5. The apparatus of claim 4 wherein the level dependent switches are arranged by pairs, one pair being for controlling motor drive in one direction of rotation, the other pair being for controlling motor drive in an opposite direction of rotation.
6. The apparatus of claim 5 wherein the switches for controlling motor drive in a first direction are biased to change state at negative discrete levels of control voltage while the pair of switches for controlling motor drive in the opposite direction of rotation are biased to change state for positive discrete levels of control voltage.
7. The apparatus of claim 6 further including a single level dependent switch biased to change state in the vicinity of zero control voltage to develop a signal for affecting the sign of the feedback current signal.
8. The apparatus of claim 3 or claim 6 wherein the determining means further includes a plurality of commutation logic means for combining signals from the respective pairs of level dependent switches and the rotor position sensing means to coordinate level dependent switch output signals with the phase signals.
9. The apparatus of claim 8 wherein the commutation logic means comprise a plurality of interconnected gates coupled to receive respective inputs from the level dependent switches and the rotor position sensing means for developing control signals to control the power switches in accordance with a predetermined set of commutation equations.
10. The apparatus of claim 9 wherein said power switches are arranged by pairs, one pair for each motor winding, and are operable to selectively apply positive and negative voltage to the corresponding windings.
11. The apparatus of claim 10 wherein said power switches correspond to the designations A+, A-, B+, B-, C+ and C-, depending on their connections to corresponding windings and to positive and negative sides of the power supply, and wherein the selected commutation equations correspond to the following:
A+ = ? 03 CWP + 01 ? CCWP
B+ = 02 ? CWP + ? 03 CCWP
C+ = 01 ? CWP + ? 02 CCWP
A- = 01 ? CWM + ? 03 CCWM
B- = ? 03 CWM + 02 ? CCWM
C- = ? 02 CWM + 01 ? CCWM
A+ = ? 03 CWP + 01 ? CCWP
B+ = 02 ? CWP + ? 03 CCWP
C+ = 01 ? CWP + ? 02 CCWP
A- = 01 ? CWM + ? 03 CCWM
B- = ? 03 CWM + 02 ? CCWM
C- = ? 02 CWM + 01 ? CCWM
12. The apparatus of claim 1 wherein each of the power switches has a diode connected in parallel with it and poled to conduct current in a direction opposite to the normal direction of current through the associated switch.
13. The apparatus of claim 12 further including means for sensing the magnitude of current in the motor windings,
14. The apparatus of claim 13 wherein said current sensing means comprises a first resistor connected in series with the power switches coupled between the windings and one side of the power supply and a second resistor connected in series between said one side of the power supply and the diodes associated with said switches.
15. The apparatus of claim 14 further including means for comparing voltages developed across said first and said second resistors to develop a current feedback signal.
16. The apparatus of claim 15 further including means for controlling the sign of said current feedback signal in accordance with the commanded direction of rotation of the motor.
17. The apparatus of claim 16 further including filter means connected in the feedback current loop, said filter means being selected to exhibit a band limiting characteristic having an upper roll-off frequency of approximately 314,000 radians per second.
18. The apparatus of claim 16 further including a single level dependent switch biased to change state when the control voltage applied thereto changes polarity.
19. The apparatus of claim 18 further including means for multiplying the derived current feedback signal with an output signal from said single level dependent logic switch in order to control the sign of the feedback current signal in accordance with the level of the error signal applied to the level dependent switches.
20. The apparatus of claim 17 further including a dynamic compensation means coupled between said signal combining means and the level dependent logic means for integrating and filtering the current error signal to develop the control signal for the level dependent logic means.
21. The apparatus of claim 20 wherein said dynamic compensation means includes filtering means for limiting the bandwith to an upper limit having a roll-off frequency of approximately 199,000 radians per second.
22. The apparatus of claim 21 wherein said dynamic compensation means further includes amplifying means having a selected gain which, together with the hysteresis characteristic of the level dependent switches, is effective to develop pulse width modulation control of the power switches.
23. The apparatus of claim 20 further including additional dynamic compensation means for processing a current command signal prior to combination thereof with the current feedback signal, said additional dynamic compensation means including a current limiting stage for limiting the command signal within a selected range corresponding to the range of permissible current levels in the motor windings.
24. The apparatus of claim 23 wherein the additional dynamic compensation means includes a band limiting filter having an upper roll-off frequency of approximately 15,000 radians per second.
25. The apparatus of claim 23 further including means for receiving a position command signal and converting said signal to a corresponding current command signal for application to said additional dynamic compensation means.
26. The apparatus of claim 25 comprising an actuator connected to be driven by the motor and a feedback loop including an actuator position sensor for providing a position feedback signal to be compared with said position command signal in developing a position error signal.
27. The apparatus of claim 26 further including a rate feedback loop coupled to develop a rate feedback signal indicative of motor speed and means for comparing the position error signal and the rate feedback signal to develop the current command signal.
28. The apparatus of claim 12 wherein said current sensing means comprises first means for sensing the current in the power switches connected to one side of the power supply and second sensing means for sensing the current in the diodes associated with switches connected to the other side of the power supply, and means for summing said two current signals.
29. A method of controlling the current and torque of a brushless DC motor in response to a current command signal, which motor has a plurality of windings and associated power switches for applying power to the windings, comprising the steps of:
applying a current error signal to a plurality of level dependent switches to selectively control the conduction states thereof, each of said switches being biased to develop a change of conduction state at a unique level of control voltage;
applying signals developed from the outputs of the level dependent switches to control the states of the power switches;
deriving a current feedback signal indicative of motor current;
and combining the current feedback signal with an applied current command signal to develop the error signal.
applying a current error signal to a plurality of level dependent switches to selectively control the conduction states thereof, each of said switches being biased to develop a change of conduction state at a unique level of control voltage;
applying signals developed from the outputs of the level dependent switches to control the states of the power switches;
deriving a current feedback signal indicative of motor current;
and combining the current feedback signal with an applied current command signal to develop the error signal.
30. The method of claim 29 wherein the motor has at least three windings, further including the step of combining the output signals of the level dependent switches with signals indicative of motor phase in accordance with a predetermined commutation logic arrangement, and applying the resulting signals to the power switches.
31. The method of claim 29 wherein the combining step comprises applying the output signals from the level dependent switches to a selectively intercoupled plurality of NOR gates to develop a plurality of commutation equations to determine the actuation of the power switches.
32. The method of claim 30 wherein the commutation equations correspond to the following:
A+ = 01 03 CWP + 01 03 CCWP
B+ = 02 03 CWP + 02 03 CCWP
C+ = 01 02 CWP + 01 02 CCWP
A- = 01 03 CWM + 01 03 CCWM
B- = 02 03 CWM + 02 03 CCWM
C- = 01 02 CWM + 01 02 CCWM
wherein A+, B+, C-, A-, B- and C- designate specific power switches; 01, 02 and 03 designate specific phase signals corresponding to rotor position; and CWP, CWM, CCWP and CCWM designate signals from the level dependent switches.
A+ = 01 03 CWP + 01 03 CCWP
B+ = 02 03 CWP + 02 03 CCWP
C+ = 01 02 CWP + 01 02 CCWP
A- = 01 03 CWM + 01 03 CCWM
B- = 02 03 CWM + 02 03 CCWM
C- = 01 02 CWM + 01 02 CCWM
wherein A+, B+, C-, A-, B- and C- designate specific power switches; 01, 02 and 03 designate specific phase signals corresponding to rotor position; and CWP, CWM, CCWP and CCWM designate signals from the level dependent switches.
33. The method of claim 29 wherein the step of deriving a current feedback signal comprises sensing the current in the power switches and the current in shunt current paths individually associated therewith, and combining said two currents to develop a signal corresponding to the higher absolute value of said currents.
34. The method of claim 33 wherein the step of deriving the current feedback signal further includes establishing the sign of said absolute value signal in accordance with a signal from a level dependent switch which is biased to change state about the zero level of control voltage.
35. The method of claim 33 further comprising the step of changing the sign of the feedback current signal when it is desired to reverse the direction of acceleration of the motor.
37. The method of claim 29 further including the step of limiting the range of the current command signal in accordance with the limits of permissible current in the motor,
37. The method of claim 29 further including the step of developing the current command signal in response to a position command signal by modifying the position command signal in accordance with position feedback and rate feedback signals derived from the motor.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US06/354,963 US4494051A (en) | 1982-03-05 | 1982-03-05 | Multi-quadrant brushless DC motor drive |
| US354,963 | 1982-03-05 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1199058A true CA1199058A (en) | 1986-01-07 |
Family
ID=23395645
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000422263A Expired CA1199058A (en) | 1982-03-05 | 1983-02-23 | Multi-quadrant brushless dc motor drive |
Country Status (5)
| Country | Link |
|---|---|
| US (1) | US4494051A (en) |
| EP (1) | EP0089150B1 (en) |
| JP (1) | JPS58195485A (en) |
| CA (1) | CA1199058A (en) |
| DE (1) | DE3366182D1 (en) |
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| FR2590420B1 (en) * | 1985-11-21 | 1994-04-08 | Valeo | SUPPLY CURRENT DEVICE FOR A DIRECT CURRENT ELECTRIC MOTOR, AND ELECTRIC MOTOR EQUIPPED WITH SUCH A DEVICE |
| US4751438A (en) * | 1985-12-18 | 1988-06-14 | Sundstrand Corporation | Brushless DC motor control |
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-
1982
- 1982-03-05 US US06/354,963 patent/US4494051A/en not_active Expired - Fee Related
-
1983
- 1983-02-23 CA CA000422263A patent/CA1199058A/en not_active Expired
- 1983-03-03 EP EP83301143A patent/EP0089150B1/en not_active Expired
- 1983-03-03 DE DE8383301143T patent/DE3366182D1/en not_active Expired
- 1983-03-04 JP JP58034731A patent/JPS58195485A/en active Pending
Also Published As
| Publication number | Publication date |
|---|---|
| EP0089150A1 (en) | 1983-09-21 |
| DE3366182D1 (en) | 1986-10-23 |
| US4494051A (en) | 1985-01-15 |
| JPS58195485A (en) | 1983-11-14 |
| EP0089150B1 (en) | 1986-09-17 |
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