CA2104252A1 - Lamp ballast circuit - Google Patents
Lamp ballast circuitInfo
- Publication number
- CA2104252A1 CA2104252A1 CA002104252A CA2104252A CA2104252A1 CA 2104252 A1 CA2104252 A1 CA 2104252A1 CA 002104252 A CA002104252 A CA 002104252A CA 2104252 A CA2104252 A CA 2104252A CA 2104252 A1 CA2104252 A1 CA 2104252A1
- Authority
- CA
- Canada
- Prior art keywords
- capacitor
- circuit
- lamp
- frequency
- voltage
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Abandoned
Links
Classifications
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
- H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
- H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
- H05B41/2825—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage
- H05B41/2828—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices by means of a bridge converter in the final stage using control circuits for the switching elements
-
- H—ELECTRICITY
- H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
- H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
- H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
- H05B41/14—Circuit arrangements
- H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
- H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
- H05B41/282—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices
- H05B41/285—Arrangements for protecting lamps or circuits against abnormal operating conditions
- H05B41/2851—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
- H05B41/2856—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against internal abnormal circuit conditions
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10—TECHNICAL SUBJECTS COVERED BY FORMER USPC
- Y10S—TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10S315/00—Electric lamp and discharge devices: systems
- Y10S315/05—Starting and operating circuit for fluorescent lamp
-
- Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10—TECHNICAL SUBJECTS COVERED BY FORMER USPC
- Y10S—TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
- Y10S315/00—Electric lamp and discharge devices: systems
- Y10S315/07—Starting and control circuits for gas discharge lamp using transistors
Abstract
ABSTRACT:
"Lamp ballast circuit"
A ballast circuit having a series inductor (L7) and capacitor (C10) in which the lamp load (LL) is connected in parallel with the capacitor. During pre-ignition of the lamp load, the driving signal supplied by an inventor generating a substantially rectangular signal includes a fundamental frequency f1. The resonant frequency f0 of the series connected L-C circuit is at least 2 times greater than the fundamental frequency f1 but less than the third harmonic of the driving signal.
Fig. 3.
"Lamp ballast circuit"
A ballast circuit having a series inductor (L7) and capacitor (C10) in which the lamp load (LL) is connected in parallel with the capacitor. During pre-ignition of the lamp load, the driving signal supplied by an inventor generating a substantially rectangular signal includes a fundamental frequency f1. The resonant frequency f0 of the series connected L-C circuit is at least 2 times greater than the fundamental frequency f1 but less than the third harmonic of the driving signal.
Fig. 3.
Description
PHA 21.745 1 08.07.1993 "LalTlp ballast circuit"
BACKGROUND OF THE INVENTION
This invention relates to a ballast circuit for generating a substantially rectangular driving signal sufficient to ignite a lamp load, comprising:
inductor means;
capacitor means serially connected to said inductor means; and generating means for applying a generated signal to said serially connected inductor means and capacitor means, said generated signal having at least a fundamental frequency fi;
the inductor means and capacitor means having a Nsonant frequency fO.
Inductor means are to understand to be means adapted to e~hibit the 10 properties of an inductor. Capacitor means are to understand to be means adapted to exhibit the properties of a capacitor.
Conventionally the lamp load is connected across the capacitor. In a known circuitry the series L-C circuit operates during pre-ignition of the lamp load substantially at its resonant frequency. That is, the driving signal applied to the series 15 L-C circuit is at or near the resonant frequency of the series L-C circuit. In this way a sufficiently high pre-ignition voltage is applied across the lamp load for ignition of the latter.
The lamp load, typically of a fluorescent type, following ignition, achieves a substantia11y steady-state sinusoidal current flow therethrough by reducing the 20 driving signal frequency well below the resonant frequency of the series L-C circuit. In determining when to switch from the resonant frequency to a different steady-state operating frequency, feedback circuitry is required in the known ballast circuit for sensing lamp ignition.
A sufficiently high voltage during pre-ignition of the la np and sinusoidal 25 lamp current following ignition (i.e. steady state operation), is commonly provided by a bridge inverter. Both full bridge and half-bridge inverters are known in the ballast circuit art. The (half)-bridge inverter includes switching to control the f~equency of the ::
PHA21.745 ~ ~ 42.32 08.07.1993 driving signal applied to the series L-C circuit. Control circuitry, responsive to the feedback circuitry, is required for controlling the speed at which the switching takes place.
Known lamp ballast circuits, as described above, suffer from several S drawbacks. For example, known lamp ballast circuits require generating two different frequencies, that is, the resonant frequency during pre-ignition of the lamp load and a different therefrom a steady-state operating frequency. Such ballast circuits also require sensing circuitry to determine when to switch from the resonant frequency to the steady state operating frequency.
It is particularly undesirable to operate at or near the resonant frequency of the series L-C circuit before lamp ignition inasmuch as unsafe, high voltages and current levels can occur (i.e. above the maximum ratings of one or more ballast circuit components). By operating below resonance during pre-ignition of the lamp load, capacitive switching of the inverter can easily occur producing high switching losses.
15 Additional circuitry is therefore required to prevent the inverter from operating below the series L-C circuit resonant frequency during pre-ignition of the lamp load.
The inductance of inductor L is normally determined based on the desired lamp current during steady state conditions. The capacitance of capacitor C is thereafter chosen so as to provide a resonant condition (typically between 20-50 kHz for a 20 fluorescent lamp). Generally, the capacitance of capacitor C is between about 5 to 10 nanofarads with the additional high voltage capability leading to a relatively costly capacitor requiring a relatively large space on a printed circuit board.
Accordingly, it is desirable to provide a lamp ballast circuit having a safe open circuit (i.e., pre-ignition) voltage and current level, with relatively low switching 25 losses. The improved lamp ballast circuit should not need a driving signal at more than one frequency, this frequency being well below resonance of the series L-C circuit. It is also desirable that the improved lamp ballast circuit permit use of a relatively less e~pensive, smaller capacitor in order to lower the lamp ballast manufacturing cost and to reduce the reactive current flowing through the capacitor after lamp ignition thus 30 lowering circuit power loss.
SU~ARY OF THE INVENTION
In accordance with the invention, a ballast circuit for generating a driving PHA 21.745 32 ~ ~ 4 2 .~ ~ 08.07.1993 signal sufficient to ignite a lam~load as mentioned in the prearnble is characterized in that for the fundamental ~requency ft and the resonant frequency fO it holds:
nf, < fO c (n+l)f, with n = an even integer.
By operating in these regions during pre-ignition, safe voltage and current 5 levels will be maintained. A single drive frequency results in safe non-resonant operation before lamp ignition as well as correct lamp current after ignition. Feedback circuitry for sensing ignition of the larnp load for switching to a diffe~ent steady-state lamp operating frequency need not be provided. By eliminating the need to operate at the resonant frequency of the series connected L-C circuit during pre-ignition of the 10 lamp load, the value and resulting si~e of the capacitor can be chosen far smaller than norrnally used in a conventional series connected L-C circuit in a known ballast circuit.
In accordance with a feature of the invention, the generated signal which is a train of square waves, is generated preferably by a half-bridge or full bridge inverter. In yet another feature of the invention, the resonant frequency of the series 15 connected L-C circuit is less than the third harmonic ~requency of the generated square wave drive thereby avoiding unsafe third harmonic voltages and current levels duAng pre-ignition of the larnp load. Substantially the sarne generated signal frequency is used during pre-ignition and steady-state operation of the lamp load.
Accordingly, it is an object invention to provide an improved ballast 20 circuit in which the unloaded, open circuit voltage and current levels are within the operating range of the ballast circuit components.
It is another object of the invention to provide an impToved ballast circuit in which the same inverter driving signal can be used during pre-ignition and steady-state operation of the larnp load.
It is a further object of the invention to provide an improved ballast circuit in which less costly components can be used to lower the manufacturing cost of the ballast.
It is still another object of the invention to provide an improved ballast circuit which eliminates the need for feedback circuitry for sensing lamp ignition for 30 changing the inverter frequency.
It is still a further object of the invention to provide an improved ballast circuit in which the inverter driving signal frequency is substantially less than the resonant frequency of a series connected L-C output circuit during pre-ignition of the PHA 21.745 ~ ) 2 0~.07.1993 lamp load.
The invention accordingly compnses several steps in a relation of one or more of such steps with respect to each of the others, and the device embodying features of construction, a combination of elements and arrangement of parts which are S adapted to effect such steps, all is exemplified in the following detailed disclosure and the scope of the invention will be indicated in the claims.
BRlEF DESCRIPIION OF DRAWINGS
For a fuller understanding of the invention, reference is made to the 10 following description taken in connection with the accompanying drawings, in which:
Fig. 1 is a circuit diagram of a baLast output circuit in accordance with the present invention;
Figs. 2(a), 2(b) and 2(c) are timing diagrams of an inverter substantially rectangular output voltage, output current at its fundarnental frequency and output 15 current at its third harmonic, respectively in the circuit according to Fig. l;
Fig. 3 is a schematic diagram of a ballast circuit in accordance with the invention;
Figs. 4(a), 4(b), 4(c) and 4(d) are timing diagr~ns of signals produced within the ballast circuit of Fig. 3 during pre-ignition and steady-state operation of the 20 lamp load; and Fig. 5 is a diagram of simulation of current in circuit of Fig. 1 as function of the ratio fundamental frequency and resonant frequency.
DETAILED DESCRIlYrION OF l~ PREFER~ED EMBODIMENT
The figures shown herein illustrate a preferred embodiment of the invention. Those elements/components shown in more than one figure of the drawings have been identified by like reference numerals/letters and are of similar construction and operation.
Referring now to Figs. 1, 2(a), 2(b) and 2(c), a ballast circuit having a 30 ballast output circuit 10 includes an inductor L and a capacitor C seAally connected across the output of a square wave generator 13. Square wave generator 13 is preferably, but not limited to, a bridge inverter generating a substantially square wave of voltage +E (i.e. the inverter output voltage). A larnp load 16 is connected across PHA 21.745 2~ 2 ~ 2 08.07.1993 capacitor C through a switch SW. A current I flowing through inductor L includes a fun,damental frequency component I~, and a third harmonic component of the fundamental frequency I3n. Other currents at higher odd harmonics are present but are significantly smaller. For the sake of simplicity in calculations with respect to the 5 preferred embodiment as described hereafter only terms concerning the fundamental frf~uency f, and the 3rd harmonic are taken into account.
In accordance with the Fourier transform square wave voltage 13 contains a sinusoidal wave at a fundarnental frequency fi and odd harmonics of the fundamental frequency including a sinusoidal wave at a third harmonic 3f,. The amplitude of third 10 harmonic component fi of voltage E is one third the amplitude of fundamental frequency component fi of voltage E.
To achieve low switching losses within square wave generator 13 during pre-ignition of lamp load 16 (generally at trailing edges Er Of voltage E), curTent I is preferably inductive (i.e., current lagging drive voltage) rather than capacitive (i.e.
15 current leading drive voltage) during the voltage transitions of voltage E. Accordingly, the sum of fundamental frequency current component In and third harmonic-currentcomponent I3n is inductive wherein Ilf and I3n are the capacitive and inductive components of I, respectively. To achieve an overall inductive current I, an impedance Z of circuit 10 as viewed from square wave generator 13 requires that the inductive 20 impedance at the third harmonic Z3n be less than one third the capacitive impedance at the fundamental frequency Zn~ In other words, third harmonic component current I3n is greater than fundamental frequency component In. This relationship is illustrated in Figs. 2(b) and 2(c) wherein an arnplitude P represents the peak value of fundamental frequency current component I,l but is less than the peak va1ue of third harmonic 25 current component I3n. In this way the sum of In and I3n remains inductive at the voltage transitions of voltage E.
Lamp load 16 prior to ignition (i.e. during pre-ignition) appears as an open circuit. This open circuit condition is represented by switch SW in an open state (turned OFF). Following ignition, lasnp load 16 is in its steady-state mode of operation and is 30 represented by switch SW being turned ON such that larnp load 16 is connected in pa~llel with capacitor C.
Impedance Z3n~ which must be less than one third impedance Zn during pre-ignition of lamp load 16, is therefore based on switch SW in its open state (i.e., PH~ 21.745 6 ~ 08.07.1993 turned OFF). This condition can be expressed as follows:
IZnl > ¦ 3 Z3nl (eq. 1) That is, ~2~f,xL-1/(2~f,xC)1 > 3 l61rfl xL-1/(6~f~xC)1 (eq. 2) S Since impedance Z is capacitive at fundamental frequency fl and inductive at the third harmonic 3fi, 1/(2~rfixC)-2~rfixL > 18~rfi~cL - l/(2~,xC) That is, 1/(2~fixC)>5(2~fixL) (eq. 3) Eq. 3 can be rewritten as follows:
l/~LC > ~5 2~fi (e~. 4) A resonant frequency f0 of cir~uit l0 during pre-ignition (i.e., with switch 15 SW open) can be defined as follows:
l/~fLC = 27 f0 (eq. 5) Substituting the value of l/~LC defined by eq. 4 for the value of l/~LC in eq. Sresu1ts in 2~fo > ~/5 27fi (eq. 6) Accordingly, resonant frequency f0 can be e~pressed as follows:
fo > ~5 fi (eq. 7) In otha words, third harmonic inductive current component I3n is greater than fundarnental f~equency capacitive current component In when resonant frequency f0 is greater than ~5 times the fundamental frequency of voltage E.
To ensure that unsafe voltages and currents present at resonant frequency f cannot occur, resonant frequency f0 also should be less than third harmonic frequency 3fi of voltage E. Therefore, the values of inductor L and capacitor C should be chosen such th~t:
~5fi < fo ~ 3f~ (eq- 8) By designing baLlast circuit 10 such that resonant frequency f0 is within the range of frequencies defined by eq. 8, the unsafe voltages and currents which occur at resonant frequency f0 during pre ignition of larnp load 16 are avoided and total current delivered by square wave generator 13 remains inductive. There is no need to vary the .
PHA 21.745 .~ ; 2 ~ 2 OB.07.1993 frequency of voltage E between resonant frequency fO during pre-ignition of lamp load 16 and a different frequency immediately thereafter as in conventional ballast circuitry.
Feedback circuitry designed to sense ignition of lamp load 16 for determining when to vary the frequency of voltage E from resonant frequency fO to a different operating S frequency can be eliminated. In accordance with the invention, a safer, simpler circuit is plrovided by maintaining resonant frequency fO within the boundaries defined by eq. 8.
Due to the fact that the calculation as shown has only taken into account the fundamental frequency f, and its 3rd harmonic 3f" the lower value of the range for chosing the resonant frequency fO is ~5 times f,. However, when taken into account the 10 existence of higher harmonics this value reaches the limit 2.
The result of a simulation in which at least the first 25 harmonics are taken into account is shown in figure 5.
In Fig. 5 the depicted cu~ve disp~lays the total current I,Yo in the circuit of Fig. 1 at the moment the voltage switches from -E to +E of the generator 13 as 15 function of ratio of the fundamental frequency fi and the resonant frequency fO. The circuit operates in the inductive mode in all those regions that the current I~o is lagging to the voltage, thus is negative. From the Fig. S it is clear that these regions fulfil the relation nfl < fO < (n+l)fi with n = an even integer.
A ballast circuit 20 in accordance with the invention is shown in Fig. 3.
An input voltage of 277 volts, 60 hertz is supplied to an electromagnetic interference (EMI) suppression filter 23. Filter 23 filters high frequency components inputted thereto lowering conducted and radiated EMI. The output of filter 20 provided at a pair of terminals 24 and 25is supplied to a full wave rectifier 30 which includes diodes Dl, D2, 25 D3 and D". The anode of diode Dl and cathode of diode D2 are connected to terminal 24. The anode of diode D3 and cathode of diode D4 are connected to terminal 25. The output of rectifier 30 (i.e. rectified a.c. signal) at a pair of output terminals 31 and 32is supplied to a boost converter 40. The cathodes of diodes D, and D3 are connected to terminal 31. The cathodes of diodes D2 and D~ are connected to terminal 32.
Converter 40 boosts the magnitude of the rectified A.C. signal supplied by rectifier 30 and produces at a pair of output terminals 41 and 42 a regulated D.C.
voltage supply. Boost converter 40 includes a choke L3, a diode D5 the anode of which is connected to one end of choke L3. The other end of choke L3 is coMected to output .
pHA21.745 8 2~a~25~ 0~.07.1993 terrninal 31 of rectifier 30. The output of boost converter 40 at output tenninals 41, 42 is applied across an electrolytic capacitor C~, one end of which is connected to the cathode of diode Ds~ A transistor (switch) Q, is connected to the junction between choke Ll and the anode of diode D5. The other ~nd of transistor Q, is connected to theS junction between the other end of capacitor Cll, output terrninal 32 of rectifier 30 and output terminal 42.
A preconditioner control SQ, which is powered by a D.C. supply voltage V, controls the switching duration and frequency of transistor Q,. Preconditioner control 50 is preferably~ but not limited to, a Motorola MC33261 Power Factor Controller10 Integrated Circuit. Transistor Q, is preferably a MOSFET, the gate of which is connected to preconditioner control 50. Rectifier 30 and boost converter 40, including preconditioner control 50, forrn a preconditioner 80 for ballast circuit 20. Output terrninals 41 and 42 of boost converter 40 serve as the output for preconditioner 80 across which a regulated D.C. voltage is produced.
A larnp drive 90, which is supplied with the regulated D.C. voltage outputted by preconditioner 80, includes a half bridge inverter having a le~el shifter 60 and a half-bridge drive 70. The half brldge inverter includes a pair of transistors Q6 and Q7, which serve as switches, a pair of capacitors C5 and C6 and a transformer T~. Half-bridge drive 70 produces a square wave driving signal to drive transistor Q7 and has a 20 S0-50 duty cycle. Level shifter 60 inverts the driving signal supplied to transistor Q7 for driving transistor Q6. The driving signals produced by level shifter 60 and half-bridge drive 70 are appro~imately 180 out of phase vith each other so as to prevent conduction of transistors Q6 and Q7 at the same time, respectively.
A source S of transistor Q6 and one end of level shifter 60 are connected to 25 output terminal 41 of boost converter 40. A drain D of transistor Q6 is connected to a terminal A. The o~er end of level shifter 60, one end of half-bridge drive 70 and a source S of transistor Q, are also are coMected to terminal A. The other end of half-bridge drive 70 and a drain D of transistor Q7 are connected to output terminal 42 of boost converter 40. Capacitor C5 iS connected at one end to output terminal 41. The 30 other end of capacitor C5 and one end of capacitor C6 are connected to a terminal B.
The other end of capacitor C6 is connected to output tersl~inal 42.
A primary winding Tp of transformer Tl is connected to terminals A and B. A secondary winding T9 is connected at one end to an inductor L7, the latter which PHA 21.745 9~ 2 08.07.1993 generally represents either the leakage inductance of transformer Tl or a discrete choke.
Connected to the other end of inductor L7 is one end of a capacitor C,0 and one end of a lamp load LL. Lamp load LL can include any combination of lasnps and is shown, but not limited to, the series combination of two fluorescent larnps LLI and LL2. The other S ends of capacitor ClO and la np load LL are connected to the other end of secondaly winding T,.
The turns ratio between primary winding Tp and secondary winding T, of transformer T, is Np/N,. Transformer T, electrically isolates lamp load LL from the output voltage produced by preconditioner 80 and provides sufficient open circuit 10 voltage during pre-ignition to ignite larnp load LL.
The inductance of inductor Lq is based on the desired current flow through laJnp load LL once the latter has ignited and is in its steady-state mode of operation.
The DC voltage a~ross each capacitor C5 and capacitor C6 is approximately half the output voltage of preconditioner 80.
lS The waveforms shown in Figs. 4(a), 4(b), 4(c) and 4(d) produced by ballast circuit 20 are based on turns ratio N,/Np of about 1.5, inductor Lq of approximately 4.3 millihenries, capacitor ClO of about 1.2 nanofarads and capacitors C3 and C4 of about 0.33 microfarads, nominally rated at 630 volts. Both lamp LLI and lamp LL2 are 40 watt low pressure mercury vapor tubular fluorescent lamps. l'he 20 fundamental frequency of the square wave produced by the half-bridge inverter is appro~imately 281~Iz. The resonant frequency of inductor L7 and capacitor ClO isapproximately 70kHz, that is, appro~imately 2.5 times fundamental frequency fi.
During pre-ignition of lamp load LL, the output of ~e half-bridge inverter, which is across terminals A-B, forms a substantially square wave voltage train.
25 Inductor Lq and capacitor C,0 form an L-C series connected circuit. During pre-ignition, lamp load LL appears as a substantially open circuit (i.e. no load condition) drawing substantially no power e~cpect for filament heating (assuming lamps LLI and LL~ are fluorescent lamps of, for e~ample, the rapid-start type).
Fig. 4(a) illustrates a voltage VAB~ that is, between terminals A and B.
30 Voltage VAI~ is square wave voltage train which is applied across primary winding Tp varying between appro~imately +240 volts and -240 volts during no load conditions.
Fig. 4(b) illustrates current Ir~U flowing through primary winding Tp du~ing no load conditions, that is, prior to ignition of larnp load LL and having a peal~ value of PHA 21.745 ~ r~ 2 08.07. 1993 approximately i~ 400 milliamperes. Once lamp load LL is ignited and is in its steady-state operation, current IPRI flowing through primary winding Tp, as shown in Fig. 4(c), has a somewhat sinusoidal wave shape with a ~c value of approximately + 800 milliamperes. Capacitor Cl0 serves to smooth this somewhat sinusoidal current 5 waveforrn resulting in a substantially sinusoidal lamp cuIrent IL~MP as shown in FIG.
4(d) having a peak value of approximately + 380 milliamperes.
Inductor L7 serves as the lamp current ballasting element. Capacitor C,0, which is placed across lamp load LL, provides a more sinusoidal open circuit voltage and keeps total half bridge current inductive while also lowering higher harmonic 10 content of current flowing through larnp load LL. Inductor L, and capacitor Cl0 together form a series coMected L-C output circuit. The value for capacitor Cl0 is chosen such that safe open circuit operation is provided, that is, within the range of resonant frequencies defined by eq. 8. Accordingly, no additional circuits to protect lamp drive circuit 90 are required.
When ballast circuit 20 is first turned on, prior to the voltage being boosted by preconditioner 80, the input voltage of approximately 277 volts results in a square wave voltage of approximately 390 volts peak to peak being applied acrossprimary winding Tp of transformer Tl which is stepped up to approximately 570 volts pealc to peak across secondary winding T,. During this time the lamp cathodes are 20 heated. After approximately 0.5 seconds, preconditioner 80 turns ON resulting in a regulated D.C. voltage of approximately 480 volts across output terminals 41, 42 of boost converter 40 and a voltage of approximately 700 volts peak to peak across secondary winding T" the latter of which is sufficient for igniting larnp load LL. Once lamp load LL is ignited (i.e. during steady-state larnp operation), the lamp voltage (i.e.
25 voltage across larnp load LL) drops to approximately + 300 volts peak with the remainder of the secondary winding T9 output voltage across inductor Lq. The number of and connections between the lamps within larnp load LL can be varied as desired with the value of inductor L, being chosen so as to provide the desired lamp current IL"MP during steady-state operation of larnp load LL.
Referring again to Fig. 3, the rectified AC (i.e. pulsating DC) signal supplied to preconditioner 80 from diode bridge rectifier 30 is boosted in magnitude by choke L3 and diode D5 to charge capacitors C8, C5 and C6. In Fig. 3, capacitor C~ is separate from capacitors C5 and C6, capacitor C~ being a large electrolytic capacitor in PHA 21.745 11 2 ~ rr~ 2 08.07.1993 the range of 5 to 100 microfarads. Capacitors C5 and C6 are high frequency bridge capacitors. Since capacitor C~ is in parallel with the series combination of capacitors C5 andl C6, these three capacitors can be reconfigured as capacitors C5' and C6'.
Preconditioner 80 is an u~converter and boosts the rectified AC input S voltage as follows. When transistor Q6 (which serves as a switch) is closed, choke L3 is short circuited to ground. Current flows through choke L3. Transistor Ql is then opened (turned OFP). Choke L3 with transistor Q, open transfers stored energy through diode D5 into capacitor C~. The amount of energy transferred to capacitor C~ is based on the time during which transistor Ql is turned ON, that is, based on the frequency and 10 duration of the driving signal supplied to the gate of transistor Ql by the preconditioner control 50. Asynchronous operation of transistor Ql with respect to voltage V~, results.
Choke L3 operates in a discontinuous mode, that is, the current through choke L3 during each cycle is reduced to substantially zero before a new cycle is initiated. The frequency at which transistor Q, is turned ON and OFF is varied by 15 preconditioner control 50 so that the peak current through choke L3 is kept constant.
Transistors Q6 and Q7 have internal diodes (not shown). These diodes, which can either be internal or external to the transistors, permit inductive currents to flow through transistors Q6 and Q7 at the initial turn ON and turn OFF of transistors Q6 and Q7.
Preferably, capacitors C5 and C6 are electrolytic capacitors having a pair of 20 discharge resistors in parallel, respectively. Transformer T, is a leakage transformer, that is, having a leakage inductor of inductance LM which serves as the ballast for larnp load LL (i.e. to limit steady state current flow through the lamp load). Alternatively, when transformer Tl has little or no leakage inductance an external inductor of inductance L~, is required for ballast purposes.
Transformer T~ has a main secondary winding TM. A resonant capacitor Cl0 is in series with inductor L7 and reflects back to the primary winding of transformer Tl as a series LC combination across the nalf-bridge inverter.
AS now can be readily appreciated, by maintaining the fundarnental sinusoidal frequency fi well below resonant frequency f0 of the series L-C output 30 circuit, the undesirable and unsafe high voltages and current levels produced in conventional ballast circuits during p~e-ignition of lamp load LI, are avoided. More particularly, by choosing the values of inductor L7 and capacitor Cl0 such that their resonant frequency f0 is defined as described hereinbefore, the voltage level across ., . ~ ., :
, 2 ~ 2 PHA 21.74~ 12 08.07.1993 inductor L7 and capacitor C10 and current flow therethrough will be maintained at levels far below conventional ballast output circuits during pre-ignition of lamp load LL.
By not requiring the combination of inductor L, and capacitor ~,0 to be operated at its resonant frequency fO during pre-ignition of larnp load LL, the value of S capacitor C,0 can be significantly reduced. For example, conventional values for capacitor C10 range from about a nominal value of 6.8 nanofarads to about a nominal value of 9.2 nanofarads. In accordance with the invention, however, capacitor C10 can be reduced in value by approximately one-fourth to one-sixth (e.g. to approximately 1.2 nanofarads). Consequently, a far smaller, less expensive capacitor C,0 is required 10 reducing the manufacturing cost and space requirements of the baUast output circuit.
The reduced value of capacitor CtO results on top of this in substantially all current flowing through larnp load LL with relatively little current flowing through capacitor C2. Power re~quirements for the baUast circuit can be reduced and/or less costly wiring (higher resistance) can be used in the series connected L-C ballast output 15 circuit while maintaining the same power requirements as in a conventional ballast output circuit. In other words, a less costly and/or more efficient ballast with smaller space requirements is provided by the present invention.
Preferably, resonant frequency fO should range from approximately 2.3 to 2.6 times fundamental frequency f, of the square wave generated by the square wave 20 generator. ~ onsequently, stray inductances and the like which may be difficult to account for will not increase the overall inductance. Resonant frequency fO wiU not approach third harmonic frequency 3f,. Unsafe operation (i.e., resonant operation of the series L-C output circuit) of baUast circuit 20 is prevented.
GeneraUy, in calculating the inductance of inductor L, for determining 25 resonant frequency fO, the leakage inductance of transformer T~ or inductance of the discrete choke used for inductor L, is far greater than the stray inductance or other inductances within baUast circuit 20. Therefore, ~ a first order appro~cimation, the inducta~ce of inductor L, can be used without taking into account stray inductances and the like in determining the resonant frequency fO. For a tightly wound transformer T, in 30 which very little or an insufficient unount of leakage inductance e~cists, a discrete inductor will be required to serve as the baUasting element for lamp load LL (i.e., to control the larnp current I, ~,p).
As now can be readily appreciated, the generated voltage (i.e. voltage E of 2.~ ~
PHA 21.745 13 08.07.1993 Fig. 1 and voltage VA-~ of Fig. 4(a)) is at a frequency which is far less than the resonant frequency of the series connected L-C circuit and therefore provides safe open circuit (pre-ignition) voltages and current levels. The frequency of this generat~d signal need not be changed following pre-ignition since it is never at or near resonantS frequency fO of the series connected L-C circuit. Feedback circuitry for sensing ignition of lamp load LL for switching to a different steady-state lamp operating frequency need not be provided. By eliminated the need to operate at resonant frequency S of the series L-C circuit during pre-ignition of lamp load LL, the value and resulting size of the capacitor for the series connected L-C circuit ~an be far smaller than normally used in a 10 conventional series connectecl L-C circuit.
BACKGROUND OF THE INVENTION
This invention relates to a ballast circuit for generating a substantially rectangular driving signal sufficient to ignite a lamp load, comprising:
inductor means;
capacitor means serially connected to said inductor means; and generating means for applying a generated signal to said serially connected inductor means and capacitor means, said generated signal having at least a fundamental frequency fi;
the inductor means and capacitor means having a Nsonant frequency fO.
Inductor means are to understand to be means adapted to e~hibit the 10 properties of an inductor. Capacitor means are to understand to be means adapted to exhibit the properties of a capacitor.
Conventionally the lamp load is connected across the capacitor. In a known circuitry the series L-C circuit operates during pre-ignition of the lamp load substantially at its resonant frequency. That is, the driving signal applied to the series 15 L-C circuit is at or near the resonant frequency of the series L-C circuit. In this way a sufficiently high pre-ignition voltage is applied across the lamp load for ignition of the latter.
The lamp load, typically of a fluorescent type, following ignition, achieves a substantia11y steady-state sinusoidal current flow therethrough by reducing the 20 driving signal frequency well below the resonant frequency of the series L-C circuit. In determining when to switch from the resonant frequency to a different steady-state operating frequency, feedback circuitry is required in the known ballast circuit for sensing lamp ignition.
A sufficiently high voltage during pre-ignition of the la np and sinusoidal 25 lamp current following ignition (i.e. steady state operation), is commonly provided by a bridge inverter. Both full bridge and half-bridge inverters are known in the ballast circuit art. The (half)-bridge inverter includes switching to control the f~equency of the ::
PHA21.745 ~ ~ 42.32 08.07.1993 driving signal applied to the series L-C circuit. Control circuitry, responsive to the feedback circuitry, is required for controlling the speed at which the switching takes place.
Known lamp ballast circuits, as described above, suffer from several S drawbacks. For example, known lamp ballast circuits require generating two different frequencies, that is, the resonant frequency during pre-ignition of the lamp load and a different therefrom a steady-state operating frequency. Such ballast circuits also require sensing circuitry to determine when to switch from the resonant frequency to the steady state operating frequency.
It is particularly undesirable to operate at or near the resonant frequency of the series L-C circuit before lamp ignition inasmuch as unsafe, high voltages and current levels can occur (i.e. above the maximum ratings of one or more ballast circuit components). By operating below resonance during pre-ignition of the lamp load, capacitive switching of the inverter can easily occur producing high switching losses.
15 Additional circuitry is therefore required to prevent the inverter from operating below the series L-C circuit resonant frequency during pre-ignition of the lamp load.
The inductance of inductor L is normally determined based on the desired lamp current during steady state conditions. The capacitance of capacitor C is thereafter chosen so as to provide a resonant condition (typically between 20-50 kHz for a 20 fluorescent lamp). Generally, the capacitance of capacitor C is between about 5 to 10 nanofarads with the additional high voltage capability leading to a relatively costly capacitor requiring a relatively large space on a printed circuit board.
Accordingly, it is desirable to provide a lamp ballast circuit having a safe open circuit (i.e., pre-ignition) voltage and current level, with relatively low switching 25 losses. The improved lamp ballast circuit should not need a driving signal at more than one frequency, this frequency being well below resonance of the series L-C circuit. It is also desirable that the improved lamp ballast circuit permit use of a relatively less e~pensive, smaller capacitor in order to lower the lamp ballast manufacturing cost and to reduce the reactive current flowing through the capacitor after lamp ignition thus 30 lowering circuit power loss.
SU~ARY OF THE INVENTION
In accordance with the invention, a ballast circuit for generating a driving PHA 21.745 32 ~ ~ 4 2 .~ ~ 08.07.1993 signal sufficient to ignite a lam~load as mentioned in the prearnble is characterized in that for the fundamental ~requency ft and the resonant frequency fO it holds:
nf, < fO c (n+l)f, with n = an even integer.
By operating in these regions during pre-ignition, safe voltage and current 5 levels will be maintained. A single drive frequency results in safe non-resonant operation before lamp ignition as well as correct lamp current after ignition. Feedback circuitry for sensing ignition of the larnp load for switching to a diffe~ent steady-state lamp operating frequency need not be provided. By eliminating the need to operate at the resonant frequency of the series connected L-C circuit during pre-ignition of the 10 lamp load, the value and resulting si~e of the capacitor can be chosen far smaller than norrnally used in a conventional series connected L-C circuit in a known ballast circuit.
In accordance with a feature of the invention, the generated signal which is a train of square waves, is generated preferably by a half-bridge or full bridge inverter. In yet another feature of the invention, the resonant frequency of the series 15 connected L-C circuit is less than the third harmonic ~requency of the generated square wave drive thereby avoiding unsafe third harmonic voltages and current levels duAng pre-ignition of the larnp load. Substantially the sarne generated signal frequency is used during pre-ignition and steady-state operation of the lamp load.
Accordingly, it is an object invention to provide an improved ballast 20 circuit in which the unloaded, open circuit voltage and current levels are within the operating range of the ballast circuit components.
It is another object of the invention to provide an impToved ballast circuit in which the same inverter driving signal can be used during pre-ignition and steady-state operation of the larnp load.
It is a further object of the invention to provide an improved ballast circuit in which less costly components can be used to lower the manufacturing cost of the ballast.
It is still another object of the invention to provide an improved ballast circuit which eliminates the need for feedback circuitry for sensing lamp ignition for 30 changing the inverter frequency.
It is still a further object of the invention to provide an improved ballast circuit in which the inverter driving signal frequency is substantially less than the resonant frequency of a series connected L-C output circuit during pre-ignition of the PHA 21.745 ~ ) 2 0~.07.1993 lamp load.
The invention accordingly compnses several steps in a relation of one or more of such steps with respect to each of the others, and the device embodying features of construction, a combination of elements and arrangement of parts which are S adapted to effect such steps, all is exemplified in the following detailed disclosure and the scope of the invention will be indicated in the claims.
BRlEF DESCRIPIION OF DRAWINGS
For a fuller understanding of the invention, reference is made to the 10 following description taken in connection with the accompanying drawings, in which:
Fig. 1 is a circuit diagram of a baLast output circuit in accordance with the present invention;
Figs. 2(a), 2(b) and 2(c) are timing diagrams of an inverter substantially rectangular output voltage, output current at its fundarnental frequency and output 15 current at its third harmonic, respectively in the circuit according to Fig. l;
Fig. 3 is a schematic diagram of a ballast circuit in accordance with the invention;
Figs. 4(a), 4(b), 4(c) and 4(d) are timing diagr~ns of signals produced within the ballast circuit of Fig. 3 during pre-ignition and steady-state operation of the 20 lamp load; and Fig. 5 is a diagram of simulation of current in circuit of Fig. 1 as function of the ratio fundamental frequency and resonant frequency.
DETAILED DESCRIlYrION OF l~ PREFER~ED EMBODIMENT
The figures shown herein illustrate a preferred embodiment of the invention. Those elements/components shown in more than one figure of the drawings have been identified by like reference numerals/letters and are of similar construction and operation.
Referring now to Figs. 1, 2(a), 2(b) and 2(c), a ballast circuit having a 30 ballast output circuit 10 includes an inductor L and a capacitor C seAally connected across the output of a square wave generator 13. Square wave generator 13 is preferably, but not limited to, a bridge inverter generating a substantially square wave of voltage +E (i.e. the inverter output voltage). A larnp load 16 is connected across PHA 21.745 2~ 2 ~ 2 08.07.1993 capacitor C through a switch SW. A current I flowing through inductor L includes a fun,damental frequency component I~, and a third harmonic component of the fundamental frequency I3n. Other currents at higher odd harmonics are present but are significantly smaller. For the sake of simplicity in calculations with respect to the 5 preferred embodiment as described hereafter only terms concerning the fundamental frf~uency f, and the 3rd harmonic are taken into account.
In accordance with the Fourier transform square wave voltage 13 contains a sinusoidal wave at a fundarnental frequency fi and odd harmonics of the fundamental frequency including a sinusoidal wave at a third harmonic 3f,. The amplitude of third 10 harmonic component fi of voltage E is one third the amplitude of fundamental frequency component fi of voltage E.
To achieve low switching losses within square wave generator 13 during pre-ignition of lamp load 16 (generally at trailing edges Er Of voltage E), curTent I is preferably inductive (i.e., current lagging drive voltage) rather than capacitive (i.e.
15 current leading drive voltage) during the voltage transitions of voltage E. Accordingly, the sum of fundamental frequency current component In and third harmonic-currentcomponent I3n is inductive wherein Ilf and I3n are the capacitive and inductive components of I, respectively. To achieve an overall inductive current I, an impedance Z of circuit 10 as viewed from square wave generator 13 requires that the inductive 20 impedance at the third harmonic Z3n be less than one third the capacitive impedance at the fundamental frequency Zn~ In other words, third harmonic component current I3n is greater than fundamental frequency component In. This relationship is illustrated in Figs. 2(b) and 2(c) wherein an arnplitude P represents the peak value of fundamental frequency current component I,l but is less than the peak va1ue of third harmonic 25 current component I3n. In this way the sum of In and I3n remains inductive at the voltage transitions of voltage E.
Lamp load 16 prior to ignition (i.e. during pre-ignition) appears as an open circuit. This open circuit condition is represented by switch SW in an open state (turned OFF). Following ignition, lasnp load 16 is in its steady-state mode of operation and is 30 represented by switch SW being turned ON such that larnp load 16 is connected in pa~llel with capacitor C.
Impedance Z3n~ which must be less than one third impedance Zn during pre-ignition of lamp load 16, is therefore based on switch SW in its open state (i.e., PH~ 21.745 6 ~ 08.07.1993 turned OFF). This condition can be expressed as follows:
IZnl > ¦ 3 Z3nl (eq. 1) That is, ~2~f,xL-1/(2~f,xC)1 > 3 l61rfl xL-1/(6~f~xC)1 (eq. 2) S Since impedance Z is capacitive at fundamental frequency fl and inductive at the third harmonic 3fi, 1/(2~rfixC)-2~rfixL > 18~rfi~cL - l/(2~,xC) That is, 1/(2~fixC)>5(2~fixL) (eq. 3) Eq. 3 can be rewritten as follows:
l/~LC > ~5 2~fi (e~. 4) A resonant frequency f0 of cir~uit l0 during pre-ignition (i.e., with switch 15 SW open) can be defined as follows:
l/~fLC = 27 f0 (eq. 5) Substituting the value of l/~LC defined by eq. 4 for the value of l/~LC in eq. Sresu1ts in 2~fo > ~/5 27fi (eq. 6) Accordingly, resonant frequency f0 can be e~pressed as follows:
fo > ~5 fi (eq. 7) In otha words, third harmonic inductive current component I3n is greater than fundarnental f~equency capacitive current component In when resonant frequency f0 is greater than ~5 times the fundamental frequency of voltage E.
To ensure that unsafe voltages and currents present at resonant frequency f cannot occur, resonant frequency f0 also should be less than third harmonic frequency 3fi of voltage E. Therefore, the values of inductor L and capacitor C should be chosen such th~t:
~5fi < fo ~ 3f~ (eq- 8) By designing baLlast circuit 10 such that resonant frequency f0 is within the range of frequencies defined by eq. 8, the unsafe voltages and currents which occur at resonant frequency f0 during pre ignition of larnp load 16 are avoided and total current delivered by square wave generator 13 remains inductive. There is no need to vary the .
PHA 21.745 .~ ; 2 ~ 2 OB.07.1993 frequency of voltage E between resonant frequency fO during pre-ignition of lamp load 16 and a different frequency immediately thereafter as in conventional ballast circuitry.
Feedback circuitry designed to sense ignition of lamp load 16 for determining when to vary the frequency of voltage E from resonant frequency fO to a different operating S frequency can be eliminated. In accordance with the invention, a safer, simpler circuit is plrovided by maintaining resonant frequency fO within the boundaries defined by eq. 8.
Due to the fact that the calculation as shown has only taken into account the fundamental frequency f, and its 3rd harmonic 3f" the lower value of the range for chosing the resonant frequency fO is ~5 times f,. However, when taken into account the 10 existence of higher harmonics this value reaches the limit 2.
The result of a simulation in which at least the first 25 harmonics are taken into account is shown in figure 5.
In Fig. 5 the depicted cu~ve disp~lays the total current I,Yo in the circuit of Fig. 1 at the moment the voltage switches from -E to +E of the generator 13 as 15 function of ratio of the fundamental frequency fi and the resonant frequency fO. The circuit operates in the inductive mode in all those regions that the current I~o is lagging to the voltage, thus is negative. From the Fig. S it is clear that these regions fulfil the relation nfl < fO < (n+l)fi with n = an even integer.
A ballast circuit 20 in accordance with the invention is shown in Fig. 3.
An input voltage of 277 volts, 60 hertz is supplied to an electromagnetic interference (EMI) suppression filter 23. Filter 23 filters high frequency components inputted thereto lowering conducted and radiated EMI. The output of filter 20 provided at a pair of terminals 24 and 25is supplied to a full wave rectifier 30 which includes diodes Dl, D2, 25 D3 and D". The anode of diode Dl and cathode of diode D2 are connected to terminal 24. The anode of diode D3 and cathode of diode D4 are connected to terminal 25. The output of rectifier 30 (i.e. rectified a.c. signal) at a pair of output terminals 31 and 32is supplied to a boost converter 40. The cathodes of diodes D, and D3 are connected to terminal 31. The cathodes of diodes D2 and D~ are connected to terminal 32.
Converter 40 boosts the magnitude of the rectified A.C. signal supplied by rectifier 30 and produces at a pair of output terminals 41 and 42 a regulated D.C.
voltage supply. Boost converter 40 includes a choke L3, a diode D5 the anode of which is connected to one end of choke L3. The other end of choke L3 is coMected to output .
pHA21.745 8 2~a~25~ 0~.07.1993 terrninal 31 of rectifier 30. The output of boost converter 40 at output tenninals 41, 42 is applied across an electrolytic capacitor C~, one end of which is connected to the cathode of diode Ds~ A transistor (switch) Q, is connected to the junction between choke Ll and the anode of diode D5. The other ~nd of transistor Q, is connected to theS junction between the other end of capacitor Cll, output terrninal 32 of rectifier 30 and output terminal 42.
A preconditioner control SQ, which is powered by a D.C. supply voltage V, controls the switching duration and frequency of transistor Q,. Preconditioner control 50 is preferably~ but not limited to, a Motorola MC33261 Power Factor Controller10 Integrated Circuit. Transistor Q, is preferably a MOSFET, the gate of which is connected to preconditioner control 50. Rectifier 30 and boost converter 40, including preconditioner control 50, forrn a preconditioner 80 for ballast circuit 20. Output terrninals 41 and 42 of boost converter 40 serve as the output for preconditioner 80 across which a regulated D.C. voltage is produced.
A larnp drive 90, which is supplied with the regulated D.C. voltage outputted by preconditioner 80, includes a half bridge inverter having a le~el shifter 60 and a half-bridge drive 70. The half brldge inverter includes a pair of transistors Q6 and Q7, which serve as switches, a pair of capacitors C5 and C6 and a transformer T~. Half-bridge drive 70 produces a square wave driving signal to drive transistor Q7 and has a 20 S0-50 duty cycle. Level shifter 60 inverts the driving signal supplied to transistor Q7 for driving transistor Q6. The driving signals produced by level shifter 60 and half-bridge drive 70 are appro~imately 180 out of phase vith each other so as to prevent conduction of transistors Q6 and Q7 at the same time, respectively.
A source S of transistor Q6 and one end of level shifter 60 are connected to 25 output terminal 41 of boost converter 40. A drain D of transistor Q6 is connected to a terminal A. The o~er end of level shifter 60, one end of half-bridge drive 70 and a source S of transistor Q, are also are coMected to terminal A. The other end of half-bridge drive 70 and a drain D of transistor Q7 are connected to output terminal 42 of boost converter 40. Capacitor C5 iS connected at one end to output terminal 41. The 30 other end of capacitor C5 and one end of capacitor C6 are connected to a terminal B.
The other end of capacitor C6 is connected to output tersl~inal 42.
A primary winding Tp of transformer Tl is connected to terminals A and B. A secondary winding T9 is connected at one end to an inductor L7, the latter which PHA 21.745 9~ 2 08.07.1993 generally represents either the leakage inductance of transformer Tl or a discrete choke.
Connected to the other end of inductor L7 is one end of a capacitor C,0 and one end of a lamp load LL. Lamp load LL can include any combination of lasnps and is shown, but not limited to, the series combination of two fluorescent larnps LLI and LL2. The other S ends of capacitor ClO and la np load LL are connected to the other end of secondaly winding T,.
The turns ratio between primary winding Tp and secondary winding T, of transformer T, is Np/N,. Transformer T, electrically isolates lamp load LL from the output voltage produced by preconditioner 80 and provides sufficient open circuit 10 voltage during pre-ignition to ignite larnp load LL.
The inductance of inductor Lq is based on the desired current flow through laJnp load LL once the latter has ignited and is in its steady-state mode of operation.
The DC voltage a~ross each capacitor C5 and capacitor C6 is approximately half the output voltage of preconditioner 80.
lS The waveforms shown in Figs. 4(a), 4(b), 4(c) and 4(d) produced by ballast circuit 20 are based on turns ratio N,/Np of about 1.5, inductor Lq of approximately 4.3 millihenries, capacitor ClO of about 1.2 nanofarads and capacitors C3 and C4 of about 0.33 microfarads, nominally rated at 630 volts. Both lamp LLI and lamp LL2 are 40 watt low pressure mercury vapor tubular fluorescent lamps. l'he 20 fundamental frequency of the square wave produced by the half-bridge inverter is appro~imately 281~Iz. The resonant frequency of inductor L7 and capacitor ClO isapproximately 70kHz, that is, appro~imately 2.5 times fundamental frequency fi.
During pre-ignition of lamp load LL, the output of ~e half-bridge inverter, which is across terminals A-B, forms a substantially square wave voltage train.
25 Inductor Lq and capacitor C,0 form an L-C series connected circuit. During pre-ignition, lamp load LL appears as a substantially open circuit (i.e. no load condition) drawing substantially no power e~cpect for filament heating (assuming lamps LLI and LL~ are fluorescent lamps of, for e~ample, the rapid-start type).
Fig. 4(a) illustrates a voltage VAB~ that is, between terminals A and B.
30 Voltage VAI~ is square wave voltage train which is applied across primary winding Tp varying between appro~imately +240 volts and -240 volts during no load conditions.
Fig. 4(b) illustrates current Ir~U flowing through primary winding Tp du~ing no load conditions, that is, prior to ignition of larnp load LL and having a peal~ value of PHA 21.745 ~ r~ 2 08.07. 1993 approximately i~ 400 milliamperes. Once lamp load LL is ignited and is in its steady-state operation, current IPRI flowing through primary winding Tp, as shown in Fig. 4(c), has a somewhat sinusoidal wave shape with a ~c value of approximately + 800 milliamperes. Capacitor Cl0 serves to smooth this somewhat sinusoidal current 5 waveforrn resulting in a substantially sinusoidal lamp cuIrent IL~MP as shown in FIG.
4(d) having a peak value of approximately + 380 milliamperes.
Inductor L7 serves as the lamp current ballasting element. Capacitor C,0, which is placed across lamp load LL, provides a more sinusoidal open circuit voltage and keeps total half bridge current inductive while also lowering higher harmonic 10 content of current flowing through larnp load LL. Inductor L, and capacitor Cl0 together form a series coMected L-C output circuit. The value for capacitor Cl0 is chosen such that safe open circuit operation is provided, that is, within the range of resonant frequencies defined by eq. 8. Accordingly, no additional circuits to protect lamp drive circuit 90 are required.
When ballast circuit 20 is first turned on, prior to the voltage being boosted by preconditioner 80, the input voltage of approximately 277 volts results in a square wave voltage of approximately 390 volts peak to peak being applied acrossprimary winding Tp of transformer Tl which is stepped up to approximately 570 volts pealc to peak across secondary winding T,. During this time the lamp cathodes are 20 heated. After approximately 0.5 seconds, preconditioner 80 turns ON resulting in a regulated D.C. voltage of approximately 480 volts across output terminals 41, 42 of boost converter 40 and a voltage of approximately 700 volts peak to peak across secondary winding T" the latter of which is sufficient for igniting larnp load LL. Once lamp load LL is ignited (i.e. during steady-state larnp operation), the lamp voltage (i.e.
25 voltage across larnp load LL) drops to approximately + 300 volts peak with the remainder of the secondary winding T9 output voltage across inductor Lq. The number of and connections between the lamps within larnp load LL can be varied as desired with the value of inductor L, being chosen so as to provide the desired lamp current IL"MP during steady-state operation of larnp load LL.
Referring again to Fig. 3, the rectified AC (i.e. pulsating DC) signal supplied to preconditioner 80 from diode bridge rectifier 30 is boosted in magnitude by choke L3 and diode D5 to charge capacitors C8, C5 and C6. In Fig. 3, capacitor C~ is separate from capacitors C5 and C6, capacitor C~ being a large electrolytic capacitor in PHA 21.745 11 2 ~ rr~ 2 08.07.1993 the range of 5 to 100 microfarads. Capacitors C5 and C6 are high frequency bridge capacitors. Since capacitor C~ is in parallel with the series combination of capacitors C5 andl C6, these three capacitors can be reconfigured as capacitors C5' and C6'.
Preconditioner 80 is an u~converter and boosts the rectified AC input S voltage as follows. When transistor Q6 (which serves as a switch) is closed, choke L3 is short circuited to ground. Current flows through choke L3. Transistor Ql is then opened (turned OFP). Choke L3 with transistor Q, open transfers stored energy through diode D5 into capacitor C~. The amount of energy transferred to capacitor C~ is based on the time during which transistor Ql is turned ON, that is, based on the frequency and 10 duration of the driving signal supplied to the gate of transistor Ql by the preconditioner control 50. Asynchronous operation of transistor Ql with respect to voltage V~, results.
Choke L3 operates in a discontinuous mode, that is, the current through choke L3 during each cycle is reduced to substantially zero before a new cycle is initiated. The frequency at which transistor Q, is turned ON and OFF is varied by 15 preconditioner control 50 so that the peak current through choke L3 is kept constant.
Transistors Q6 and Q7 have internal diodes (not shown). These diodes, which can either be internal or external to the transistors, permit inductive currents to flow through transistors Q6 and Q7 at the initial turn ON and turn OFF of transistors Q6 and Q7.
Preferably, capacitors C5 and C6 are electrolytic capacitors having a pair of 20 discharge resistors in parallel, respectively. Transformer T, is a leakage transformer, that is, having a leakage inductor of inductance LM which serves as the ballast for larnp load LL (i.e. to limit steady state current flow through the lamp load). Alternatively, when transformer Tl has little or no leakage inductance an external inductor of inductance L~, is required for ballast purposes.
Transformer T~ has a main secondary winding TM. A resonant capacitor Cl0 is in series with inductor L7 and reflects back to the primary winding of transformer Tl as a series LC combination across the nalf-bridge inverter.
AS now can be readily appreciated, by maintaining the fundarnental sinusoidal frequency fi well below resonant frequency f0 of the series L-C output 30 circuit, the undesirable and unsafe high voltages and current levels produced in conventional ballast circuits during p~e-ignition of lamp load LI, are avoided. More particularly, by choosing the values of inductor L7 and capacitor Cl0 such that their resonant frequency f0 is defined as described hereinbefore, the voltage level across ., . ~ ., :
, 2 ~ 2 PHA 21.74~ 12 08.07.1993 inductor L7 and capacitor C10 and current flow therethrough will be maintained at levels far below conventional ballast output circuits during pre-ignition of lamp load LL.
By not requiring the combination of inductor L, and capacitor ~,0 to be operated at its resonant frequency fO during pre-ignition of larnp load LL, the value of S capacitor C,0 can be significantly reduced. For example, conventional values for capacitor C10 range from about a nominal value of 6.8 nanofarads to about a nominal value of 9.2 nanofarads. In accordance with the invention, however, capacitor C10 can be reduced in value by approximately one-fourth to one-sixth (e.g. to approximately 1.2 nanofarads). Consequently, a far smaller, less expensive capacitor C,0 is required 10 reducing the manufacturing cost and space requirements of the baUast output circuit.
The reduced value of capacitor CtO results on top of this in substantially all current flowing through larnp load LL with relatively little current flowing through capacitor C2. Power re~quirements for the baUast circuit can be reduced and/or less costly wiring (higher resistance) can be used in the series connected L-C ballast output 15 circuit while maintaining the same power requirements as in a conventional ballast output circuit. In other words, a less costly and/or more efficient ballast with smaller space requirements is provided by the present invention.
Preferably, resonant frequency fO should range from approximately 2.3 to 2.6 times fundamental frequency f, of the square wave generated by the square wave 20 generator. ~ onsequently, stray inductances and the like which may be difficult to account for will not increase the overall inductance. Resonant frequency fO wiU not approach third harmonic frequency 3f,. Unsafe operation (i.e., resonant operation of the series L-C output circuit) of baUast circuit 20 is prevented.
GeneraUy, in calculating the inductance of inductor L, for determining 25 resonant frequency fO, the leakage inductance of transformer T~ or inductance of the discrete choke used for inductor L, is far greater than the stray inductance or other inductances within baUast circuit 20. Therefore, ~ a first order appro~cimation, the inducta~ce of inductor L, can be used without taking into account stray inductances and the like in determining the resonant frequency fO. For a tightly wound transformer T, in 30 which very little or an insufficient unount of leakage inductance e~cists, a discrete inductor will be required to serve as the baUasting element for lamp load LL (i.e., to control the larnp current I, ~,p).
As now can be readily appreciated, the generated voltage (i.e. voltage E of 2.~ ~
PHA 21.745 13 08.07.1993 Fig. 1 and voltage VA-~ of Fig. 4(a)) is at a frequency which is far less than the resonant frequency of the series connected L-C circuit and therefore provides safe open circuit (pre-ignition) voltages and current levels. The frequency of this generat~d signal need not be changed following pre-ignition since it is never at or near resonantS frequency fO of the series connected L-C circuit. Feedback circuitry for sensing ignition of lamp load LL for switching to a different steady-state lamp operating frequency need not be provided. By eliminated the need to operate at resonant frequency S of the series L-C circuit during pre-ignition of lamp load LL, the value and resulting size of the capacitor for the series connected L-C circuit ~an be far smaller than normally used in a 10 conventional series connectecl L-C circuit.
Claims (6)
1. A ballast circuit for generating a substantially rectangular driving signal sufficient to ignite a lamp load, comprising:
inductor means;
capacitor means serially connected to said inductor means; and generating means for applying a generated signal to said serially connected inductor means and capacitor means, said generated signal having at least a fundamental frequency f1;
the inductor means and capacitor means having a resonant frequency f0, characterized in that for the fundamental frequency and the resonant frequency it holds:
nf1 < f0 < (n+1)f1 with n = an even integer.
inductor means;
capacitor means serially connected to said inductor means; and generating means for applying a generated signal to said serially connected inductor means and capacitor means, said generated signal having at least a fundamental frequency f1;
the inductor means and capacitor means having a resonant frequency f0, characterized in that for the fundamental frequency and the resonant frequency it holds:
nf1 < f0 < (n+1)f1 with n = an even integer.
2. A ballast circuit of claim 1, characterized in that the generating means includes a half-bridge inverter.
3. A ballast circuit of claim 1 or 2, wherein the resonant frequency f0 is less than a third harmonic of said fundamental frequency f1.
4. A ballast circuit of claim 1, 2 or 3, wherein the lamp load after ignition enters into a steady-state mode of operation in which current therethrough is maintained at a substantially constant level, characterized in that in the steady-state mode the generating means apply said generated signal to said serially connected inductor means and capacitor means.
5. A ballast circuit of claim 1, 2, 3 or 4 wherein said lamp load is connected across the capacitor means.
6. A ballast circuit of claim 1, 2, 3, 4 or 5, wherein the lamp load includes at least one fluorescent lamp.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US93284092A | 1992-08-20 | 1992-08-20 | |
| US932,840 | 1992-08-20 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2104252A1 true CA2104252A1 (en) | 1994-02-21 |
Family
ID=25463035
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002104252A Abandoned CA2104252A1 (en) | 1992-08-20 | 1993-08-17 | Lamp ballast circuit |
Country Status (12)
| Country | Link |
|---|---|
| US (2) | US5463284A (en) |
| EP (1) | EP0583838B1 (en) |
| JP (1) | JPH06176881A (en) |
| KR (1) | KR100289019B1 (en) |
| AT (1) | ATE147925T1 (en) |
| CA (1) | CA2104252A1 (en) |
| DE (1) | DE69307427T2 (en) |
| ES (1) | ES2099369T3 (en) |
| FI (1) | FI108910B (en) |
| MX (1) | MX9305064A (en) |
| SG (1) | SG48129A1 (en) |
| TW (1) | TW394493U (en) |
Families Citing this family (42)
| Publication number | Priority date | Publication date | Assignee | Title |
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| CA2104252A1 (en) * | 1992-08-20 | 1994-02-21 | Charles B. Mattas | Lamp ballast circuit |
| US5563473A (en) * | 1992-08-20 | 1996-10-08 | Philips Electronics North America Corp. | Electronic ballast for operating lamps in parallel |
| US5545955A (en) * | 1994-03-04 | 1996-08-13 | International Rectifier Corporation | MOS gate driver for ballast circuits |
| US5463283A (en) * | 1994-05-24 | 1995-10-31 | Bkl, Inc. | Drive circuit for electroluminescent lamp |
| US5550436A (en) * | 1994-09-01 | 1996-08-27 | International Rectifier Corporation | MOS gate driver integrated circuit for ballast circuits |
| US5834906A (en) * | 1995-05-31 | 1998-11-10 | Philips Electronics North America Corporation | Instant start for an electronic ballast preconditioner having an active power factor controller |
| US5825139A (en) * | 1995-11-02 | 1998-10-20 | Hubbell Incorporated | Lamp driven voltage transformation and ballasting system |
| US5962988A (en) * | 1995-11-02 | 1999-10-05 | Hubbell Incorporated | Multi-voltage ballast and dimming circuits for a lamp drive voltage transformation and ballasting system |
| DE29605913U1 (en) * | 1996-03-29 | 1996-06-13 | Trilux Lenze Gmbh & Co Kg | Fluorescent lamp ballast |
| CN1143604C (en) * | 1996-05-10 | 2004-03-24 | 皇家菲利浦电子有限公司 | Circuit arrangement |
| US5747942A (en) * | 1996-07-10 | 1998-05-05 | Enersol Systems, Inc. | Inverter for an electronic ballast having independent start-up and operational output voltages |
| US5912812A (en) * | 1996-12-19 | 1999-06-15 | Lucent Technologies Inc. | Boost power converter for powering a load from an AC source |
| CA2198173A1 (en) * | 1997-02-21 | 1998-08-21 | Exacta Transformers Of Canada Ltd. | Micro-controller-operated high intensity discharge lamp ballast system and method |
| EP1023818A4 (en) * | 1997-07-25 | 2003-08-13 | Jorge M Parra | Low-voltage non-thermionic ballast-free fluorescent light system and method |
| US5969483A (en) * | 1998-03-30 | 1999-10-19 | Motorola | Inverter control method for electronic ballasts |
| US5945788A (en) * | 1998-03-30 | 1999-08-31 | Motorola Inc. | Electronic ballast with inverter control circuit |
| IL125328A0 (en) * | 1998-07-13 | 1999-03-12 | Univ Ben Gurion | Modular apparatus for regulating the harmonics of current drawn from power lines |
| US6111369A (en) * | 1998-12-18 | 2000-08-29 | Clalight Israel Ltd. | Electronic ballast |
| JP2003525518A (en) * | 2000-02-29 | 2003-08-26 | コーニンクレッカ フィリップス エレクトロニクス エヌ ヴィ | Electronic ballast |
| DE10013342A1 (en) * | 2000-03-17 | 2001-09-27 | Trilux Lenze Gmbh & Co Kg | Producing ignition voltage for fluorescent lamps involves applying start voltages of reducing frequency until lamp voltage reaches desired value |
| US6661185B2 (en) * | 2001-03-22 | 2003-12-09 | Matsushita Electric Industrial Co., Ltd. | Dimmable self-ballasted fluorescent lamp and discharge lamp operating apparatus |
| DE10128687A1 (en) * | 2001-06-13 | 2002-12-19 | Philips Corp Intellectual Pty | DC converter |
| US6677715B2 (en) * | 2001-09-19 | 2004-01-13 | General Electric Company | Portable electronic ballast |
| WO2003039211A1 (en) * | 2001-10-31 | 2003-05-08 | Koninklijke Philips Electronics N.V. | Circuit arrangement |
| US6731075B2 (en) * | 2001-11-02 | 2004-05-04 | Ampr Llc | Method and apparatus for lighting a discharge lamp |
| NL1020276C2 (en) * | 2002-03-28 | 2003-09-30 | Nedap Nv | Electronic ballast for gas discharge lamps. |
| US7247998B2 (en) * | 2002-07-31 | 2007-07-24 | Universal Lighting Technologies, Inc. | Transient detection of end of lamp life condition apparatus and method |
| US6781352B2 (en) | 2002-12-16 | 2004-08-24 | International Rectifer Corporation | One cycle control continuous conduction mode PFC boost converter integrated circuit with integrated power switch and boost converter |
| JP2005063821A (en) * | 2003-08-13 | 2005-03-10 | Koito Mfg Co Ltd | Discharge lamp lighting circuit and method |
| JP2005129004A (en) * | 2003-10-03 | 2005-05-19 | Sharp Corp | Driving system and a.c. converter |
| KR100840683B1 (en) * | 2004-03-25 | 2008-06-24 | 엘지디스플레이 주식회사 | back light assembly |
| DE102004017479A1 (en) * | 2004-04-08 | 2005-10-27 | Patent-Treuhand-Gesellschaft für elektrische Glühlampen mbH | ECG with resonance stimulus for transfer voltage generation |
| GB2417592B (en) * | 2004-08-13 | 2006-07-26 | Ingenia Technology Ltd | Authenticity verification of articles |
| KR100675224B1 (en) * | 2005-05-09 | 2007-01-26 | 삼성전기주식회사 | Driving method of external electrode fluorescent lamp inverter for backlight |
| US7715698B2 (en) * | 2005-08-31 | 2010-05-11 | Thor Power Corporation | Control electronics for brushless motors |
| US7701153B2 (en) * | 2006-12-15 | 2010-04-20 | Panasonic Corporation | Visible indication of mistaken lamp use |
| ES2522968T3 (en) | 2008-06-04 | 2014-11-19 | Synergy Pharmaceuticals Inc. | Guanylate cyclase agonists useful for the treatment of gastrointestinal disorders, inflammation, cancer and other disorders |
| CN102388530B (en) | 2008-10-01 | 2014-09-03 | 雷斯特克有限公司 | Circuit and method for coupling electrical energy to a resonated inductive load |
| RU2566736C2 (en) * | 2009-02-26 | 2015-10-27 | Конинклейке Филипс Электроникс Н.В. | Resonant converter |
| US20110273118A1 (en) * | 2010-05-10 | 2011-11-10 | David Bonner | Power Factor Correction Circuit |
| CN104282281B (en) * | 2014-10-20 | 2016-11-09 | 深圳市华星光电技术有限公司 | A kind of LED backlight drive circuit and fault detection method thereof |
| DE102014226716A1 (en) * | 2014-12-19 | 2016-06-23 | Dialog Semiconductor (Uk) Limited | Voltage doubling and voltage doubling methods for use in PMW mode |
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| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| NL7402779A (en) * | 1974-03-01 | 1975-09-03 | Philips Nv | DEVICE EQUIPPED WITH A GAS AND / OR VAPOR DISCHARGE LAMP. |
| US4060752A (en) * | 1976-03-01 | 1977-11-29 | General Electric Company | Discharge lamp auxiliary circuit with dI/dt switching control |
| US4053813A (en) * | 1976-03-01 | 1977-10-11 | General Electric Company | Discharge lamp ballast with resonant starting |
| US4060751A (en) * | 1976-03-01 | 1977-11-29 | General Electric Company | Dual mode solid state inverter circuit for starting and ballasting gas discharge lamps |
| US5214356A (en) * | 1978-12-28 | 1993-05-25 | Nilssen Ole K | Dimmable fluorescent lamp ballast |
| WO1981003731A1 (en) * | 1980-06-10 | 1981-12-24 | Kaunassk Polt Inst Antanasa | Lighting system |
| DE3441992A1 (en) * | 1984-11-16 | 1986-05-22 | Patent-Treuhand-Gesellschaft für elektrische Glühlampen mbH, 8000 München | CIRCUIT ARRANGEMENT FOR IGNITING A LOW-PRESSURE DISCHARGE LAMP |
| US4641061A (en) * | 1985-04-22 | 1987-02-03 | Emerson Electric Co. | Solid state ballast for gaseous discharge lamps |
| US4730147A (en) * | 1986-08-19 | 1988-03-08 | Siemens Aktiengesellschaft | Method and arrangement for the operation of a gas discharge lamp |
| US5013974A (en) * | 1987-08-24 | 1991-05-07 | Nilssen Ole K | Electronic ballast with improved lamp current crest factor |
| ATE103458T1 (en) * | 1988-04-20 | 1994-04-15 | Zumtobel Ag | BALLAST FOR A DISCHARGE LAMP. |
| DE3843029A1 (en) * | 1988-12-21 | 1990-06-28 | Hella Kg Hueck & Co | DEVICE FOR IGNITING AND OPERATING ELECTRIC GAS DISCHARGE LAMPS |
| US5289083A (en) * | 1989-04-03 | 1994-02-22 | Etta Industries, Inc. | Resonant inverter circuitry for effecting fundamental or harmonic resonance mode starting of a gas discharge lamp |
| DE9005084U1 (en) * | 1989-05-23 | 1990-07-12 | Siemens Ag, 1000 Berlin Und 8000 Muenchen, De | |
| US5075599A (en) * | 1989-11-29 | 1991-12-24 | U.S. Philips Corporation | Circuit arrangement |
| CA2104252A1 (en) * | 1992-08-20 | 1994-02-21 | Charles B. Mattas | Lamp ballast circuit |
-
1993
- 1993-08-17 CA CA002104252A patent/CA2104252A1/en not_active Abandoned
- 1993-08-17 JP JP5203376A patent/JPH06176881A/en active Pending
- 1993-08-17 DE DE69307427T patent/DE69307427T2/en not_active Expired - Fee Related
- 1993-08-17 AT AT93202406T patent/ATE147925T1/en not_active IP Right Cessation
- 1993-08-17 ES ES93202406T patent/ES2099369T3/en not_active Expired - Lifetime
- 1993-08-17 FI FI933626A patent/FI108910B/en not_active Application Discontinuation
- 1993-08-17 EP EP93202406A patent/EP0583838B1/en not_active Expired - Lifetime
- 1993-08-17 SG SG1996007205A patent/SG48129A1/en unknown
- 1993-08-20 MX MX9305064A patent/MX9305064A/en not_active IP Right Cessation
- 1993-08-20 KR KR1019930016192A patent/KR100289019B1/en not_active IP Right Cessation
- 1993-09-14 TW TW086218665U patent/TW394493U/en not_active IP Right Cessation
-
1994
- 1994-10-26 US US08/329,700 patent/US5463284A/en not_active Expired - Fee Related
-
1995
- 1995-06-05 US US08/461,459 patent/US5686798A/en not_active Expired - Fee Related
Also Published As
| Publication number | Publication date |
|---|---|
| US5463284A (en) | 1995-10-31 |
| JPH06176881A (en) | 1994-06-24 |
| MX9305064A (en) | 1994-06-30 |
| TW394493U (en) | 2000-06-11 |
| KR940005193A (en) | 1994-03-16 |
| EP0583838A3 (en) | 1994-03-09 |
| SG48129A1 (en) | 1998-04-17 |
| EP0583838B1 (en) | 1997-01-15 |
| FI933626A0 (en) | 1993-08-17 |
| KR100289019B1 (en) | 2001-05-02 |
| ATE147925T1 (en) | 1997-02-15 |
| FI108910B (en) | 2002-04-15 |
| DE69307427D1 (en) | 1997-02-27 |
| EP0583838A2 (en) | 1994-02-23 |
| DE69307427T2 (en) | 1997-07-17 |
| ES2099369T3 (en) | 1997-05-16 |
| US5686798A (en) | 1997-11-11 |
| FI933626A (en) | 1994-02-21 |
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Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| FZDE | Discontinued |