CA2104658A1 - Ils signal analysis device and method - Google Patents
Ils signal analysis device and methodInfo
- Publication number
- CA2104658A1 CA2104658A1 CA002104658A CA2104658A CA2104658A1 CA 2104658 A1 CA2104658 A1 CA 2104658A1 CA 002104658 A CA002104658 A CA 002104658A CA 2104658 A CA2104658 A CA 2104658A CA 2104658 A1 CA2104658 A1 CA 2104658A1
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- CA
- Canada
- Prior art keywords
- signal
- ils
- nte
- frequency
- analysed
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Abandoned
Links
Classifications
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S1/00—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith
- G01S1/02—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith using radio waves
- G01S1/08—Systems for determining direction or position line
- G01S1/14—Systems for determining direction or position line using amplitude comparison of signals transmitted simultaneously from antennas or antenna systems having differently oriented overlapping directivity-characteristics
-
- G—PHYSICS
- G01—MEASURING; TESTING
- G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
- G01S1/00—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith
- G01S1/02—Beacons or beacon systems transmitting signals having a characteristic or characteristics capable of being detected by non-directional receivers and defining directions, positions, or position lines fixed relatively to the beacon transmitters; Receivers co-operating therewith using radio waves
- G01S1/022—Means for monitoring or calibrating
- G01S1/024—Means for monitoring or calibrating of beacon transmitters
Abstract
ABSTRACT OF THE DISCLOSURE
The invention relates to an instrument landing system (ILS) signal analysis device including an analog/
digital converter receiving the composite signal to be analysed and delivering a succession of values which can be processed in digital form, the signal to be analysed being furthermore applied to phase-locking means (2, 3) which deliver for the said analog/digital converter (10) a sampling signal of frequency greater than the largest frequency of the components of the signal to be analysed, the digital processing of the values from the converter (10), carried out in real time between two sampling instants, allowing determination of the parameters for modulation of the ILS signal.
The phase-lock loop is synchronized with the frequency of the signal to be analysed, which frequency is extracted from a sub-harmonic of the 90 Hz and 150 Hz components constituting this signal.
The invention relates to an instrument landing system (ILS) signal analysis device including an analog/
digital converter receiving the composite signal to be analysed and delivering a succession of values which can be processed in digital form, the signal to be analysed being furthermore applied to phase-locking means (2, 3) which deliver for the said analog/digital converter (10) a sampling signal of frequency greater than the largest frequency of the components of the signal to be analysed, the digital processing of the values from the converter (10), carried out in real time between two sampling instants, allowing determination of the parameters for modulation of the ILS signal.
The phase-lock loop is synchronized with the frequency of the signal to be analysed, which frequency is extracted from a sub-harmonic of the 90 Hz and 150 Hz components constituting this signal.
Description
46~8 ILS SIGNAI- ANALYSIS DEVICE AND METHOD
The present invention relates to a method and a device for analysing landing ~ystem ~ignal~ known as ILS
"Instrument Landing System" signals. The device i~
intended in particular for the ground control of all the characteristics of instrument landing installations.
In French Patent Application FR-A-l 487 441, the Applicant has described a method of measuring an ILS
signal implementing an analog technique evaluating the deviation of the aircraft with respect to the landing path, from the determination of the difference in modula-tion existing between the two components, 90 and 150 ~z, constituting the composite ILS signal.
The French Patent Application FR-A-2 596 547 shows a device for formulating~ radionavigation information relying, in one embodiment, on a digital architecture implementing a microprocessor assembly at the level of which the ILS signal is processed with an algorithm calling upon a recursive fast Fourier transform ` 20 calculation.
These two requirements, which describe devices intended to serve as on-board navigation receivers, give rise to constraints distinct from that of a device for ; measurement and analysis; in particular, they necessitate integration of the flaws due, among other things, to the obstacles which may appear in the landing path, 80 as to prevent any tacking during approach. This "smoothing" of the information only allows small data renewal rates which, however, permit recourse to processing operations ~uch as the recursive fast Fourier transform which demands long observation times.
By contrast, in an analysis device ~uch a~ that ; of the invention, it is e~sential to be able to a~se~s all the imperfections of the ~ystem to be controlled, this requiring particularly fa~t respon~e times.
One purpose of the invention is to produce a measurement device having high performance, that is to say great accuracy and a not lower processing speed.
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The present invention relates to a method and a device for analysing landing ~ystem ~ignal~ known as ILS
"Instrument Landing System" signals. The device i~
intended in particular for the ground control of all the characteristics of instrument landing installations.
In French Patent Application FR-A-l 487 441, the Applicant has described a method of measuring an ILS
signal implementing an analog technique evaluating the deviation of the aircraft with respect to the landing path, from the determination of the difference in modula-tion existing between the two components, 90 and 150 ~z, constituting the composite ILS signal.
The French Patent Application FR-A-2 596 547 shows a device for formulating~ radionavigation information relying, in one embodiment, on a digital architecture implementing a microprocessor assembly at the level of which the ILS signal is processed with an algorithm calling upon a recursive fast Fourier transform ` 20 calculation.
These two requirements, which describe devices intended to serve as on-board navigation receivers, give rise to constraints distinct from that of a device for ; measurement and analysis; in particular, they necessitate integration of the flaws due, among other things, to the obstacles which may appear in the landing path, 80 as to prevent any tacking during approach. This "smoothing" of the information only allows small data renewal rates which, however, permit recourse to processing operations ~uch as the recursive fast Fourier transform which demands long observation times.
By contrast, in an analysis device ~uch a~ that ; of the invention, it is e~sential to be able to a~se~s all the imperfections of the ~ystem to be controlled, this requiring particularly fa~t respon~e times.
One purpose of the invention is to produce a measurement device having high performance, that is to say great accuracy and a not lower processing speed.
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- 2 -Another purpo~e of the invention i8 to produce a ~imple and compact device employing a restricted number of components and thereby permitting fast and reduced maintenance.
These purposes are achieved with an instrument landing system (ILS) signal analysis device including an analog/digital converter receiving the composite signal to be analysed and delivering a succession of values which can be processed in digital form, characterized in that the said signal to be analysed is furthermore applied to phase-locking means which deliver for the said analog/digital converter a sampling signal of freguency greater than the largest frequency of the components of the signal to be analysed,..the digital proces~ing of the values from the converter, carried out in real time between two sampling instants, allowing determination of the parameters for modulation of the ILS signal.
- The phase-lock loop is synchronized directly with - the frequency of the signal to be analysed, which freque-ncy is extracted from a sub-harmonic of the 90 Bz and 150 Hz components constituting this signal.
. This specific architecture allows the calculation procedure, implemented at the level of the digital processing assembly, to be synchronized directly with the composite ILS signal 80 as constantly to assess the maximum amplitude of the 90 and 150 Hz components which is contained in this ILS ~ignal, and this despite the frequency or phase drift existing at the level of the signal or despite it~ distortion factor. Furthermore, the instantaneous processing between each sample of the numerical value~ acquired makes it possible to limit the memory capacity of the device and thus to afford it maximum compactne~s.
The digital processing of the values from the converter is carried out by a processing unit with which are as~ociated a random-acces~ memory, a non-volatile memory and an input~output module.
The non-volatile memory includes a first table in which are written numerical values corresponding to the , .: , . . ~
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These purposes are achieved with an instrument landing system (ILS) signal analysis device including an analog/digital converter receiving the composite signal to be analysed and delivering a succession of values which can be processed in digital form, characterized in that the said signal to be analysed is furthermore applied to phase-locking means which deliver for the said analog/digital converter a sampling signal of freguency greater than the largest frequency of the components of the signal to be analysed,..the digital proces~ing of the values from the converter, carried out in real time between two sampling instants, allowing determination of the parameters for modulation of the ILS signal.
- The phase-lock loop is synchronized directly with - the frequency of the signal to be analysed, which freque-ncy is extracted from a sub-harmonic of the 90 Bz and 150 Hz components constituting this signal.
. This specific architecture allows the calculation procedure, implemented at the level of the digital processing assembly, to be synchronized directly with the composite ILS signal 80 as constantly to assess the maximum amplitude of the 90 and 150 Hz components which is contained in this ILS ~ignal, and this despite the frequency or phase drift existing at the level of the signal or despite it~ distortion factor. Furthermore, the instantaneous processing between each sample of the numerical value~ acquired makes it possible to limit the memory capacity of the device and thus to afford it maximum compactne~s.
The digital processing of the values from the converter is carried out by a processing unit with which are as~ociated a random-acces~ memory, a non-volatile memory and an input~output module.
The non-volatile memory includes a first table in which are written numerical values corresponding to the , .: , . . ~
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- 3 -value of a weighting window function for each sampling instant, as well as at least two table~ in each of which are written numerical values repre~entative, over a quarter period, of a co~ine function of frequency equal to that of a characteristic component of the signal to be analysed.
Recourse to these tables of values avoid~ calcu-lation, for each sample acquired, of the values of the functions employed in the calculation procedure.
Preferably, for the processing of the identifica-tion signal contained in the ILS signal, the analysis device according to the invention includes a digital filter using a weighting function whose equation i~ given by: .
sin k~ 2~t W(t) - 2 [~+(l~~cos~] with ~ -~ - T
where T represents the-duration of observation of the ~ signal to be analysed and k~ and a are specified numeri-cal coefficients.
The choice of this window (and in particular with k' - 4 and a - 0.54) makes it-possible to obtain a filter having a very selective and quasi-linear frequency response in the pass-band.
As before, this weighting window is advantageously stored in a non-volatile memory in the form of a table of numerical values.
The device according to the invention is imple-mented in accordance with a method including the follow-ing steps:
a) acquisition of a sample S(nTe) of the ILS ~ignal to be analysed by the analog/digital conversion of this signal, Te representing the sampling period and n being an integer number, the sampling frequency being greater than the largest of the frequencies k of the components of the signal to be analysed, b) multiplication of the value of the sample taken by a first numerical value corresponding to the vaiue of a weighting window function at the acquisition instant:
S(nTe) - Fen(nTe~ x S(nTe)
Recourse to these tables of values avoid~ calcu-lation, for each sample acquired, of the values of the functions employed in the calculation procedure.
Preferably, for the processing of the identifica-tion signal contained in the ILS signal, the analysis device according to the invention includes a digital filter using a weighting function whose equation i~ given by: .
sin k~ 2~t W(t) - 2 [~+(l~~cos~] with ~ -~ - T
where T represents the-duration of observation of the ~ signal to be analysed and k~ and a are specified numeri-cal coefficients.
The choice of this window (and in particular with k' - 4 and a - 0.54) makes it-possible to obtain a filter having a very selective and quasi-linear frequency response in the pass-band.
As before, this weighting window is advantageously stored in a non-volatile memory in the form of a table of numerical values.
The device according to the invention is imple-mented in accordance with a method including the follow-ing steps:
a) acquisition of a sample S(nTe) of the ILS ~ignal to be analysed by the analog/digital conversion of this signal, Te representing the sampling period and n being an integer number, the sampling frequency being greater than the largest of the frequencies k of the components of the signal to be analysed, b) multiplication of the value of the sample taken by a first numerical value corresponding to the vaiue of a weighting window function at the acquisition instant:
S(nTe) - Fen(nTe~ x S(nTe)
- 4 -c) determination, at the frequencies k of the character-istic components of the ILS signal, of the real and imaginary parts of the spectrum of the sampled and weighted signal S(t):
R(k) ~ R(k) + S(nTe) x cos 2~k nTe and I(k) - I(k) + S(nTe) x sin 2~k nTe d) repetition of steps a) to c) for all of the ~amples over a duration of observation T of the ILS signal, e) calculation of the modulation factors for the charac-teristic component~ of the ILS signal.
M O = ~ .
HF being the average value of S(nTe) calculated over all the samples taken.
f) calculation of the difference (DDM) and sum (SDM) of the modulations (M (90) and M~150)) of the 90 and 150 Hz - 15 components of the ILS signal:
-SDM ~ M(90) + N(150) and DDM ~ M(90) - M(150).
By performing the above ¢alculation~ on numerous samples, advantageously 540, the effect of the analog or guantization noise becomes virtually nil. Moreover, measurement of the amplitude of each component of the signal is effected selectively about each frequency, this i having the effect of further limiting the wide band noise.
In a particular embodiment, it iB possible to take just one sample of the signal to be analysed every second or third value from the table~ of the cosine function, thereby making it possible to determine also the amplitude of the 2nd or 3rd harmonic components of this signal.
Other characteristics and advantages of the present invention will emerge better on reading the following description given, by way of non-llm;ting illustrative example, in connection with the attached drawing~ in which:
- Figure 1 shows the structure of an ~LS signal analysis device according to the invention, . - -,- '' ,' ' ' .
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` ` ~lU4~8 - 5 - Figures 2a to 2e, 3a to 3e and 4 are a graphical approximation making it possible to observe the trans-formations carried out on the ILS signal in the time domain and in the frequency domain as well as the rela-tions between these two domains, - Figure 5 is a flow chart describing the digital pro-cessing of the ILS signals in the device according to the invention, - Figure 6 ~hows one form of the signal stored in the tables of the device and utilized in the digital process-ing described in Figure 4, - Figure 7 i8 a flow chart describing the processing of the identification signal in the device according to the invention, --- Figure 8 represents, in the time domain, a ~weighting window" function employed in the processing of the identification signal, and - - Figure 9 represents the frequency r~ponse of the - digital filter corresponding to the weighting window of Figure 8.
It i~ known that, in an instrument landing system, two transmitters are employed; one, at the frequency 110 MBz, known as the localizing signal or ~localizer~ and emitting a beam 2.5 wide on either side of the axis of the landing ~trip, and the other, at the frequency 332 MBz, known as the "glide-path~ signal and emitting a be~m 0.7 wide about a-typical 3 glide path.
In practice, these beams each consist of the zone common to two low-frequency modulated main beam~, one at 90 Hz the other at 150 Bz. The resulting ILS signal therefore in theory assumes the form of a high-frequency signal (110 or 332 MHz) amplitude modulated by two 90 and 150 Hz sinusoidal voltages.
In the particular case of the localizing ILS
signal, a 1020 Bz identification signal is furthermore superimposed on the 90 and 150 Bz components.
Figure 1 is a preferred illustrative embodiment of a device for proce3sing such ILS signals, according to the invention.
:
:
6 ~ 8 ~ his ILS signal constitutes an input ~ignal, on the one hand, for an anti-alia~ing low-pass filter 1 and, on the other hand, for phaso-locking moans consisting of a phase-lock loop (PLL)2 who~e output is connected to a frequency synthesizer 3. Output from the low-pa~s filter and from the synthesizer 3 constitute inputs for a digital proces~ing assembly 4.
In a known manner, a phase-lock loop con~ists of a phase comparator who~e output controls a voltage-controlled oscillator (VCO) which in turn feeds back tothe input of the comparator. In the context of the pre~ent invention, the synchronizing of thi~ circuit is carried out, not with a fundamental frequency of the analysed signal, as is frequently done, but with a sub-harmonic of this signal.-In this-instance, this is the 3rd sub-harmonic of the 90 Hz signal or the 5th sub-harmonic of the 150 Hz signal, that is to say the 30 ~z - frequency.
~Likewise in a known manner, a frequency synthe-sizer takes the form of a phase-lock loop into which a programmable divider is inserted within the feedback circuit.
In the example of the invention, the programmable divider is a divider by 540 making it pos~ible to obtain a frequency of 16.2 k~z at the output of the synthe~izer ! 3 from the stable frequency of 30 Hz provided by the pha~e-lock loop 2. This sampling frequency i8 greater than the highest frequency of the components of the signal to be analysed. Advantageously, in a ratio at least 1 to 10 for the identification signal and at least 1 to 100 for the 150 ~z component.
The processing as~embly 4 includes an analog/digital converter 10 which receives the ILS signal from tho anti-aliasing filter 1 and deliver~ on 10 bits a ~ampled signal S*(t) clocked by a sampling signal delivered by an input/output module 11. The result of this conversion iB 3tored in a random-acces~ memory 12 which is connected to a processing unit 13, which is al~o connected to a non-volatile memory 14.
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~ ~ u ~ 8 _ 7 _ The input/output module 11 which receive~ the 16.2 kHz synchronization ~ignal generated by the fre-quency synthesizer 3 i8 connected to the proce~ing unit 13 and also deliver~ signals for exploitation of the analysed ILS signal.
The transfers of information between the proce~s-ing unit and its a~oeiated circuits, memories 12, 14 or input/output module 11, are advantageously carried out on 16 bit~.
The non-volatile memory contains a program for manaqing all the processing as well as several tables of values, the utility of whieh will emerge hereafter, in conneetion with Figures 2 to 9 which explain the oper-ation of the proceseing unit according to the invention.
Figuree 2 and 3 show the various operations performed, by the processing as~embly, on the signal both in the time domain and in the frequency domain.
- Figure 2a shows the ILS signal on input to the deviee aecording to the invention. By way of in~truetion, the analysed signal i8 the loealizing signal whieh contains only the 90 Hz and 150 Hz eomponents to the exclusion of the 1020 Hz identification signal. However, it i8 obvious that a comparable analysis may be condueted with the loealizing ~y~tem whose demodulated signal would then also yield this 1020 Hz signal. Furthermore, the ; ~ignal will firstly be assumed to be complete, without harmonies.
The form of the demodulated signal is given by the following equation:
S(t) ~ HF (1 + M90 ~in 2~.90.t + M150 sin 2~.150.t) with HF the continuous eomponent refleeting the 332 M~z high-frequeney carrier on which the ~um of the 90 Hz and 150 Hz waves is ~uperimpo~ed, N90 the modulation faetor for the 90 Hz wave, M150 the modulation faetor for the 150 Hz wave.
These modulation factors serve to reconstitute the difference of the modulations (DDM) and the ~um of the modulations (SDM), which parameters, in the ca~e of the former, characterizea the position of the aircraft , , , , ' .
~lU~6~8 with respect to the glide path (tracking path) and, in the case of the latter, serves in the validation of this previou~ measurement.
The sampling of the signal S(t~ at a sampling frequency Fe - 1/Te i8 achieved by multiplying thi~
~ignal by a Dirac comb ~(t) of period Te (see Figure 2b) namely S*(t) ~ S(t)x~(t) (Figure 2c), S* being a seguence of values of S(t) taken at the instants nTe.
In the frequency domain, this sampling corre-sponds to a convolution between the signal S(F) (Figure3a) and the signal ~(F) (Figure 3b), which gives a spectrum S*(F) which is the repeat of S~F) about the multiple frequencies of Fe (see Figure 3c).
It can then~ be seen that in order to sample, without lo~ing information, this signal S(t) who~e spectrum i~ bounded by a maximum freguency Fm; it is absolutely neces~ary for the sampling frequency Fe to be - greater than 2Fm. -~However, it must be certain that the spectrum of S(t) is bounded by this value Fm, now, this signal is subjected to fnst variatione and multiple interference depending on the configuration of the strip at the time of landing. This is why, in order to avoid these disad-vantages, the sampling ha~ been preceded by a low-pass filter which necessarily limits the spectrum to Fm, thus avoiding any unde~ired disturbances; this is the anti-~; aliasing filter 1. The frequency spectrum output by this filter is identical to that of S(t) and it is this signal, and not S(t), which i~ in reality the subject of the aforesaid sampling.
Figures 2d and 3d repressnt respectively in thetime domain and in the frequency domain a weighting function known as a Hamming window.
The-sampling may not be carried out over the whole of the signal S(t), that is to say for an infinite number of samples corresponding to an infinite ~ignal observation time. The number of sample~ must be finite.
This limitation reduces the period of observ~tion to a value T, this being eguivalent to multiplying the sampled :
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6 ~ 8 _ 9 _ function S*(t) by a window function of width T. So as to reduce the effects due to this truncation, it i~ advis-able to resort to a specific window such as the Ha~ming window which, in respect of its frequency spectrum, ha~
secondary side lobes of very small amplitudes, the main lobe having a witth of 1/T.
Thi~ window function is defined by the following equation:
t ham(t) - a + (1-) C08 (2n T
Preferably, the value 0.54 will be chosen for a, without however being limited to this value alone (the value a-0.5 which corresponds to a Hann window could, for example, also be envisaged).
Figures 2e and 3e show the shape of the signal S(t) after sampling and truncation and that of the j~ corrosponding spectrum.
~~ The latter i8 given by the convolution of the spectrum of the s~mpled signal S~(F) and of the ~pectrum of the ham window (F) namely: S*(F)* ham (F). In the frequency domain, this convolution amounts to repeating the ham spectrum (F) about the various freguencie~ of the : spectrum S~(F). It can then be seen that, in order to avoid losing any information and achieve complete selec-tion of the 90 and 150 Hz frequencies, it is necessary that the frequency width of the spectrum of the weighting window should not be greater than 60 Hz. In the case of . the pre~ent invention, this would imply a theoretical .~ 30 : observation of the signal for 16.6 ms~ - ~ 30 ~z, namely 1 . 2T
T ~ - seconds), namely one half-period of the ILS
~; signal which has a periodicity of 33.3 ms (30 Hz).
In practice, it will however be otherwise.
Indeed, owing to flaws pre~ent in respect of the modula-~ tion and generation of the 90 and 150 Hz signals, the ILS
-~ 40 signal also includes harmonic components of these signal~
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~lU~6~8 and, in particular, as Figure 4 shows, the - harmonic of the 150 Hz (75 ~z) and 2nd harmonic of the 90 Hz (180 Hz). Hence, in order in practice to achieve complete selection of the 90 and 150 Hz useful signals, it will be necessary for the frequency width of the observation window to be not greater than lS Hz, this corres-ponding to a temporal width of 66.6 ms (- - 7.5 Hz ' namely T - - seconds).
Figure S shows, in the form of a fiow diagram, the various operations effected by the processing as~em-bly 4 in order to determine the modulation difference DDM
and sum of modulations SDM parameters characteristic of the analysed ILS signal.
These parameters depend on the modulation factors M90 and M150 which can be determined simply frsm the Fourier transform of the signal S(t).
Indeed, the modulus of S~F) for 90 HZ is equal to HF.M90 HF.M150 and, for lS0 Hz, it is equal to , the cen-2 - ~ 2 tinuous component having modulus HF.
It follows therefore that ¦S(90)1 = ~nd ¦S(150)1- .h~150 or again M90= ~ S(90=HF~R~90)+~90) and M150= ~ S(150))=HF~R~150)~150) R(90), I(90), R(150), I(150) being the real and imaginary parts of the moduli of S(F) at the frequencies 90 and lS0 Hz.
~. . . . .
U~658 It is therefore expedient to determine the~e real R (F) and imaginary I (F) parts in order subsequently to recover the modulation factors ~imply. Now, R(F) and I(F) are given by the following formulae:
..~
R(~ =~S(t)~s2~Ftdt I~=rS(t)sin2~Ftdt -~o Which may also be written, taking into account the sampling and truncation, over the period T, afforded by the weighting window:
R(k) = ~ S(nTe) C09 2 ~knTe I(k) Y 2N S~nTe) ~in 2 ~knTe.
N being the nnmher of samples.taken over the period T, Te being the ~ampling period and n.representing an integer number.
- These calculation~ will bs performed from each - sample of the ILS eignal and from tables stored in the non-volatile memory 14 and each containing a sequence of value~ defining a 90 or 150 ~z co~ine or sine curve.
Figure 6 shows an example .of such a table. The values are stored .in the table in the form o~ 16-bit integer~ and therefore vary between 2'5 and -215. Advan-20 tageou31y, each table, 90 or 150 Hz, will include a quarter sine period. Indeed, starting.at the beginning of the table, a cosine i8 obtained over.n points, whilst starting after the first quarter period, a ~ine is likewise obtained over n point~.
Th~ sampling frequency for the ~ignal S(t) being fixed at 16.2 k~z, namely one acqui~ition of a sample ev~ry 61.728 ~8, the number of s2mples taken over the duration of observation of the ~ignal will be equal to 1080, this duration having been fixed beforahand at two 30 periods of the ILS ~ignal, namely 66.6 ms (15 ~z).
One period of thé 90 ~z signal can then be described exactly on 180 points, (16200/90), the table then containing in total 225 point~ (1 period + 1/4 period). Similarly, one period of the 150 Hz signal can ~lU'~58 be described exactly on 108 points (16200/150), the table then containing a maximum of 135 points.
It may be noted that it i8 possible, by extract-ing from these tables only one point every two or three, to gather information about the 2nd or 3rd harmonic distortion factors existing in respect of the signal, when the latter doe3 not take the form of a signal with ideal spectrum such as that represented in Figure 3a.
Reference i~ again made to Figure 5 which describes the program for calculating the modulation factors M90 and M150. It is a loop synchronized to the 16.2 kHz sampling signal.
After a first step 20, consisting in initializing the various parameters required for the calculations, the synchronization signal controlling the sampling of the signal S(t) is awaited in a second step 21. Upon recep-tion of this command, step 22 undertakes the acquisition - of a first sample S(nTe) with n-l, Te being the sampling - period.
In a next step 23, the value of S(nTe) obtained is multiplied by the value of the weighting window at the corresponding instant Ham (nTe), this last value being extracted from a first table 140 containing predetermined values of the function Ham (t) for each sampling instant.
A first determination of the continuous level HF, then equal to the amplitude A of the signal relating to the first sample acquired, is next undertaken in a step 24.
Step 25 carries out the calculation of the real and imaginary parts of the 90 Hz component of the ILS
signal.
The real part R90 i8 given by multiplying the sampled and weighted value determined at the previous calculation by cos(nTe) and the imaginary part I9O by multiplying this value by sin(nTe), the value- of cos(nTe) and sin(nTe) being extracted from a single 90 Hz table 142.
Step 26 carries out an identical calculation for the 150 Hz component of the ILS signal, the values cos(nTe) and sin(nTe) being extracted from a single 150 Hz table 143.
The number of samples n is next incremented in step 27 and the preceding operations of steps 21 to 26 may be restarted for a new sample and 80 on until comple-tion, at step 28, of the ob~ervation time which isadvantageously l;mited to 66.6 ms, namely exactly 1080 points of the signal S(t).
The modulation factors are then calculated at step 29 through the following formulae already described:
~0= ~ ~90 A ~R~90)~90) and M150= ~ ~150 A JRyl50)+P~150) the ratio A/n giving the average value HF of the signal S(t).
Finally, in step 30, prior to the end-of-proce~s-ing step 31, the difference and sum of the modulations DDM and SDM are determined by differencing or summing the - 15 modulation factors calculated in the preceding step 29.
These data will advantageously be delivered on twelve bits.
It i~ fundamental to note that the sampled signal is not stored in any way, the calculations being effected during the period s~parating two acquisitions of the signal. The method is carried out in real time and therefore require~ no significant memory capacity. Thus, the random-access me ry containing in particular the conversion result may be just 256 bytes.
These calculations having to be carried out between two sampling instants, namely 61.728 ~8, the processing unit will preferably permit calculations on 16 and 32-bit integers as well as in floating point.
Operation at 16 MHz will advantageously be envisaged without this frequency of calculation being l;miting.
Figure 7 describes, with the aid of a flow chart, the analysis of the 1020 Hz identification signal.
Thi~ signal is an audio wave modulated according to morse code and superimpo~ed on the ILS localizing ~ignal.
.
.. .: : ' ' ~lU~6~8 In this case, the signal S(t) will therefore have a spectral component ~lightly different to that of Figure 3b with, in addition, in particular a line at the fre-HF.M1020 S quency of 1020 Hz of amplitude 2 It is therefore possible, by adopting the principles defined for calcula-tion of the 90 and 150 Hz modulation factors, to evaluate the 1020 Hz modulation factor also. However, with the identification signal being able to vary within a range of ~ 50Hz with respect to its nominal value of 1020 Hz, the Hamming window defined earlier no longer allows valued gathering of the amplitude of this frequency line.
It is necessary to define a new weighting window lS which will allow a filtering of the 1020 -Hz frequency line with a pass-band of at least 100 Hz and if po~sible a virtually non-existent ripple. -Figure 8 shows the weighting window developed for filtering the 1020 Hz component.
~ 20 This window ha~ a damped (sin n)/n ~hape result-ing from combining a (sin n)/n function with a Hamming function. The equation of this window function is given by:
WlO~t)=2~ a+(l-~)cos~]
2~t with e~ where T represents the duration of observation T
and advantageously being chosen equal to 0.54 and k~ is equal to 4.
The application of this specific window W1020(t) to the sampled signal S~(t) defines a digital filter whose frequency response is represented in Figure 9, and in which can be noted the absence of ripple in the pas~-band as well a~ the steepness of the edges of this filter, which clearly shows its very high ~electivity.
Reference is again made to Figure 7 which explains the calculation of the modulation factor N1020.
; As for the determination of the M90 and MlS0 factors, tha calculation is carried out in a 16.2 kHz synchronized .: , , ............ , . :
... .. . .
~ - 15- ~lU4~8 loop.
After a first step 40 for initializing variou~
parameters of calculation, the synchronization signal controlling the sampling and therefore the analog/digital conversion of the ILS ~ignal i~ awaited in a ~econd step 41. Having received this conver~ion command, the acqui-sition of the fir~t sample is undertaken in a ~tep 42, and then, in a step 43, the value of this sample i8 multiplied by the value of the aforesaid weighing window W1020 for this sampling instant, this latter value being extracted from a second table 141 containing all the values defining the function W1020(t). A first deter-mination of the continuous level HF is next undertaken in a step 44. The next step-45 carries out the calculation of the real and imaginary parts of the 1020 Hz component via the product of the sampled and weighted value times values extracted from a single 1020 Hz table 144 deliver-ing cosine and sine values for each sampling instant. A
~ new evaluation is then possible after passing, in step 46, to a next ~ample, these calculations being performed throughout the duration of observation, that is to say over 1080 points (step 47).
The modulation factor can then be calculated, in step 48, from the values of R(1020) and I(1020) and from the continuoue component HF, ~tep 49 terminating the processing of the 1020 Hz component. The ~020 Hz modu-lation factor is, like the values DDM and SDM, advanta-geously delivered on 12 bits.
It may be noted that the contents of the ~in/cos 1020 table (and likewi~e for the 90 and 150 Hz tables) can be accessed very easily by using two pointers, one moving cyclically over the fir~t n points of the table, thus defining the cosine function, and the other movinq cyclically over n points starting from the first quarter period of this table, in order to define the sine function.
Contrary to the 90 and 150 Hz tables from which each period can be defined with an integer multiple of the sampling period, one period of the 1020 ~z table is .
: ' ' ' ,:
: ' 4 ~ ~ 8 not equal to an integer number times this sampling period (16200/1020-15.882). Hence, the closest integer, namely 16, will be cho~en to define one period of the identi-fication signal, this in practice reducing the frequency of this signal to 1012.5 Hz, a value which still lies within the admissible range of variation (1020 ~ 50 Hz).
This results in the 1020 Hz table containing 16 + 4 namely 20 points in total.
It is clear that the structure thus described allows very accurate calculations, all the more 80 since these calculations are synchronized with the signal via the phase-lock loop synchronization assembly. Constant positioning on the maximum of the main lobes is thus a8BUred a8 i8 therefore, correspondingly, the avoidance of any amplitude error prejudicial to tho determination of the modulation factors and hence to the definition of the tracking and glide paths.
Furthermore, the accuracy in the determination of the parameters DDM and SDM is heightened further through the fact that the measurement is carried out from the real modulation factors defined, like the ratio of the amplitudes of the modulating signals, with respect to the amplitude of the carrier, and not by regarding the latter amplitude as constant and then evaluating solely the amplitude of the modulating signals, as was customary in the prior art methods, in particular in the patents described at the start of the description.
.
~, , , . :
. . ~ , .. . .. , , ,, ~ :. , :
, ' ' '' '.
R(k) ~ R(k) + S(nTe) x cos 2~k nTe and I(k) - I(k) + S(nTe) x sin 2~k nTe d) repetition of steps a) to c) for all of the ~amples over a duration of observation T of the ILS signal, e) calculation of the modulation factors for the charac-teristic component~ of the ILS signal.
M O = ~ .
HF being the average value of S(nTe) calculated over all the samples taken.
f) calculation of the difference (DDM) and sum (SDM) of the modulations (M (90) and M~150)) of the 90 and 150 Hz - 15 components of the ILS signal:
-SDM ~ M(90) + N(150) and DDM ~ M(90) - M(150).
By performing the above ¢alculation~ on numerous samples, advantageously 540, the effect of the analog or guantization noise becomes virtually nil. Moreover, measurement of the amplitude of each component of the signal is effected selectively about each frequency, this i having the effect of further limiting the wide band noise.
In a particular embodiment, it iB possible to take just one sample of the signal to be analysed every second or third value from the table~ of the cosine function, thereby making it possible to determine also the amplitude of the 2nd or 3rd harmonic components of this signal.
Other characteristics and advantages of the present invention will emerge better on reading the following description given, by way of non-llm;ting illustrative example, in connection with the attached drawing~ in which:
- Figure 1 shows the structure of an ~LS signal analysis device according to the invention, . - -,- '' ,' ' ' .
.., :- -. : ' . . .
.. .: . :
:, ,: : ~,, :.
..
` ` ~lU4~8 - 5 - Figures 2a to 2e, 3a to 3e and 4 are a graphical approximation making it possible to observe the trans-formations carried out on the ILS signal in the time domain and in the frequency domain as well as the rela-tions between these two domains, - Figure 5 is a flow chart describing the digital pro-cessing of the ILS signals in the device according to the invention, - Figure 6 ~hows one form of the signal stored in the tables of the device and utilized in the digital process-ing described in Figure 4, - Figure 7 i8 a flow chart describing the processing of the identification signal in the device according to the invention, --- Figure 8 represents, in the time domain, a ~weighting window" function employed in the processing of the identification signal, and - - Figure 9 represents the frequency r~ponse of the - digital filter corresponding to the weighting window of Figure 8.
It i~ known that, in an instrument landing system, two transmitters are employed; one, at the frequency 110 MBz, known as the localizing signal or ~localizer~ and emitting a beam 2.5 wide on either side of the axis of the landing ~trip, and the other, at the frequency 332 MBz, known as the "glide-path~ signal and emitting a be~m 0.7 wide about a-typical 3 glide path.
In practice, these beams each consist of the zone common to two low-frequency modulated main beam~, one at 90 Hz the other at 150 Bz. The resulting ILS signal therefore in theory assumes the form of a high-frequency signal (110 or 332 MHz) amplitude modulated by two 90 and 150 Hz sinusoidal voltages.
In the particular case of the localizing ILS
signal, a 1020 Bz identification signal is furthermore superimposed on the 90 and 150 Bz components.
Figure 1 is a preferred illustrative embodiment of a device for proce3sing such ILS signals, according to the invention.
:
:
6 ~ 8 ~ his ILS signal constitutes an input ~ignal, on the one hand, for an anti-alia~ing low-pass filter 1 and, on the other hand, for phaso-locking moans consisting of a phase-lock loop (PLL)2 who~e output is connected to a frequency synthesizer 3. Output from the low-pa~s filter and from the synthesizer 3 constitute inputs for a digital proces~ing assembly 4.
In a known manner, a phase-lock loop con~ists of a phase comparator who~e output controls a voltage-controlled oscillator (VCO) which in turn feeds back tothe input of the comparator. In the context of the pre~ent invention, the synchronizing of thi~ circuit is carried out, not with a fundamental frequency of the analysed signal, as is frequently done, but with a sub-harmonic of this signal.-In this-instance, this is the 3rd sub-harmonic of the 90 Hz signal or the 5th sub-harmonic of the 150 Hz signal, that is to say the 30 ~z - frequency.
~Likewise in a known manner, a frequency synthe-sizer takes the form of a phase-lock loop into which a programmable divider is inserted within the feedback circuit.
In the example of the invention, the programmable divider is a divider by 540 making it pos~ible to obtain a frequency of 16.2 k~z at the output of the synthe~izer ! 3 from the stable frequency of 30 Hz provided by the pha~e-lock loop 2. This sampling frequency i8 greater than the highest frequency of the components of the signal to be analysed. Advantageously, in a ratio at least 1 to 10 for the identification signal and at least 1 to 100 for the 150 ~z component.
The processing as~embly 4 includes an analog/digital converter 10 which receives the ILS signal from tho anti-aliasing filter 1 and deliver~ on 10 bits a ~ampled signal S*(t) clocked by a sampling signal delivered by an input/output module 11. The result of this conversion iB 3tored in a random-acces~ memory 12 which is connected to a processing unit 13, which is al~o connected to a non-volatile memory 14.
. ~ . : - . :
.
.. . ~ .
.
.
- : :. ~: .
. .
~ ~ u ~ 8 _ 7 _ The input/output module 11 which receive~ the 16.2 kHz synchronization ~ignal generated by the fre-quency synthesizer 3 i8 connected to the proce~ing unit 13 and also deliver~ signals for exploitation of the analysed ILS signal.
The transfers of information between the proce~s-ing unit and its a~oeiated circuits, memories 12, 14 or input/output module 11, are advantageously carried out on 16 bit~.
The non-volatile memory contains a program for manaqing all the processing as well as several tables of values, the utility of whieh will emerge hereafter, in conneetion with Figures 2 to 9 which explain the oper-ation of the proceseing unit according to the invention.
Figuree 2 and 3 show the various operations performed, by the processing as~embly, on the signal both in the time domain and in the frequency domain.
- Figure 2a shows the ILS signal on input to the deviee aecording to the invention. By way of in~truetion, the analysed signal i8 the loealizing signal whieh contains only the 90 Hz and 150 Hz eomponents to the exclusion of the 1020 Hz identification signal. However, it i8 obvious that a comparable analysis may be condueted with the loealizing ~y~tem whose demodulated signal would then also yield this 1020 Hz signal. Furthermore, the ; ~ignal will firstly be assumed to be complete, without harmonies.
The form of the demodulated signal is given by the following equation:
S(t) ~ HF (1 + M90 ~in 2~.90.t + M150 sin 2~.150.t) with HF the continuous eomponent refleeting the 332 M~z high-frequeney carrier on which the ~um of the 90 Hz and 150 Hz waves is ~uperimpo~ed, N90 the modulation faetor for the 90 Hz wave, M150 the modulation faetor for the 150 Hz wave.
These modulation factors serve to reconstitute the difference of the modulations (DDM) and the ~um of the modulations (SDM), which parameters, in the ca~e of the former, characterizea the position of the aircraft , , , , ' .
~lU~6~8 with respect to the glide path (tracking path) and, in the case of the latter, serves in the validation of this previou~ measurement.
The sampling of the signal S(t~ at a sampling frequency Fe - 1/Te i8 achieved by multiplying thi~
~ignal by a Dirac comb ~(t) of period Te (see Figure 2b) namely S*(t) ~ S(t)x~(t) (Figure 2c), S* being a seguence of values of S(t) taken at the instants nTe.
In the frequency domain, this sampling corre-sponds to a convolution between the signal S(F) (Figure3a) and the signal ~(F) (Figure 3b), which gives a spectrum S*(F) which is the repeat of S~F) about the multiple frequencies of Fe (see Figure 3c).
It can then~ be seen that in order to sample, without lo~ing information, this signal S(t) who~e spectrum i~ bounded by a maximum freguency Fm; it is absolutely neces~ary for the sampling frequency Fe to be - greater than 2Fm. -~However, it must be certain that the spectrum of S(t) is bounded by this value Fm, now, this signal is subjected to fnst variatione and multiple interference depending on the configuration of the strip at the time of landing. This is why, in order to avoid these disad-vantages, the sampling ha~ been preceded by a low-pass filter which necessarily limits the spectrum to Fm, thus avoiding any unde~ired disturbances; this is the anti-~; aliasing filter 1. The frequency spectrum output by this filter is identical to that of S(t) and it is this signal, and not S(t), which i~ in reality the subject of the aforesaid sampling.
Figures 2d and 3d repressnt respectively in thetime domain and in the frequency domain a weighting function known as a Hamming window.
The-sampling may not be carried out over the whole of the signal S(t), that is to say for an infinite number of samples corresponding to an infinite ~ignal observation time. The number of sample~ must be finite.
This limitation reduces the period of observ~tion to a value T, this being eguivalent to multiplying the sampled :
" ' , . ~ . ..
.. . . .. .
- , . . ,' . ' " .
, .
6 ~ 8 _ 9 _ function S*(t) by a window function of width T. So as to reduce the effects due to this truncation, it i~ advis-able to resort to a specific window such as the Ha~ming window which, in respect of its frequency spectrum, ha~
secondary side lobes of very small amplitudes, the main lobe having a witth of 1/T.
Thi~ window function is defined by the following equation:
t ham(t) - a + (1-) C08 (2n T
Preferably, the value 0.54 will be chosen for a, without however being limited to this value alone (the value a-0.5 which corresponds to a Hann window could, for example, also be envisaged).
Figures 2e and 3e show the shape of the signal S(t) after sampling and truncation and that of the j~ corrosponding spectrum.
~~ The latter i8 given by the convolution of the spectrum of the s~mpled signal S~(F) and of the ~pectrum of the ham window (F) namely: S*(F)* ham (F). In the frequency domain, this convolution amounts to repeating the ham spectrum (F) about the various freguencie~ of the : spectrum S~(F). It can then be seen that, in order to avoid losing any information and achieve complete selec-tion of the 90 and 150 Hz frequencies, it is necessary that the frequency width of the spectrum of the weighting window should not be greater than 60 Hz. In the case of . the pre~ent invention, this would imply a theoretical .~ 30 : observation of the signal for 16.6 ms~ - ~ 30 ~z, namely 1 . 2T
T ~ - seconds), namely one half-period of the ILS
~; signal which has a periodicity of 33.3 ms (30 Hz).
In practice, it will however be otherwise.
Indeed, owing to flaws pre~ent in respect of the modula-~ tion and generation of the 90 and 150 Hz signals, the ILS
-~ 40 signal also includes harmonic components of these signal~
.
.
....... .
. .
.
. .:
., :
~lU~6~8 and, in particular, as Figure 4 shows, the - harmonic of the 150 Hz (75 ~z) and 2nd harmonic of the 90 Hz (180 Hz). Hence, in order in practice to achieve complete selection of the 90 and 150 Hz useful signals, it will be necessary for the frequency width of the observation window to be not greater than lS Hz, this corres-ponding to a temporal width of 66.6 ms (- - 7.5 Hz ' namely T - - seconds).
Figure S shows, in the form of a fiow diagram, the various operations effected by the processing as~em-bly 4 in order to determine the modulation difference DDM
and sum of modulations SDM parameters characteristic of the analysed ILS signal.
These parameters depend on the modulation factors M90 and M150 which can be determined simply frsm the Fourier transform of the signal S(t).
Indeed, the modulus of S~F) for 90 HZ is equal to HF.M90 HF.M150 and, for lS0 Hz, it is equal to , the cen-2 - ~ 2 tinuous component having modulus HF.
It follows therefore that ¦S(90)1 = ~nd ¦S(150)1- .h~150 or again M90= ~ S(90=HF~R~90)+~90) and M150= ~ S(150))=HF~R~150)~150) R(90), I(90), R(150), I(150) being the real and imaginary parts of the moduli of S(F) at the frequencies 90 and lS0 Hz.
~. . . . .
U~658 It is therefore expedient to determine the~e real R (F) and imaginary I (F) parts in order subsequently to recover the modulation factors ~imply. Now, R(F) and I(F) are given by the following formulae:
..~
R(~ =~S(t)~s2~Ftdt I~=rS(t)sin2~Ftdt -~o Which may also be written, taking into account the sampling and truncation, over the period T, afforded by the weighting window:
R(k) = ~ S(nTe) C09 2 ~knTe I(k) Y 2N S~nTe) ~in 2 ~knTe.
N being the nnmher of samples.taken over the period T, Te being the ~ampling period and n.representing an integer number.
- These calculation~ will bs performed from each - sample of the ILS eignal and from tables stored in the non-volatile memory 14 and each containing a sequence of value~ defining a 90 or 150 ~z co~ine or sine curve.
Figure 6 shows an example .of such a table. The values are stored .in the table in the form o~ 16-bit integer~ and therefore vary between 2'5 and -215. Advan-20 tageou31y, each table, 90 or 150 Hz, will include a quarter sine period. Indeed, starting.at the beginning of the table, a cosine i8 obtained over.n points, whilst starting after the first quarter period, a ~ine is likewise obtained over n point~.
Th~ sampling frequency for the ~ignal S(t) being fixed at 16.2 k~z, namely one acqui~ition of a sample ev~ry 61.728 ~8, the number of s2mples taken over the duration of observation of the ~ignal will be equal to 1080, this duration having been fixed beforahand at two 30 periods of the ILS ~ignal, namely 66.6 ms (15 ~z).
One period of thé 90 ~z signal can then be described exactly on 180 points, (16200/90), the table then containing in total 225 point~ (1 period + 1/4 period). Similarly, one period of the 150 Hz signal can ~lU'~58 be described exactly on 108 points (16200/150), the table then containing a maximum of 135 points.
It may be noted that it i8 possible, by extract-ing from these tables only one point every two or three, to gather information about the 2nd or 3rd harmonic distortion factors existing in respect of the signal, when the latter doe3 not take the form of a signal with ideal spectrum such as that represented in Figure 3a.
Reference i~ again made to Figure 5 which describes the program for calculating the modulation factors M90 and M150. It is a loop synchronized to the 16.2 kHz sampling signal.
After a first step 20, consisting in initializing the various parameters required for the calculations, the synchronization signal controlling the sampling of the signal S(t) is awaited in a second step 21. Upon recep-tion of this command, step 22 undertakes the acquisition - of a first sample S(nTe) with n-l, Te being the sampling - period.
In a next step 23, the value of S(nTe) obtained is multiplied by the value of the weighting window at the corresponding instant Ham (nTe), this last value being extracted from a first table 140 containing predetermined values of the function Ham (t) for each sampling instant.
A first determination of the continuous level HF, then equal to the amplitude A of the signal relating to the first sample acquired, is next undertaken in a step 24.
Step 25 carries out the calculation of the real and imaginary parts of the 90 Hz component of the ILS
signal.
The real part R90 i8 given by multiplying the sampled and weighted value determined at the previous calculation by cos(nTe) and the imaginary part I9O by multiplying this value by sin(nTe), the value- of cos(nTe) and sin(nTe) being extracted from a single 90 Hz table 142.
Step 26 carries out an identical calculation for the 150 Hz component of the ILS signal, the values cos(nTe) and sin(nTe) being extracted from a single 150 Hz table 143.
The number of samples n is next incremented in step 27 and the preceding operations of steps 21 to 26 may be restarted for a new sample and 80 on until comple-tion, at step 28, of the ob~ervation time which isadvantageously l;mited to 66.6 ms, namely exactly 1080 points of the signal S(t).
The modulation factors are then calculated at step 29 through the following formulae already described:
~0= ~ ~90 A ~R~90)~90) and M150= ~ ~150 A JRyl50)+P~150) the ratio A/n giving the average value HF of the signal S(t).
Finally, in step 30, prior to the end-of-proce~s-ing step 31, the difference and sum of the modulations DDM and SDM are determined by differencing or summing the - 15 modulation factors calculated in the preceding step 29.
These data will advantageously be delivered on twelve bits.
It i~ fundamental to note that the sampled signal is not stored in any way, the calculations being effected during the period s~parating two acquisitions of the signal. The method is carried out in real time and therefore require~ no significant memory capacity. Thus, the random-access me ry containing in particular the conversion result may be just 256 bytes.
These calculations having to be carried out between two sampling instants, namely 61.728 ~8, the processing unit will preferably permit calculations on 16 and 32-bit integers as well as in floating point.
Operation at 16 MHz will advantageously be envisaged without this frequency of calculation being l;miting.
Figure 7 describes, with the aid of a flow chart, the analysis of the 1020 Hz identification signal.
Thi~ signal is an audio wave modulated according to morse code and superimpo~ed on the ILS localizing ~ignal.
.
.. .: : ' ' ~lU~6~8 In this case, the signal S(t) will therefore have a spectral component ~lightly different to that of Figure 3b with, in addition, in particular a line at the fre-HF.M1020 S quency of 1020 Hz of amplitude 2 It is therefore possible, by adopting the principles defined for calcula-tion of the 90 and 150 Hz modulation factors, to evaluate the 1020 Hz modulation factor also. However, with the identification signal being able to vary within a range of ~ 50Hz with respect to its nominal value of 1020 Hz, the Hamming window defined earlier no longer allows valued gathering of the amplitude of this frequency line.
It is necessary to define a new weighting window lS which will allow a filtering of the 1020 -Hz frequency line with a pass-band of at least 100 Hz and if po~sible a virtually non-existent ripple. -Figure 8 shows the weighting window developed for filtering the 1020 Hz component.
~ 20 This window ha~ a damped (sin n)/n ~hape result-ing from combining a (sin n)/n function with a Hamming function. The equation of this window function is given by:
WlO~t)=2~ a+(l-~)cos~]
2~t with e~ where T represents the duration of observation T
and advantageously being chosen equal to 0.54 and k~ is equal to 4.
The application of this specific window W1020(t) to the sampled signal S~(t) defines a digital filter whose frequency response is represented in Figure 9, and in which can be noted the absence of ripple in the pas~-band as well a~ the steepness of the edges of this filter, which clearly shows its very high ~electivity.
Reference is again made to Figure 7 which explains the calculation of the modulation factor N1020.
; As for the determination of the M90 and MlS0 factors, tha calculation is carried out in a 16.2 kHz synchronized .: , , ............ , . :
... .. . .
~ - 15- ~lU4~8 loop.
After a first step 40 for initializing variou~
parameters of calculation, the synchronization signal controlling the sampling and therefore the analog/digital conversion of the ILS ~ignal i~ awaited in a ~econd step 41. Having received this conver~ion command, the acqui-sition of the fir~t sample is undertaken in a ~tep 42, and then, in a step 43, the value of this sample i8 multiplied by the value of the aforesaid weighing window W1020 for this sampling instant, this latter value being extracted from a second table 141 containing all the values defining the function W1020(t). A first deter-mination of the continuous level HF is next undertaken in a step 44. The next step-45 carries out the calculation of the real and imaginary parts of the 1020 Hz component via the product of the sampled and weighted value times values extracted from a single 1020 Hz table 144 deliver-ing cosine and sine values for each sampling instant. A
~ new evaluation is then possible after passing, in step 46, to a next ~ample, these calculations being performed throughout the duration of observation, that is to say over 1080 points (step 47).
The modulation factor can then be calculated, in step 48, from the values of R(1020) and I(1020) and from the continuoue component HF, ~tep 49 terminating the processing of the 1020 Hz component. The ~020 Hz modu-lation factor is, like the values DDM and SDM, advanta-geously delivered on 12 bits.
It may be noted that the contents of the ~in/cos 1020 table (and likewi~e for the 90 and 150 Hz tables) can be accessed very easily by using two pointers, one moving cyclically over the fir~t n points of the table, thus defining the cosine function, and the other movinq cyclically over n points starting from the first quarter period of this table, in order to define the sine function.
Contrary to the 90 and 150 Hz tables from which each period can be defined with an integer multiple of the sampling period, one period of the 1020 ~z table is .
: ' ' ' ,:
: ' 4 ~ ~ 8 not equal to an integer number times this sampling period (16200/1020-15.882). Hence, the closest integer, namely 16, will be cho~en to define one period of the identi-fication signal, this in practice reducing the frequency of this signal to 1012.5 Hz, a value which still lies within the admissible range of variation (1020 ~ 50 Hz).
This results in the 1020 Hz table containing 16 + 4 namely 20 points in total.
It is clear that the structure thus described allows very accurate calculations, all the more 80 since these calculations are synchronized with the signal via the phase-lock loop synchronization assembly. Constant positioning on the maximum of the main lobes is thus a8BUred a8 i8 therefore, correspondingly, the avoidance of any amplitude error prejudicial to tho determination of the modulation factors and hence to the definition of the tracking and glide paths.
Furthermore, the accuracy in the determination of the parameters DDM and SDM is heightened further through the fact that the measurement is carried out from the real modulation factors defined, like the ratio of the amplitudes of the modulating signals, with respect to the amplitude of the carrier, and not by regarding the latter amplitude as constant and then evaluating solely the amplitude of the modulating signals, as was customary in the prior art methods, in particular in the patents described at the start of the description.
.
~, , , . :
. . ~ , .. . .. , , ,, ~ :. , :
, ' ' '' '.
Claims (17)
1. Instrument landing system (ILS) signal analysis device including an analog/digital converter receiving the composite signal to be analysed and delivering a succession of values which can be processed in digital form, characterized in that the said signal to be ana-lysed is furthermore applied to phase-locking means (2, 3) which deliver for the said analog/digital converter (10) a sampling signal of frequency greater than the largest frequency of the components of the signal to be analysed, the digital processing of the values from the converter (10), carried out in real time between two sampling instants, allowing determination of the para-meters for modulation of the ILS signal.
2. ILS signal analysis device according to Claim 1, characterized in that the phase-locking means include a phase-lock loop (2) whose output is connected to the input of a frequency synthesizer(3).
3. ILS signal analysis device according to Claim 2, characterized in that the said phase-lock loop is syn-chronized directly with the frequency of the signal to be analysed, which frequency is extracted from a sub-har-monic of the 90 Hz and 150 Hz components constituting this signal.
4. ILS signal analysis device according to Claim 3, characterized in that the sampling frequency is equal to 16.2 kHz.
5. ILS signal analysis device according to Claim 1, characterized in that the digital processing of the values from the converter is carried out by a processing unit (13) with which are associated a random-access memory (12), a non-volatile memory (14) and an input/output module (11).
6. ILS signal analysis device according to Claim 5, characterized in that the said non-volatile memory includes a first table (140) in which are written numeri-cal values corresponding to the value of a weighting function for each sampling instant.
7. ILS signal analysis device according to Claim 5, characterized in that the said non-volatile memory includes at least two tables (142, 143) in each of which are written numerical values representative, over a quarter period, of a cosine function of frequency equal to that of a characteristic component of the signal to be analysed.
8. Instrument landing system (ILS) signal analysis device including an analog/digital converter receiving the composite signal to be analysed and delivering a succession of values which can be processed in digital form, characterized in that in order to undertake the determination of the modulation factor for the identifi-cation signal contained in the ILS signal, it includes a digital filter using a weighting function whose equation is given by the following relation:
and which carries out an extraction of this signal for identification of the ILS signal delivered by the con-verter, where T represents the duration of observation of the signal to be analysed and k' and .alpha. are specified numerical coefficients.
and which carries out an extraction of this signal for identification of the ILS signal delivered by the con-verter, where T represents the duration of observation of the signal to be analysed and k' and .alpha. are specified numerical coefficients.
9. ILS signal analysis device according to Claim 8, characterized in that the coefficient .alpha. is equal to 0.54 and the coefficient k' is equal to 4.
10. ILS signal analysis device according to Claim 8 or Claim 9, characterized in that the said transfer function is stored in digital form in a table (141) contained in a non-volatile memory (14) of a processing assembly (4) carrying out the digital processing of the values from the analog/digital converter.
11. Instrument landing system (ILS) signal analysis method, characterized in that it includes the following steps:
a) acquisition of a sample S(nTe) of the ILS signal to be analysed by the analog/digital conversion of this signal, Te representing the sampling period and n being an integer number, the sampling frequency being greater than the largest of the frequencies k of the components of the signal to be analysed, b) multiplication of the value of the sample taken by a first numerical value corresponding to the value of a weighting window function at the acquisition instant:
S(nTe) = Fen(nTe) x S(nTe) c) determination, at the frequencies k of the character-istic components of the ILS signal, of the real and imaginary parts of the spectrum of the sampled and weighted signal S(t):
R(k) = R(k) + S(nTe) x cos 2.pi.k nTe and I(k) = I(k) + S(nTe) x sin 2.pi.k nTe d) repetition of steps a) to c) for all of the samples over a duration of observation T of the ILS signal, e) calculation of the modulation factors for the charac-teristic components of the ILS signal.
HF being the average value of S(nTe) calculated over all the samples taken.
a) acquisition of a sample S(nTe) of the ILS signal to be analysed by the analog/digital conversion of this signal, Te representing the sampling period and n being an integer number, the sampling frequency being greater than the largest of the frequencies k of the components of the signal to be analysed, b) multiplication of the value of the sample taken by a first numerical value corresponding to the value of a weighting window function at the acquisition instant:
S(nTe) = Fen(nTe) x S(nTe) c) determination, at the frequencies k of the character-istic components of the ILS signal, of the real and imaginary parts of the spectrum of the sampled and weighted signal S(t):
R(k) = R(k) + S(nTe) x cos 2.pi.k nTe and I(k) = I(k) + S(nTe) x sin 2.pi.k nTe d) repetition of steps a) to c) for all of the samples over a duration of observation T of the ILS signal, e) calculation of the modulation factors for the charac-teristic components of the ILS signal.
HF being the average value of S(nTe) calculated over all the samples taken.
12. ILS signal analysis method according to Claim 11, characterized in that it furthermore includes the follow-ing step:
f) calculation of the difference (DDM) and sum (SDM) of the modulations (M (90) and M(150)) of the 90 and 150 Hz components of the ILS signal:
SDM - M(90) + M(150) and DDM = M(90) - M(150).
f) calculation of the difference (DDM) and sum (SDM) of the modulations (M (90) and M(150)) of the 90 and 150 Hz components of the ILS signal:
SDM - M(90) + M(150) and DDM = M(90) - M(150).
13. ILS signal analysis method according to Claim 11, characterized in that the said weighting window function is a Hamming function.
14. ILS signal analysis method according to Claim 11, characterized in that the said window function is a function corresponding to the following equation:
T being the duration of observation, k' and .alpha. being specified constants.
T being the duration of observation, k' and .alpha. being specified constants.
15. ILS signal analysis method according to Claim 14, characterized in that the coefficient a is equal to 0.54 and the coefficient k' equal to 4.
16. ILS signal analysis method according to Claim 11, characterized in that in order to undertake the determi-nation of the harmonics of order r of the characteristic components of the ILS signal, the calculation of the real and imaginary parts of the spectrum of the weighted signal S?(t) carried out at step c) is modified as follows:
R(k) = R(k) + S(nTe) x cos 2.pi.krnTe I(k) = I(k) + S(nTe) x sin 2.pi.krnTe
R(k) = R(k) + S(nTe) x cos 2.pi.krnTe I(k) = I(k) + S(nTe) x sin 2.pi.krnTe
17. ILS signal analysis method according to any one of Claims 11 to 14, characterized in that the said functions cos 2.pi.knTe and sin 2.pi.knTe are stored in the form of tables of values, n varying from 1 to N, N being the total number of samples taken over the duration of observation T.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| FR9210290 | 1992-08-26 | ||
| FR9210290A FR2695211B1 (en) | 1992-08-26 | 1992-08-26 | Device and method for analyzing ILS signals. |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2104658A1 true CA2104658A1 (en) | 1994-02-27 |
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ID=9433030
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002104658A Abandoned CA2104658A1 (en) | 1992-08-26 | 1993-08-23 | Ils signal analysis device and method |
Country Status (5)
| Country | Link |
|---|---|
| US (1) | US5483245A (en) |
| CA (1) | CA2104658A1 (en) |
| DE (1) | DE4328269A1 (en) |
| FR (1) | FR2695211B1 (en) |
| GB (1) | GB2270226A (en) |
Families Citing this family (23)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FR2731799B1 (en) * | 1995-03-13 | 1997-06-20 | Concept Aero System Sarl Soc | METHOD AND DEVICE FOR ANALYZING RADIONAVIGATION SIGNALS |
| US6061551A (en) * | 1998-10-21 | 2000-05-09 | Parkervision, Inc. | Method and system for down-converting electromagnetic signals |
| US7515896B1 (en) | 1998-10-21 | 2009-04-07 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same, and aperture relationships |
| US7236754B2 (en) * | 1999-08-23 | 2007-06-26 | Parkervision, Inc. | Method and system for frequency up-conversion |
| US7039372B1 (en) * | 1998-10-21 | 2006-05-02 | Parkervision, Inc. | Method and system for frequency up-conversion with modulation embodiments |
| US6370371B1 (en) | 1998-10-21 | 2002-04-09 | Parkervision, Inc. | Applications of universal frequency translation |
| US6542722B1 (en) * | 1998-10-21 | 2003-04-01 | Parkervision, Inc. | Method and system for frequency up-conversion with variety of transmitter configurations |
| US6879817B1 (en) * | 1999-04-16 | 2005-04-12 | Parkervision, Inc. | DC offset, re-radiation, and I/Q solutions using universal frequency translation technology |
| US6853690B1 (en) * | 1999-04-16 | 2005-02-08 | Parkervision, Inc. | Method, system and apparatus for balanced frequency up-conversion of a baseband signal and 4-phase receiver and transceiver embodiments |
| US6873836B1 (en) * | 1999-03-03 | 2005-03-29 | Parkervision, Inc. | Universal platform module and methods and apparatuses relating thereto enabled by universal frequency translation technology |
| US7110435B1 (en) * | 1999-03-15 | 2006-09-19 | Parkervision, Inc. | Spread spectrum applications of universal frequency translation |
| US7693230B2 (en) * | 1999-04-16 | 2010-04-06 | Parkervision, Inc. | Apparatus and method of differential IQ frequency up-conversion |
| US7110444B1 (en) * | 1999-08-04 | 2006-09-19 | Parkervision, Inc. | Wireless local area network (WLAN) using universal frequency translation technology including multi-phase embodiments and circuit implementations |
| US7065162B1 (en) | 1999-04-16 | 2006-06-20 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same |
| US8295406B1 (en) | 1999-08-04 | 2012-10-23 | Parkervision, Inc. | Universal platform module for a plurality of communication protocols |
| US7010286B2 (en) | 2000-04-14 | 2006-03-07 | Parkervision, Inc. | Apparatus, system, and method for down-converting and up-converting electromagnetic signals |
| US7454453B2 (en) * | 2000-11-14 | 2008-11-18 | Parkervision, Inc. | Methods, systems, and computer program products for parallel correlation and applications thereof |
| US7072427B2 (en) * | 2001-11-09 | 2006-07-04 | Parkervision, Inc. | Method and apparatus for reducing DC offsets in a communication system |
| US7460584B2 (en) * | 2002-07-18 | 2008-12-02 | Parkervision, Inc. | Networking methods and systems |
| US7379883B2 (en) | 2002-07-18 | 2008-05-27 | Parkervision, Inc. | Networking methods and systems |
| JP4420223B2 (en) * | 2004-11-16 | 2010-02-24 | 信越化学工業株式会社 | Method for producing sulfide chain-containing organosilicon compound |
| JP4386189B2 (en) * | 2004-11-16 | 2009-12-16 | 信越化学工業株式会社 | Method for producing sulfide chain-containing organosilicon compound |
| CN115508867B (en) * | 2022-09-07 | 2023-06-09 | 湖北第二师范学院 | GNSS-R receiver double-antenna signal collaborative correlation processing system and method |
Family Cites Families (8)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| FR1487441A (en) * | 1966-05-25 | 1967-07-07 | Artus | Method for measuring the modulation difference of a complex signal |
| DE2852506C2 (en) * | 1978-12-05 | 1980-09-25 | Standard Elektrik Lorenz Ag, 7000 Stuttgart | Monitoring device for the instrument landing system |
| US4301455A (en) * | 1979-10-17 | 1981-11-17 | United States Of America | Groundspeed measurement system |
| US4414632A (en) * | 1981-04-07 | 1983-11-08 | Murrell Robert A | Automatic monitoring system for radio signal |
| US4609921A (en) * | 1983-11-09 | 1986-09-02 | Rockwell International Corporation | Digital modulation depth detector |
| US4586049A (en) * | 1984-06-25 | 1986-04-29 | Davidson Eldon F | Apparatus and method for use in instrument landing system localizer and glide-slope receivers for deriving course deviation information |
| FR2591048B1 (en) * | 1985-11-29 | 1988-01-15 | Trt Telecom Radio Electr | DIGITAL FREQUENCY DISCRIMINATOR, AND APPLICATION TO A VOR COMPOSITE SIGNAL |
| FR2596547B1 (en) * | 1986-03-28 | 1992-07-03 | Socrat Sa Nouvelle Societe | DEVICE FOR PROVIDING RADIONAVIGATION INFORMATION WITH INDICATION OF VALIDITY |
-
1992
- 1992-08-26 FR FR9210290A patent/FR2695211B1/en not_active Expired - Fee Related
-
1993
- 1993-08-16 GB GB9317009A patent/GB2270226A/en not_active Withdrawn
- 1993-08-23 DE DE4328269A patent/DE4328269A1/en not_active Ceased
- 1993-08-23 CA CA002104658A patent/CA2104658A1/en not_active Abandoned
- 1993-08-24 US US08/110,932 patent/US5483245A/en not_active Expired - Fee Related
Also Published As
| Publication number | Publication date |
|---|---|
| FR2695211A1 (en) | 1994-03-04 |
| US5483245A (en) | 1996-01-09 |
| GB9317009D0 (en) | 1993-09-29 |
| GB2270226A (en) | 1994-03-02 |
| FR2695211B1 (en) | 1994-11-18 |
| DE4328269A1 (en) | 1994-03-03 |
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