CA2161901C - A method for a transmitter to compensate for varying loading without an isolator - Google Patents
A method for a transmitter to compensate for varying loading without an isolator Download PDFInfo
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- CA2161901C CA2161901C CA002161901A CA2161901A CA2161901C CA 2161901 C CA2161901 C CA 2161901C CA 002161901 A CA002161901 A CA 002161901A CA 2161901 A CA2161901 A CA 2161901A CA 2161901 C CA2161901 C CA 2161901C
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- gain
- transmitter
- reflected energy
- loop gain
- overall loop
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Classifications
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
- H03F1/3294—Acting on the real and imaginary components of the input signal
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
- H03F1/3247—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits using feedback acting on predistortion circuits
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/34—Negative-feedback-circuit arrangements with or without positive feedback
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/34—Negative-feedback-circuit arrangements with or without positive feedback
- H03F1/345—Negative-feedback-circuit arrangements with or without positive feedback using hybrid or directional couplers
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
- H03F3/24—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers of transmitter output stages
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/57—Separate feedback of real and complex signals being present
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- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/02—Transmitters
- H04B1/04—Circuits
- H04B2001/0408—Circuits with power amplifiers
- H04B2001/0416—Circuits with power amplifiers having gain or transmission power control
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/02—Transmitters
- H04B1/04—Circuits
- H04B2001/0408—Circuits with power amplifiers
- H04B2001/0433—Circuits with power amplifiers with linearisation using feedback
Abstract
A transmitter (100) that includes an amplifying element (107), an antenna (109), a gain stage (104, 105), and a closed loop feedback may compensate for varying antenna loads without an isolator. This may be accomplished by determining the effects of the varying loading on overall loop gain. Knowing the effects, the transmitter adjusts the gain of the gain stage to maintain a constant overall loop gain, thus eliminating the need for an isolator.
Description
WO 95/01009 21~ ~. 9 01 PCT/US94/06268 A Method For A Transll~iLler To Co~ e.-~te For Varying Loading Without An Isolator Field of the Invention This invention relates generally to radio frequency tr~n~mitters and, in particular, to a linear radio frequency 15 tr~n~ .or having a varying ~nt~nn~ load.
Background of the Invention As is known, radio frequency (RF) tr~nsmitters modulate
Background of the Invention As is known, radio frequency (RF) tr~nsmitters modulate
2 0 baseband si~n~l~, such as analog voice or digital voice samples, onto an RF carrier, amplify the RF carrier, and tr~n~mit the RF carrier, via an ~ltell,la, through the air as ele.,llo,.-~gnetic energy. The electrom~gnPtic energy is subsequently received by a receiver's e.-l.~, demodulated back to the baseband signal, and rendered 2 5 audible (if voice was tr~n~mitted) by the receiver.
As is also known, many co.. l.. ic~tion systems, such as cellular telephone and trlmking, utilize spectrally efficient modulation techniques, such as quadrature amplitude modulation
As is also known, many co.. l.. ic~tion systems, such as cellular telephone and trlmking, utilize spectrally efficient modulation techniques, such as quadrature amplitude modulation
3 0 (QAM) and quaternary phase shift keying (QPSK) in a time division mllltiple access (TDMA) format. These spectrally efficient modulation techniques typically correlate the baseband signal to changes in RF carrier amplitude and phase via a digital symbol constellation format such as the QAM format, which is described in 3 5 co-pending United States Patent application serial number 2l6l90l 2 07/783,289, entitled "Commnnication Signal Having A Time Domain Pilot Component", assigned to Motorola Inc. Since the spectrally efficient modulation techniques require variation of the RF carrier amplitude, a linear class A or class AB amplifier must be used. If 5 thè amplifier is not linear, it provides unwanted RF energy, or splaKer, at frequencies adjacent to the RF carrier. This splatter may subsequently interfere with two-way communications in process on the adjacent frequencies, or rh~nnels.
Linearity of an amplifier is affected by the varying loads presented by the alltenlla of the tr~n~mitter. Typically, an ~ntçnn~ is designed to provide a fixed load, fifty ohms for example. However, due to reflected energy received, the load changes.
To minimi~e variations in lo~lin~, tr~n~mitters generally incorporate isolators to provide a subst~nti~lly con~t~nt load impedance to the amplifier. The isolator incllldes a circulator and a termin~tin~ impedance which is typically fifty ohms. The circulator is a three tçrmin~l device that provides directir.n~l flow of t~he RF
2 0 energy from the amplifier to the ~ and from the ~nte~n~ to the termin~ting impe~3~nce. Therefore, the RF energy sourced by the amplifier is provided to the ~--t~ and any RF energy elltelillg the antenna is absorbed in the tçrmin~tin~ impe~l~nce. Thus, the isolator presents a con~t~nt impedance to the amplifler.
Although the isolator provides a conct~nt load irnpedance to the RF amplifier, size, cost, and bandwidth limit~tions typically inhibit the use of a universal isolator in mobile radios, portable radios, and cellular telephones. For example, a radio that operates at 3 0 132 MHz requires an isolator that has a volume of 8.19 cubic ce..~ eters (0.5 cubic inches), weighs 227 grams (0.5 pounds), and costs $30 each in qll~ntities of 100,000 per year. As a result, an isolator puts obvious constraints on the design of such radios.
Additionally, isolators have fixed bandwidths; therefore, multiple 3 5 isolators may be required in tr~n~mitters which operate over wide wo gS/oloog ~1619 01 PCT/US94/06268 ~.
frequency ranges. This bandwidth limitation is most noticeable at lower RF carrier frequencies, such as VHF, where the allocated frequency band covers a large ~e-cel.t~e bandwidth.
5 To avoid the use of the isolator, existing frequency modulation (FM) transmitters, which inclllde nollline~r amplifiers, utilize protective fee~lb~clc circuitry. The protective feedback monitors the voltage standing wave ratio (VSWR) at the nonline~r amplifier's output and correspondingly reduces the amount of output power 10 provided by the nonlinP~r amplifier to the ~ e~ This approach generally reduces the nonline~r amplifier's output power by a fixed arnount when the VSWR exceeds a predete~ l level. For example, when a 3:1 VSWR is letected at the nonline~r amplifier's output, the nonline~r amplifier's output power may be reduced by 3 dB. This approach works for nonline~r amplifiers, but does not include any provisions for m~ g amplifier linearity.
Therefore, a need exists for a method to co~ te for effects of varying lo~ling without the need for an isolator in a 2 0 tr~n~mitter that has a linear amplifier.
Brief Description of the Drawings 2 5 FIG. 1 illustrates a block diagram depiction of a linear transmitter in accordance with the present invention.
FIG. 2 illustrates a flow diagram of steps executed by a transmitter in accordance widl the present invention.
Description of a Prefe~d Embo~lim~o-nt Generally, the present invention provides a method for a tr~n~mitter to compensate for ~..te~ load variations without using of an isolator (circulator plus resistive termin~tion). This is accomplished by determinin~ the effects that the load variations have 5 on the loop gain of the tr~n~ 's fee-lb~rl~ loop. Upon detel,..i..;l.~; the effects, the tr~n~mitter adjusts either the gain of a variable gain stage within the feedback loop or a signal source level to m~int~in the transmitter's linearity requirement.
The present invention may be more fully described with refel~llce to FIGs. 1 and 2. FIG. 1 illustrates a block diagram of a tr~n~mitter 100 that includes a signal source 101, two variable gain stages 104 and 105, a frequency up-converter 106, an amplifying elçment 107, a sampler 108, an ~ 109, a feedback element 111, l S and a reflected energy representer 114. In a quadrature amplitude modulation (QAM) sch~me, the signal source 101 is typically a digital signal processor (DSP) that generates in-phase (I) and quadrature (Q) signal represent~tions of digital data and processes these I and Q signal represent~tion~ into b~eb~n~ analog waveforms.
2 0 Sllmmin~ junctions 102 and 103 receive the baseband analog I and Q
waveforms, respectively, and subtract them from the baseband outputs of the feedback elem~ont 111. The resultant waveforms proceed to their respective gain stages 104 and lOS for amplification and continue to the frequency up-converter 106 for tr~n~l~tion to a 2 5 selected radio frequency (RP) carrier. The amplifying elçment 107 amplifies the RF carrier and sllbmit~ the amplified RF carrier to the sampler 108 and the ~ e....~ 109 for radio tr~n~mi~sion.
The sampler 108 samples the amplified RF signal provided by 3 0 ~e amplifying element 107 and supplies forward and reflected signal samples 112 and 113 to the reflected energy representer 114 and supplies forward signal samples 112 to the feedback elçmtont 111.
The feedback element 111, which may incl~ e a frequency down-converter, gain stages, and loop phase adjustment circuits, receives 3 5 the forward signal sample 112, frequency down-converts it by reversing the processing order of the frequency up-converter 106, and produces baseband analog I and Q data representations. These baseband I and Q sigT~ provide the inverting inputs to the sllmming junctions 102 and 103, thus completing a feedback path. Providing 5 closed loop feedback in such tr~n.~mitters is known, thus no further discussion will be presente(l except to facilitate the underst~n~lin~ of this invention. The refl~cte~ energy representer 114 accepts the fol~vald and reflected signal samples 112 and 113 to generate represent~tions of the reflected energy based on its input signal 10 samples. As let~ile(l below, the represent~tions of the reflected energy contribute to dete....;..;.~ the gains of the two variable gain stages 104 and 105 and the signal levels of the I and Q signal representations generated by the signal source 101.
The functional blocks of the tr~ncmitter 100 incl~e a variety of circuit topologies. The gain stages 104 and 105 are typically variable high gain, low frequency amplifiers whose operating bandwidth is restricted to the loop bandwidth required for proper operation of the negative fee~b~ loop, thereby encomp~sing the 2 0 bandwidth of the data mo~ tion sigr~l~ supplied by the signal source 101. The frequency up-converter 106 inc]ll~les mixers 116 and 120, a local oscillator 117, a 90 degree phase shifter 118, and a signal combiner 119. The frequency tr~ncl~tion from baseband to the RF carrier frequency is achieved in the frequency up-converter 2 5 106 by mixing the baseband I signal waveform with the local oscillator frequency, mixin~ the b~ceb~n~l Q signal waveform with a 90 degree phase shifted version of the local oscillator frequency, and ~IIIIIIIIil~g the two mixer outputs to produce indepçn~lent I and Q data signals at the common RF carrier frequency. The amplifying 3 0 element 107 is an RF power amplifier that provides power amplification of the RF carrier frequency, which cont~in~ the up-converted data si n~l~, and is commonly formed by cascading several amplifier stages or transistors. Since the envelopes of the analog I
and Q waveforms are amplitude depen~lent, the amplifying el~ment 3 5 107 is a linear, class A or class AB, amplifier as opposed to a constant envelope, or class C, nonline~r amplifier. The sampler 108 is preferably a directional coupler that provides scaled representations of forward and reflecte-l voltage, or current, waveforms present at the ~lte,l~la 109.
The sampler 108, feedback loop elemP,nt 111, and sllmming junctions 102 and 103 are configured to e~h~nce the linear amplifying properties of the amplifying elemto,nt 107. As briefly mentioned above, these el~,me,nt~ form a negative feedback system 10 which reduces power levels of ullwa~lted sign~l~ which fall outside the ~si~ed frequency band. These ullwallled signals result from nonlinç~rities in the amplifying el~,m~,nt 107 and may cause intelrelcllce in adjacent RF ch~nnels, thus dislu~lillg commlmic~tions on those CllA~ . The ullw~ ed si n~ in the adjacent RF r.h~nn~.ls 1 5 are known as splatter. For a time division mllhirle access (TDMA) system, the adjacent ch~nnel splatter specification may be in the -60 dBc range when comparing the signal powers of the signal in an cçnt ~1IAII~ to the RF carrier sign~l This specification is typically achieved by lltili7.in~ an amplifying elem~nt 107 whose 2 0 linearity provides adjacent rh~nnP,l splatter performance of -30 dBc and improving the adjacent ch~nnel splatter by 30 dB using known Cartesian fee~b~ck correction technillues.
Effects of varying lo~ling at the output of the amplifying 2 5 eleme.nt 107 due to changes in All~ellll~ 109 impe-l~nce are dete~nine~l by ei~er measuring ~e gain changes of the amplifying element 107, detenninin~ represe.~t~tions of the lo~tlin~ via reflected energy received by the antenna 109, or both. To measure the gain variation of the amplifying elP-m~.nt 107, a training signal 115, such 3 0 as a signal discussed in United States Patent number 5,066,923, entitled "T,ine~r Tr~n.cmitter Training Method And Apparatus" and ~signed to Motorola, is input from the signal source 101 to point B
122 of FIG. 1 when the variable gain stages 104 and 105 are inactivated. Inactivating the variable gain stages 104 and 105 3 5 effectively opens the fol~ rd path of a fee~lb~c.k loop within the wo gs/oloog ~ fil ~ PCT/US94106268 tr~n.cmitter 100. The training signal 115 is up-converted by the frequency up-converter 106, amplified by the amplifying element 107, and sampled by the sampler 108. The fol~ard sampled signal 112 returns to point A 121 via the feedb~ element 111 and the 5 ~ g junction 102, thus completing a transition through the feedb~ck loop. A loop gain determiner 110 is activated during this training sequence to measure the feedback loop's open loop gain by detecting the voltages at points A 121 and B 122 and forming the ratio of the two voltages, i.e. V(A)/V(B). This ratio represents the 1 0 overall feedback loop gain minus the gain contributions of the inactivated variable gain stages 104 and 105, i.e. the open loop gain.
The loop gain de~lllliller 110 includes voltage gain stages, such as operational amplifiers, sample and hold circuitry, analog to digital converters, and a miclol~rocessor.
As briefly mentioned above, the reflected energy representer 114 processes the fol~vard 112 and reflecte~l 113 signal samples to produce represent~tions of the loading presented to the amplifying elem~nt 107. The fol~var~ signal sample 112 co~ lises a scaled 2 0 reproduction of the forward RF carrier energy present at the output of the amplifying element 107. The reflP.cte~l signal sample 113 co~ l;ses a scaled replication of the reflecte~l RF carrier energy received by the ~ntel~ 109. As is known, a ratio formed by dividing the m~itllcle of the re.flto.cted signal sample 113 by the 2 5 magnitude of the fol ~ard signal sample 112 is proportional to the m~gnitllde of a load reflection coefficient. This ratio may be conl~uled within the reflected energy representer 114 by indep~ntle.ntly rectifying and filtering the reflected and forward signal samples 113 and 112 and subsequently determinin~: the ratio of 3 0 the resulting quantities using voltage gain stages, such as operational amplifiers, sample and hold cin;~ y, analog to digital converters, and a microprocessor. This ratio provides a scalar representation of the load mi~m~tch provided by the ~ ellll~ 109 such that a small ratio denotes a minim~l mi~m~tch An ~lte.m~tive method of 3 5 processin,~ the forward and refl~.cterl signal samples 112 and 113 wO 9~i/OlOO9 619 01 I r PCT/US94/06268 includes forming a direct ratio of the reflected and forward signal samples 112 and 113, thus yielding a vector representation of the load reflection coefficient. The vector representation includes m~gnitll~le and phase portions and provides a more detailed electrical 5 description of the load mi~m~tc~ due to the ~ntenn~ 109 than does the scalar or m~gnitllfle representation; however, it also requires more complex circuitry within the reflected energy representer 114 to co,llyLlle an accurate ratio.
Upon obt~inin~ the effects of the varying loading, the reflected energy representer 114 or the loop gain determiner 110 may modify the gain of the variable gain stages, via a DC control voltage or a voltage variable aKenuator, to sul,~ lly m~int~in the overall feedb~ loop gain at a predetç~ine-l gain level based on the 15 linearity specification of the tr~n.~mitter 100. Alternatively, the reflected energy representer 114 or the loop gain detenniner 110 may instruct the signal source 101 to adjust the levels of the I and Q
si n~l~ it produces. This adj~lstm~nt ~h~nges the output power level of the amplifying elem~nt 107 to accommodate the varying lo~lin,~
2 0 while m~int~inin~ the linearity specification of the transmitter 100.
FIG. 2 illustrates a flow diagram of steps that may be executed by the ~ itter to implemPnt the present invelllion. Entering the flow diagram at the START block, logic flow proceeds to block 200 2 5 where the tr~n~mitter determines effects of the varying lo~ding on the overall loop gain. As discllsse-l above, varying lo~-ling at the tr~n~mitter's output is a result of the time depend~nt load impedance presented to the ~ntenn~, i.e. receiving reflected signals that are reflected off of blliklin~, walls, mo-mt~in~, etc. As the loading at 3 0 the output of the ~nt~nn~ varies, the gain of the amplifying elem~ont varies, which changes the overall gain of the tr~n~ el's feedback loop. To determine the overall loop gain variation and its impact on transmitter pelrol,llance, the logic flow may proceed on one of two parallel paths depending upon the distribution of possible load 3 5 impeA~nces and the effect upon power amplifier gain and linearity.
~ WO 95/01009 21 61 gOl ~ . PCT/US94/06268 When the range of load impedances, due to reflected energy received by the antenna, does not diverge appreciably from an impedance for which the amplifying el~mP-nt has been designed (i.e.
5 the overall loop gain change is moderate) the path comm~ncing with block 202 is followed. When large load variations are present (i.e.
large variations in tr~n~mitter gain and linearity) the path initi~tin~
with block 206 is followed. Selection of which method to utilize in a itter is based on the known environment in which the 10 tr~n~mitter will reside. For example, a transmitter operating iri an open area, such as the center of a room or the middle of a field, will have slight load variations, thus the path from block 202 will suffice;
whereas, a transmiller o~e~aling near a highly reflective load, such as a metal wall or building, will have large load variation, thus the 1 5 path from block 206 would be used.
When the former path is selPcte~l, logic flow proceeds to block 202 where the variable gain stages are temporarily deactivated.
Deactivation of the variable gain stages is gellelally accomplished by 2 0 reducing the gain of each stage to approxim~tely zero through a reduction in the DC supply voltage sourcing each stage's active devices. The deactivation is typically ~elrollned at the be~inning of each training sequence, which may occur as often as once a time slot in a TDMA co...~ .ication system (e.g. once every fifteen 2 5 milliseconds), thus allowing the gain of the variable gain stages to be periodically adjusted in response to the varying lo~-lin~.
Upon deactivating the variable gain stages, logic flow continll~s to block 203 where the signal source provides the 3 0 tr~n~mitter with the training signal which is sampled upon exiting the amplifying elçme-nt The training signal, which is applied at the beginnin~ of each training sequence, is used to measure and adjust the fee-lb~ loop phase parameters and to establish the maximllm signal level to which the signal source may drive the power 3 5 amplifier, or amplifying elemçnt. A detailed discussion of the WO 95/OlOOg PCT/US94/06268 2161901 `:
methodology used during the training sequence is provided in the aforementioned United States Patent number 5,066,923, thus no further discussion will be presented except to facilitate the underst~n~lin~ of the present invention. During the training S sequence, the signal source applies the training signal to the output of the I ch~nn~l's variable gain stage in the absence of I and Q data modulation. The training signal may be used either periodically or on an as n~e-le-l basis. For example, in a TDMA system that ~-tili~es a three slot time frame, the training signal may be applied during a 10 portion of each time slot when co....~ ..ications are inactive. After the signal source supplies the training sign~l, it is up-converted to the RF carrier frequency, amplified by the amplifying element, and sampled by a directional coupler, or equivalent sampler. The sampled training signal is down-converted and phase adjusted in the 1 5 feedk~ elemP-nt such that the baseband signal delivered to the I
ch~nnel's sllmming junction is ~lo~elly inverted. The baseband signal proceeds through the snmming junction to the input of the I
ch~nnel's deactivated variable gain stage, thus completing a transition through the complete feedb~ loop.
Upon arrival of the baseband signal to the input of the I
ch~nnçl's deactivated variable gain stage, the logic flow advances to block 204 where the tr~n~mitter detellllilles the open loop gain changes of the feedback loop. The tr~n~ er~s loop gain 2 5 cleterminer measures the signal voltages provided by the training signal and the baseband feedb~ signal and forms the ratio of the two signal voltages to produce the open loop voltage gain. The open loop gain represents the overall feedb~ck loop gain less the gain of the deactivated gain stages. Since the power amplifier is inchlded in 3 0 the feedb~ck loop's forward path, power amplifier gain and linearity changes resulting from tr~n~mhter load variations are expressed as changes to the open loop gain.
Once the open loop gain changes are determin~l and the 3 5 training sequence concl--des, the logic flow proceeds either to block 205, block 212, or both. The logic flow transition to block 205 occurs when loop gain compensation for changes in power amplifier gain is necessary. At block 205, the tr~n~mitter adjusts the gain of the I and Q rh~nnPls' variable gain stages based on the open loop S gain changes. The gain of these stages is adjusted in a known m~nner by varying loop compensation components, such as variable resistors, or via a gain control signal, such as a DC voltage, to m~int~in a predetellllilled overall loop gain. The predeterminP~l loop gain may be a constant value for small deviations in overall l 0 loop gain or a range of values corresponding to a range of overall loop gain changes. The logic flow transition to block 212 occurs when signal source adjustment is nPcess~ry to colll~e.lc~te for changes in power amplifier linearity. At block 212, the transmitter adjusts the signal source levels of the I and Q sign~l~ based on the 15 open loop gain changes. The loop gain delelllliller sends the signal source a s~lin& comm~ntl depen~nt on the extent of the open loop gain changes. The sc~lin~ comm~n~ls in~lic~te the direction and amount of signal source level adjustmP-nt When power amplifier linearity is adversely affected by the tr~n~ ler load, lowering the I
2 0 and Q signal levels assists o~l~tion of the negative feedback loop in avoiding adjacent rh~nnel splatter. For tr~n~ er loads which offer an improvement in power ~mplifiçr linearity, raising the data signal level may be ~lopliate, lller~y taking advantage of a favorable load condition which allows the power output level of the transmil~er 2 5 to be increased. The signal source level and variable gain stage gain adjustmPrlts of blocks 212 and 205 may also be performed ~imlllt~neously, or sequentially, when loop gain colll~ellsation for changes in both power amplifier gain and linearity is necessary.
3 0 When the latter of the two path choices available at block 200 is selected, the logic flow proceeds to block 206 where the transmitter determines the reflected energy. The reflected energy is the RF carrier energy received by the ~ntenn~ when the transmitter is transmitting the RF carrier. Reflected energy results from 3 5 impe~nce mi5m~tches between the ~te ~ impe~l~nce and its load WO 95/01009 . PCT/US94/06268 impe~nce. As briefly discussed with efer~;,lce to block 200, the ~ntenn~ is typically designed such that its input impedance subst~nti~lly m~tch~s the output imped~nce of the power amplifier and its output impe-l~nce m~tches the impedance of free space.
5 Therefore, when the antenna is brought near an object, such as a metal wall, the load presented to it deviates from the desired free space impe~n~e and an amount of energy is reflected into the ~-lte~ proportional to the amount of deviation the ~-telll.a's load imped~nce presents relative to free space. Similarly, as is known, 1 0 the reflected energy received by the ~ntPnn~ enters the output of the power amplifier, thereby effectively varying its load, i.e. t-h-e ~ntenn~'S input impedance. Thus, the load impedance presented to the power amplifier by the ~.~le....~ is a function of objects within proximity of the ~-lte---l~ and is de~ illable by the level of power 15 reflecte-l from the ~ntenn~.
The refl~cte!l energy detel,..i.,~tion may be achieved by selPctinp either one of the two paths available at block 206. When the first path is chosen, the logic flow proceeds to block 208 where 2 0 the tr~n~mitter samples the reflectetl energy and produces a scalar, or m~gnitll(le, represent~tion of the reflected energy. The fol~vard and reflected RF carrier sign~lc present at the ~ntenn~ input are sampled via a directional coupler or other device capable of dirrele-~ *ng the directional nature of these qu~ntities. As 2 5 discussed with l~rerellce to FIG. 1, the forward and reflected signal samples are recti~le~l and filtered to produce DC voltages proportional to their respective m~gnibldes. The ratio of the reflecte~l signal sample m~ niPlde. to the forward signal sample m~gnitllde provides the scalar representation of the reflected energy 3 0 and is known as the m~gnitude of the load reflection coef~lcient.
Thus, the m~xi.~l.... value of this ratio is one, which corresponds to all the fo,-vard energy being reflected.
When the second path at block 206 is selected, the logic flow 3 5 progresses to block 209 where the tr~n~mi~ter samples the reflected ~ WO 95/01009 PCT/US94/06268 '~161901 energy and produces a vector representation of the reflected energy.
The sampling performed in this block is similar to the sampling discussed with reference to block 208, except the forward and reflected signal samples are not simply rectified and filtered. The 5 transmitter operates on the fol~ald and reflected signal samples using known techniques to exlr~olate the m~nitllde and phase q~ntities associated with the ratio of the reflçcte~l signal sample to the fol~vard signal sample. This ratio forms the vector representation of the reflected energy and i~lentifies the m~gnitllde 10 and phase of the load reflection coefficient.
The value of the load reflection coefficient in-lic~tes the degree to which the varying ~ e....~ load deviates from the nominal loading that produces optimal power amplifier ~elro~ nce. Generally, 1 5 ~..t~ loads which are most likely to cause tr~n~mitter instability are those associated with large reflection coefficients, i.e. those which produce load reflection coefficient magnitudes greater than one half. Simil~rly, these highly reflective loads usually degrade the linearity ~e.ro...-~nGe of the power amplifier and are likely to 2 0 require source level adjustment in order to avoid adjacent ch~nnel splatter.
The choice to use either the scalar or vector representation is dete...~ l by the trade off between circuit complexity and 2 5 obtainable level of tr~nsmitter perform~nce under varying ~ntenn~
load conditions. The distribution of possible ~ntenn~ loads covers the complex impedance plane resulting in two-~limP-nsional variables.
Accordingly, the reflection coefficient is a two-dimensional variable, having both m~gnitllde and phase information. The scalar 3 0 represent~tion of the refl~cte-l energy expresses the two--limell~ional load as a single ~lim~n~ional quantity, ignoring load infollllation cont~ine-l by the phase term. Although the circuit complexity to produce such a reflection coefficient is reduced with respect to the vector representation, the scalar representation's 3 5 inability to rii~tin~uish phase may result in sub-optimal adjustment of WO 9S/01009 21619 01 ; ; PCT/US94/06268 ~
the loop gain and signal source levels. Conversely, knowledge of both m~nihlcle and phase of the ~ntenn~ load, as provided by the vector representation, may permit precise setting of loop gain and signal source levels in response to the varying load presented by the 5 ~ntenn~. Providing for such precision typically requires greater circuit complexity.
Upon obt~inin~ either the scalar or vector representation of the reflected energy, the logic flow continlles from blocks 208 and 1 0 209 to blocks 210 and 211 where the transmitter adjusts the gain of the I and Q ch~nnPls' variable gain stages and the signal source levels of the I and Q sign~l~ based on the de~lmilled reflected energy representation. Similar to the ~ cllssion presented with l~fe-c;nce to blocks 205 and 212, the tr~n~mitter's reflected energy representer 1 5 generates a gain control signal to adjust the variable gain stages and a sc~ling comm~n~l to adjust the signal source. Depen~in~ on ~,te,..~
load condition and tr~n~mitter linearity requirements, the tr~nsmitter may adjust both the variable gain stages and the signal source levels or may make only one of the two adjllstmPnt~. Additionally, the 2 0 extent of ~ load variation, as detel.. .ined by the chosen reflected energy representation, may be lltili7e-1 to influence the adj~lstmPnts made to the variable gain stages and the signal source by the loop gain detelllliller in blocks 205 and 212. For large load reflection coefficient m~gnitlldes, the gain setting~ for the variable 2 5 gain stages and the sc-~lin~ comm~nd values ~irnini~tered to the signal source may require reductions relative to their respective settings provided by the loop gain leterminer.
Load reflection coefficient information obtained via the 3 0 reflected energy's vector representation may be applied in digital format to the address lines of a read only memory's (ROM) look-up table, which is cont~ine-l in the tr~ncmitters reflected energy representer. Resident in the ROM is a set of predetermine~l data that may be used to control both the variable gain stages and the signal 3 5 source levels. The data w~illel~ to the ROM is detem~ined by a wo gS/oloog ~1619 01 PCT/US94/06268 pretraining operation at the time of transmitter manufacture. This pretraining procedure operates the tr~nemitter into a series of selected non-optimal loads. Under each load configuration, the loop gain and signal source levels may be adjusted to provide optimal 5 tr~ne.~ er performance within the constraints of stability and m~xi"~ permissible adjacent ch~nnel splatter. In lieu of the ROM
table, the vector representation of the load reflection coefficient may be sllbmitted to a microprocessor or ~igit~l signal processor (DSP) as an input to a m~th~m~tical equation. The microprocessor or DSP
1 0 evaluates the m~th~m~tic~l equation and produces ayy~ iate control qll~ntitiP!s which set loop gain and signal source levels.
The present invention may be further understood by way of anexample. ConsiderTDMAcb.... -l.. ic~tionssystemswhere 1 5 subscriber units, such as mobile or portable radiotelephones, ...it digiti7ed voice or data tlllrin~ pre~esi~ed slot times of TDMA time frames. In this particular e~c~mrle, six ~ eell millieecond tr~nemiesion slots coln~lise a llillely millieecond time frame. A particular subscriber's ~ e~ission will typically be 2 0 ~esigne~l to at least one of the time slots per time frame while the tr~nemi.esion is in progress. The first one millieecond portion of each time slot in each frame is dedicated to tr~nemiesion of the training sequence. The training sequence provides a me~n.e for adjusting the loop phase at the beginning of the tr~nemiesion slot.
2 5 Since the phase around the loop is initi~lly unknown, the loop phase adjustment is made with the feedba~ loop open. In the present invention, this is accomplished by inactivating the baseband variable gain stages present in the feedb~k loop. Known techniques are then employed to alternately m~e~lre and adjust loop phase to produce the 3 0 required feedback signal polarity required for negative feedback.
During the first portion of the training sequence when the baseband variable gain stages are inactivated and the loop phase has been subst~nti~lly adjusted to the value necess~ry for negative 3 5 feedb~k, the transl~ ler measures the signal voltages across the I
WO 95/01009 I'CT/US94/06268 2161901 ''' ch~nnel's variable baseband stage. This measurement represents ~e fee~lb~k open loop gain, i.e. feedback loop gain minus the gain contribution of the variable baseband gain stage, and provides an indication of the change in power amplifier gain associated with the S variable ~ e~ load when an isolator is not utilized in the tr~n~mitter. Variations in power amplifier gain result in variations in fee~lb~ck loop gain and irnpact the amount of linearity m~ntation provided by the overall tr~n~mitter. When the transmitter ~etects changes in the open loop gain, it adjusts the gain 1 0 of the variable baseband gain stages to c~ e,.~te for the gain variation incurred by the power amplifier due to changes in the 'S load.
The second portion of the training sequence is used to 1 5 determine the m~ximnm permissible data signal level which the signal source may supply. When the scalar represçnt~tion of the reflected energy is lltili7e~, this portion of the training sequence is also used as the test signal ~lllrin~ which the m~inlde of the load reflection coefficient is determinP-l Generally, the amount of gain 2 0 reduction required in the variable baseband gain stages is proportional to the magnitude of the load reflection coefficient. For reflection coefficient m~ nit~ldes less than 0.15, the transmitter may make no adjustmPnt to the variable baseband gain stages or the signal source. Alternatively, for reflection coef~lcient m~gnitlldes in the 2 5 0.15 to 0.5 range, the tr~n~mitter may reduce the gain of the variable b~eb~n-l gain stages.
Re-lllctions in the gains of the variable baseband gain stages are typically necess~ry to prevent unstable operation of the feedback 3 0 loop due to mi~m~tch~d loads present at the ~,.le"~ ntenn~ loads which are highly reactive or resonant near the frequency of operation may consume the fee~b~k loop phase margin, resulting in fee~lb~rk loop instability. When a significant reflection coefficient magnitude is m~ lred, the present invention initi~tes a reduction of 3 5 the signal source level in addition to the gain reduction of the WO 95/01009 ~ 1. 619 01 PCT/US94/06268 baseband variable gain stages. Thus, the resulting signal source level is below the ma~in~ - permissible level determine(l during the second portion of the training sequence. The signal source level reduction allows the power amplifier linearity to be m~int~ined, thus 5 inhibiting the occu~ lce of adjacent ch~nnel splatter due to the varying ~..te.~ load.
When the vector represent~tion of the refl~cted energy is incorporated into this example, the ~letel...i.~tion of the open loop 10 gain performed during the first portion of the training sequence may be lmnecess~ry. Rather, adjllctmlont~ to the variable baseband gain stages and the signal source are directly dete~ l from data stored in the tr~n~n~iue~s ROM, or supplied via DSP c~lclll~tion.
1 5 At the conrlll~ion of the one milli~econd training sequence period, the rem~ining fourteen milli.~econds of the slot is used for the tr~n.~mi.csion of user data or digitized voice illfollllation. The training process may be repeated as often as every slot in order to allow the ~ ...itter to track the effects of the time varying ~.~te...
2 0 load on loop gain and linearity.
The present invention provides a method for a transmitter to colllp~n~te for the effects of varying ~ntçnn~ loading without lltili7~tion of an isolator. With this method, the isolator is not 2 5 required between the tr~n~...iller's amplifying elem~nt, or power amplifier, and its ~lt~lllla to insure optimal ~ -nitter perform~nce.
Moreover, the method of the present invention yelllli~s elimin~tion of the isolator, thus enabling the transmitter to occupy less volume, cost less, weigh less, and be less band limited than with the isolator, 3 0 while still m~int~inin~: desired tr~n~mitter functionality.
Linearity of an amplifier is affected by the varying loads presented by the alltenlla of the tr~n~mitter. Typically, an ~ntçnn~ is designed to provide a fixed load, fifty ohms for example. However, due to reflected energy received, the load changes.
To minimi~e variations in lo~lin~, tr~n~mitters generally incorporate isolators to provide a subst~nti~lly con~t~nt load impedance to the amplifier. The isolator incllldes a circulator and a termin~tin~ impedance which is typically fifty ohms. The circulator is a three tçrmin~l device that provides directir.n~l flow of t~he RF
2 0 energy from the amplifier to the ~ and from the ~nte~n~ to the termin~ting impe~3~nce. Therefore, the RF energy sourced by the amplifier is provided to the ~--t~ and any RF energy elltelillg the antenna is absorbed in the tçrmin~tin~ impe~l~nce. Thus, the isolator presents a con~t~nt impedance to the amplifler.
Although the isolator provides a conct~nt load irnpedance to the RF amplifier, size, cost, and bandwidth limit~tions typically inhibit the use of a universal isolator in mobile radios, portable radios, and cellular telephones. For example, a radio that operates at 3 0 132 MHz requires an isolator that has a volume of 8.19 cubic ce..~ eters (0.5 cubic inches), weighs 227 grams (0.5 pounds), and costs $30 each in qll~ntities of 100,000 per year. As a result, an isolator puts obvious constraints on the design of such radios.
Additionally, isolators have fixed bandwidths; therefore, multiple 3 5 isolators may be required in tr~n~mitters which operate over wide wo gS/oloog ~1619 01 PCT/US94/06268 ~.
frequency ranges. This bandwidth limitation is most noticeable at lower RF carrier frequencies, such as VHF, where the allocated frequency band covers a large ~e-cel.t~e bandwidth.
5 To avoid the use of the isolator, existing frequency modulation (FM) transmitters, which inclllde nollline~r amplifiers, utilize protective fee~lb~clc circuitry. The protective feedback monitors the voltage standing wave ratio (VSWR) at the nonline~r amplifier's output and correspondingly reduces the amount of output power 10 provided by the nonlinP~r amplifier to the ~ e~ This approach generally reduces the nonline~r amplifier's output power by a fixed arnount when the VSWR exceeds a predete~ l level. For example, when a 3:1 VSWR is letected at the nonline~r amplifier's output, the nonline~r amplifier's output power may be reduced by 3 dB. This approach works for nonline~r amplifiers, but does not include any provisions for m~ g amplifier linearity.
Therefore, a need exists for a method to co~ te for effects of varying lo~ling without the need for an isolator in a 2 0 tr~n~mitter that has a linear amplifier.
Brief Description of the Drawings 2 5 FIG. 1 illustrates a block diagram depiction of a linear transmitter in accordance with the present invention.
FIG. 2 illustrates a flow diagram of steps executed by a transmitter in accordance widl the present invention.
Description of a Prefe~d Embo~lim~o-nt Generally, the present invention provides a method for a tr~n~mitter to compensate for ~..te~ load variations without using of an isolator (circulator plus resistive termin~tion). This is accomplished by determinin~ the effects that the load variations have 5 on the loop gain of the tr~n~ 's fee-lb~rl~ loop. Upon detel,..i..;l.~; the effects, the tr~n~mitter adjusts either the gain of a variable gain stage within the feedback loop or a signal source level to m~int~in the transmitter's linearity requirement.
The present invention may be more fully described with refel~llce to FIGs. 1 and 2. FIG. 1 illustrates a block diagram of a tr~n~mitter 100 that includes a signal source 101, two variable gain stages 104 and 105, a frequency up-converter 106, an amplifying elçment 107, a sampler 108, an ~ 109, a feedback element 111, l S and a reflected energy representer 114. In a quadrature amplitude modulation (QAM) sch~me, the signal source 101 is typically a digital signal processor (DSP) that generates in-phase (I) and quadrature (Q) signal represent~tions of digital data and processes these I and Q signal represent~tion~ into b~eb~n~ analog waveforms.
2 0 Sllmmin~ junctions 102 and 103 receive the baseband analog I and Q
waveforms, respectively, and subtract them from the baseband outputs of the feedback elem~ont 111. The resultant waveforms proceed to their respective gain stages 104 and lOS for amplification and continue to the frequency up-converter 106 for tr~n~l~tion to a 2 5 selected radio frequency (RP) carrier. The amplifying elçment 107 amplifies the RF carrier and sllbmit~ the amplified RF carrier to the sampler 108 and the ~ e....~ 109 for radio tr~n~mi~sion.
The sampler 108 samples the amplified RF signal provided by 3 0 ~e amplifying element 107 and supplies forward and reflected signal samples 112 and 113 to the reflected energy representer 114 and supplies forward signal samples 112 to the feedback elçmtont 111.
The feedback element 111, which may incl~ e a frequency down-converter, gain stages, and loop phase adjustment circuits, receives 3 5 the forward signal sample 112, frequency down-converts it by reversing the processing order of the frequency up-converter 106, and produces baseband analog I and Q data representations. These baseband I and Q sigT~ provide the inverting inputs to the sllmming junctions 102 and 103, thus completing a feedback path. Providing 5 closed loop feedback in such tr~n.~mitters is known, thus no further discussion will be presente(l except to facilitate the underst~n~lin~ of this invention. The refl~cte~ energy representer 114 accepts the fol~vald and reflected signal samples 112 and 113 to generate represent~tions of the reflected energy based on its input signal 10 samples. As let~ile(l below, the represent~tions of the reflected energy contribute to dete....;..;.~ the gains of the two variable gain stages 104 and 105 and the signal levels of the I and Q signal representations generated by the signal source 101.
The functional blocks of the tr~ncmitter 100 incl~e a variety of circuit topologies. The gain stages 104 and 105 are typically variable high gain, low frequency amplifiers whose operating bandwidth is restricted to the loop bandwidth required for proper operation of the negative fee~b~ loop, thereby encomp~sing the 2 0 bandwidth of the data mo~ tion sigr~l~ supplied by the signal source 101. The frequency up-converter 106 inc]ll~les mixers 116 and 120, a local oscillator 117, a 90 degree phase shifter 118, and a signal combiner 119. The frequency tr~ncl~tion from baseband to the RF carrier frequency is achieved in the frequency up-converter 2 5 106 by mixing the baseband I signal waveform with the local oscillator frequency, mixin~ the b~ceb~n~l Q signal waveform with a 90 degree phase shifted version of the local oscillator frequency, and ~IIIIIIIIil~g the two mixer outputs to produce indepçn~lent I and Q data signals at the common RF carrier frequency. The amplifying 3 0 element 107 is an RF power amplifier that provides power amplification of the RF carrier frequency, which cont~in~ the up-converted data si n~l~, and is commonly formed by cascading several amplifier stages or transistors. Since the envelopes of the analog I
and Q waveforms are amplitude depen~lent, the amplifying el~ment 3 5 107 is a linear, class A or class AB, amplifier as opposed to a constant envelope, or class C, nonline~r amplifier. The sampler 108 is preferably a directional coupler that provides scaled representations of forward and reflecte-l voltage, or current, waveforms present at the ~lte,l~la 109.
The sampler 108, feedback loop elemP,nt 111, and sllmming junctions 102 and 103 are configured to e~h~nce the linear amplifying properties of the amplifying elemto,nt 107. As briefly mentioned above, these el~,me,nt~ form a negative feedback system 10 which reduces power levels of ullwa~lted sign~l~ which fall outside the ~si~ed frequency band. These ullwallled signals result from nonlinç~rities in the amplifying el~,m~,nt 107 and may cause intelrelcllce in adjacent RF ch~nnels, thus dislu~lillg commlmic~tions on those CllA~ . The ullw~ ed si n~ in the adjacent RF r.h~nn~.ls 1 5 are known as splatter. For a time division mllhirle access (TDMA) system, the adjacent ch~nnel splatter specification may be in the -60 dBc range when comparing the signal powers of the signal in an cçnt ~1IAII~ to the RF carrier sign~l This specification is typically achieved by lltili7.in~ an amplifying elem~nt 107 whose 2 0 linearity provides adjacent rh~nnP,l splatter performance of -30 dBc and improving the adjacent ch~nnel splatter by 30 dB using known Cartesian fee~b~ck correction technillues.
Effects of varying lo~ling at the output of the amplifying 2 5 eleme.nt 107 due to changes in All~ellll~ 109 impe-l~nce are dete~nine~l by ei~er measuring ~e gain changes of the amplifying element 107, detenninin~ represe.~t~tions of the lo~tlin~ via reflected energy received by the antenna 109, or both. To measure the gain variation of the amplifying elP-m~.nt 107, a training signal 115, such 3 0 as a signal discussed in United States Patent number 5,066,923, entitled "T,ine~r Tr~n.cmitter Training Method And Apparatus" and ~signed to Motorola, is input from the signal source 101 to point B
122 of FIG. 1 when the variable gain stages 104 and 105 are inactivated. Inactivating the variable gain stages 104 and 105 3 5 effectively opens the fol~ rd path of a fee~lb~c.k loop within the wo gs/oloog ~ fil ~ PCT/US94106268 tr~n.cmitter 100. The training signal 115 is up-converted by the frequency up-converter 106, amplified by the amplifying element 107, and sampled by the sampler 108. The fol~ard sampled signal 112 returns to point A 121 via the feedb~ element 111 and the 5 ~ g junction 102, thus completing a transition through the feedb~ck loop. A loop gain determiner 110 is activated during this training sequence to measure the feedback loop's open loop gain by detecting the voltages at points A 121 and B 122 and forming the ratio of the two voltages, i.e. V(A)/V(B). This ratio represents the 1 0 overall feedback loop gain minus the gain contributions of the inactivated variable gain stages 104 and 105, i.e. the open loop gain.
The loop gain de~lllliller 110 includes voltage gain stages, such as operational amplifiers, sample and hold circuitry, analog to digital converters, and a miclol~rocessor.
As briefly mentioned above, the reflected energy representer 114 processes the fol~vard 112 and reflecte~l 113 signal samples to produce represent~tions of the loading presented to the amplifying elem~nt 107. The fol~var~ signal sample 112 co~ lises a scaled 2 0 reproduction of the forward RF carrier energy present at the output of the amplifying element 107. The reflP.cte~l signal sample 113 co~ l;ses a scaled replication of the reflecte~l RF carrier energy received by the ~ntel~ 109. As is known, a ratio formed by dividing the m~itllcle of the re.flto.cted signal sample 113 by the 2 5 magnitude of the fol ~ard signal sample 112 is proportional to the m~gnitllde of a load reflection coefficient. This ratio may be conl~uled within the reflected energy representer 114 by indep~ntle.ntly rectifying and filtering the reflected and forward signal samples 113 and 112 and subsequently determinin~: the ratio of 3 0 the resulting quantities using voltage gain stages, such as operational amplifiers, sample and hold cin;~ y, analog to digital converters, and a microprocessor. This ratio provides a scalar representation of the load mi~m~tch provided by the ~ ellll~ 109 such that a small ratio denotes a minim~l mi~m~tch An ~lte.m~tive method of 3 5 processin,~ the forward and refl~.cterl signal samples 112 and 113 wO 9~i/OlOO9 619 01 I r PCT/US94/06268 includes forming a direct ratio of the reflected and forward signal samples 112 and 113, thus yielding a vector representation of the load reflection coefficient. The vector representation includes m~gnitll~le and phase portions and provides a more detailed electrical 5 description of the load mi~m~tc~ due to the ~ntenn~ 109 than does the scalar or m~gnitllfle representation; however, it also requires more complex circuitry within the reflected energy representer 114 to co,llyLlle an accurate ratio.
Upon obt~inin~ the effects of the varying loading, the reflected energy representer 114 or the loop gain determiner 110 may modify the gain of the variable gain stages, via a DC control voltage or a voltage variable aKenuator, to sul,~ lly m~int~in the overall feedb~ loop gain at a predetç~ine-l gain level based on the 15 linearity specification of the tr~n.~mitter 100. Alternatively, the reflected energy representer 114 or the loop gain detenniner 110 may instruct the signal source 101 to adjust the levels of the I and Q
si n~l~ it produces. This adj~lstm~nt ~h~nges the output power level of the amplifying elem~nt 107 to accommodate the varying lo~lin,~
2 0 while m~int~inin~ the linearity specification of the transmitter 100.
FIG. 2 illustrates a flow diagram of steps that may be executed by the ~ itter to implemPnt the present invelllion. Entering the flow diagram at the START block, logic flow proceeds to block 200 2 5 where the tr~n~mitter determines effects of the varying lo~ding on the overall loop gain. As discllsse-l above, varying lo~-ling at the tr~n~mitter's output is a result of the time depend~nt load impedance presented to the ~ntenn~, i.e. receiving reflected signals that are reflected off of blliklin~, walls, mo-mt~in~, etc. As the loading at 3 0 the output of the ~nt~nn~ varies, the gain of the amplifying elem~ont varies, which changes the overall gain of the tr~n~ el's feedback loop. To determine the overall loop gain variation and its impact on transmitter pelrol,llance, the logic flow may proceed on one of two parallel paths depending upon the distribution of possible load 3 5 impeA~nces and the effect upon power amplifier gain and linearity.
~ WO 95/01009 21 61 gOl ~ . PCT/US94/06268 When the range of load impedances, due to reflected energy received by the antenna, does not diverge appreciably from an impedance for which the amplifying el~mP-nt has been designed (i.e.
5 the overall loop gain change is moderate) the path comm~ncing with block 202 is followed. When large load variations are present (i.e.
large variations in tr~n~mitter gain and linearity) the path initi~tin~
with block 206 is followed. Selection of which method to utilize in a itter is based on the known environment in which the 10 tr~n~mitter will reside. For example, a transmitter operating iri an open area, such as the center of a room or the middle of a field, will have slight load variations, thus the path from block 202 will suffice;
whereas, a transmiller o~e~aling near a highly reflective load, such as a metal wall or building, will have large load variation, thus the 1 5 path from block 206 would be used.
When the former path is selPcte~l, logic flow proceeds to block 202 where the variable gain stages are temporarily deactivated.
Deactivation of the variable gain stages is gellelally accomplished by 2 0 reducing the gain of each stage to approxim~tely zero through a reduction in the DC supply voltage sourcing each stage's active devices. The deactivation is typically ~elrollned at the be~inning of each training sequence, which may occur as often as once a time slot in a TDMA co...~ .ication system (e.g. once every fifteen 2 5 milliseconds), thus allowing the gain of the variable gain stages to be periodically adjusted in response to the varying lo~-lin~.
Upon deactivating the variable gain stages, logic flow continll~s to block 203 where the signal source provides the 3 0 tr~n~mitter with the training signal which is sampled upon exiting the amplifying elçme-nt The training signal, which is applied at the beginnin~ of each training sequence, is used to measure and adjust the fee-lb~ loop phase parameters and to establish the maximllm signal level to which the signal source may drive the power 3 5 amplifier, or amplifying elemçnt. A detailed discussion of the WO 95/OlOOg PCT/US94/06268 2161901 `:
methodology used during the training sequence is provided in the aforementioned United States Patent number 5,066,923, thus no further discussion will be presented except to facilitate the underst~n~lin~ of the present invention. During the training S sequence, the signal source applies the training signal to the output of the I ch~nn~l's variable gain stage in the absence of I and Q data modulation. The training signal may be used either periodically or on an as n~e-le-l basis. For example, in a TDMA system that ~-tili~es a three slot time frame, the training signal may be applied during a 10 portion of each time slot when co....~ ..ications are inactive. After the signal source supplies the training sign~l, it is up-converted to the RF carrier frequency, amplified by the amplifying element, and sampled by a directional coupler, or equivalent sampler. The sampled training signal is down-converted and phase adjusted in the 1 5 feedk~ elemP-nt such that the baseband signal delivered to the I
ch~nnel's sllmming junction is ~lo~elly inverted. The baseband signal proceeds through the snmming junction to the input of the I
ch~nnel's deactivated variable gain stage, thus completing a transition through the complete feedb~ loop.
Upon arrival of the baseband signal to the input of the I
ch~nnçl's deactivated variable gain stage, the logic flow advances to block 204 where the tr~n~mitter detellllilles the open loop gain changes of the feedback loop. The tr~n~ er~s loop gain 2 5 cleterminer measures the signal voltages provided by the training signal and the baseband feedb~ signal and forms the ratio of the two signal voltages to produce the open loop voltage gain. The open loop gain represents the overall feedb~ck loop gain less the gain of the deactivated gain stages. Since the power amplifier is inchlded in 3 0 the feedb~ck loop's forward path, power amplifier gain and linearity changes resulting from tr~n~mhter load variations are expressed as changes to the open loop gain.
Once the open loop gain changes are determin~l and the 3 5 training sequence concl--des, the logic flow proceeds either to block 205, block 212, or both. The logic flow transition to block 205 occurs when loop gain compensation for changes in power amplifier gain is necessary. At block 205, the tr~n~mitter adjusts the gain of the I and Q rh~nnPls' variable gain stages based on the open loop S gain changes. The gain of these stages is adjusted in a known m~nner by varying loop compensation components, such as variable resistors, or via a gain control signal, such as a DC voltage, to m~int~in a predetellllilled overall loop gain. The predeterminP~l loop gain may be a constant value for small deviations in overall l 0 loop gain or a range of values corresponding to a range of overall loop gain changes. The logic flow transition to block 212 occurs when signal source adjustment is nPcess~ry to colll~e.lc~te for changes in power amplifier linearity. At block 212, the transmitter adjusts the signal source levels of the I and Q sign~l~ based on the 15 open loop gain changes. The loop gain delelllliller sends the signal source a s~lin& comm~ntl depen~nt on the extent of the open loop gain changes. The sc~lin~ comm~n~ls in~lic~te the direction and amount of signal source level adjustmP-nt When power amplifier linearity is adversely affected by the tr~n~ ler load, lowering the I
2 0 and Q signal levels assists o~l~tion of the negative feedback loop in avoiding adjacent rh~nnel splatter. For tr~n~ er loads which offer an improvement in power ~mplifiçr linearity, raising the data signal level may be ~lopliate, lller~y taking advantage of a favorable load condition which allows the power output level of the transmil~er 2 5 to be increased. The signal source level and variable gain stage gain adjustmPrlts of blocks 212 and 205 may also be performed ~imlllt~neously, or sequentially, when loop gain colll~ellsation for changes in both power amplifier gain and linearity is necessary.
3 0 When the latter of the two path choices available at block 200 is selected, the logic flow proceeds to block 206 where the transmitter determines the reflected energy. The reflected energy is the RF carrier energy received by the ~ntenn~ when the transmitter is transmitting the RF carrier. Reflected energy results from 3 5 impe~nce mi5m~tches between the ~te ~ impe~l~nce and its load WO 95/01009 . PCT/US94/06268 impe~nce. As briefly discussed with efer~;,lce to block 200, the ~ntenn~ is typically designed such that its input impedance subst~nti~lly m~tch~s the output imped~nce of the power amplifier and its output impe-l~nce m~tches the impedance of free space.
5 Therefore, when the antenna is brought near an object, such as a metal wall, the load presented to it deviates from the desired free space impe~n~e and an amount of energy is reflected into the ~-lte~ proportional to the amount of deviation the ~-telll.a's load imped~nce presents relative to free space. Similarly, as is known, 1 0 the reflected energy received by the ~ntPnn~ enters the output of the power amplifier, thereby effectively varying its load, i.e. t-h-e ~ntenn~'S input impedance. Thus, the load impedance presented to the power amplifier by the ~.~le....~ is a function of objects within proximity of the ~-lte---l~ and is de~ illable by the level of power 15 reflecte-l from the ~ntenn~.
The refl~cte!l energy detel,..i.,~tion may be achieved by selPctinp either one of the two paths available at block 206. When the first path is chosen, the logic flow proceeds to block 208 where 2 0 the tr~n~mitter samples the reflectetl energy and produces a scalar, or m~gnitll(le, represent~tion of the reflected energy. The fol~vard and reflected RF carrier sign~lc present at the ~ntenn~ input are sampled via a directional coupler or other device capable of dirrele-~ *ng the directional nature of these qu~ntities. As 2 5 discussed with l~rerellce to FIG. 1, the forward and reflected signal samples are recti~le~l and filtered to produce DC voltages proportional to their respective m~gnibldes. The ratio of the reflecte~l signal sample m~ niPlde. to the forward signal sample m~gnitllde provides the scalar representation of the reflected energy 3 0 and is known as the m~gnitude of the load reflection coef~lcient.
Thus, the m~xi.~l.... value of this ratio is one, which corresponds to all the fo,-vard energy being reflected.
When the second path at block 206 is selected, the logic flow 3 5 progresses to block 209 where the tr~n~mi~ter samples the reflected ~ WO 95/01009 PCT/US94/06268 '~161901 energy and produces a vector representation of the reflected energy.
The sampling performed in this block is similar to the sampling discussed with reference to block 208, except the forward and reflected signal samples are not simply rectified and filtered. The 5 transmitter operates on the fol~ald and reflected signal samples using known techniques to exlr~olate the m~nitllde and phase q~ntities associated with the ratio of the reflçcte~l signal sample to the fol~vard signal sample. This ratio forms the vector representation of the reflected energy and i~lentifies the m~gnitllde 10 and phase of the load reflection coefficient.
The value of the load reflection coefficient in-lic~tes the degree to which the varying ~ e....~ load deviates from the nominal loading that produces optimal power amplifier ~elro~ nce. Generally, 1 5 ~..t~ loads which are most likely to cause tr~n~mitter instability are those associated with large reflection coefficients, i.e. those which produce load reflection coefficient magnitudes greater than one half. Simil~rly, these highly reflective loads usually degrade the linearity ~e.ro...-~nGe of the power amplifier and are likely to 2 0 require source level adjustment in order to avoid adjacent ch~nnel splatter.
The choice to use either the scalar or vector representation is dete...~ l by the trade off between circuit complexity and 2 5 obtainable level of tr~nsmitter perform~nce under varying ~ntenn~
load conditions. The distribution of possible ~ntenn~ loads covers the complex impedance plane resulting in two-~limP-nsional variables.
Accordingly, the reflection coefficient is a two-dimensional variable, having both m~gnitllde and phase information. The scalar 3 0 represent~tion of the refl~cte-l energy expresses the two--limell~ional load as a single ~lim~n~ional quantity, ignoring load infollllation cont~ine-l by the phase term. Although the circuit complexity to produce such a reflection coefficient is reduced with respect to the vector representation, the scalar representation's 3 5 inability to rii~tin~uish phase may result in sub-optimal adjustment of WO 9S/01009 21619 01 ; ; PCT/US94/06268 ~
the loop gain and signal source levels. Conversely, knowledge of both m~nihlcle and phase of the ~ntenn~ load, as provided by the vector representation, may permit precise setting of loop gain and signal source levels in response to the varying load presented by the 5 ~ntenn~. Providing for such precision typically requires greater circuit complexity.
Upon obt~inin~ either the scalar or vector representation of the reflected energy, the logic flow continlles from blocks 208 and 1 0 209 to blocks 210 and 211 where the transmitter adjusts the gain of the I and Q ch~nnPls' variable gain stages and the signal source levels of the I and Q sign~l~ based on the de~lmilled reflected energy representation. Similar to the ~ cllssion presented with l~fe-c;nce to blocks 205 and 212, the tr~n~mitter's reflected energy representer 1 5 generates a gain control signal to adjust the variable gain stages and a sc~ling comm~n~l to adjust the signal source. Depen~in~ on ~,te,..~
load condition and tr~n~mitter linearity requirements, the tr~nsmitter may adjust both the variable gain stages and the signal source levels or may make only one of the two adjllstmPnt~. Additionally, the 2 0 extent of ~ load variation, as detel.. .ined by the chosen reflected energy representation, may be lltili7e-1 to influence the adj~lstmPnts made to the variable gain stages and the signal source by the loop gain detelllliller in blocks 205 and 212. For large load reflection coefficient m~gnitlldes, the gain setting~ for the variable 2 5 gain stages and the sc-~lin~ comm~nd values ~irnini~tered to the signal source may require reductions relative to their respective settings provided by the loop gain leterminer.
Load reflection coefficient information obtained via the 3 0 reflected energy's vector representation may be applied in digital format to the address lines of a read only memory's (ROM) look-up table, which is cont~ine-l in the tr~ncmitters reflected energy representer. Resident in the ROM is a set of predetermine~l data that may be used to control both the variable gain stages and the signal 3 5 source levels. The data w~illel~ to the ROM is detem~ined by a wo gS/oloog ~1619 01 PCT/US94/06268 pretraining operation at the time of transmitter manufacture. This pretraining procedure operates the tr~nemitter into a series of selected non-optimal loads. Under each load configuration, the loop gain and signal source levels may be adjusted to provide optimal 5 tr~ne.~ er performance within the constraints of stability and m~xi"~ permissible adjacent ch~nnel splatter. In lieu of the ROM
table, the vector representation of the load reflection coefficient may be sllbmitted to a microprocessor or ~igit~l signal processor (DSP) as an input to a m~th~m~tical equation. The microprocessor or DSP
1 0 evaluates the m~th~m~tic~l equation and produces ayy~ iate control qll~ntitiP!s which set loop gain and signal source levels.
The present invention may be further understood by way of anexample. ConsiderTDMAcb.... -l.. ic~tionssystemswhere 1 5 subscriber units, such as mobile or portable radiotelephones, ...it digiti7ed voice or data tlllrin~ pre~esi~ed slot times of TDMA time frames. In this particular e~c~mrle, six ~ eell millieecond tr~nemiesion slots coln~lise a llillely millieecond time frame. A particular subscriber's ~ e~ission will typically be 2 0 ~esigne~l to at least one of the time slots per time frame while the tr~nemi.esion is in progress. The first one millieecond portion of each time slot in each frame is dedicated to tr~nemiesion of the training sequence. The training sequence provides a me~n.e for adjusting the loop phase at the beginning of the tr~nemiesion slot.
2 5 Since the phase around the loop is initi~lly unknown, the loop phase adjustment is made with the feedba~ loop open. In the present invention, this is accomplished by inactivating the baseband variable gain stages present in the feedb~k loop. Known techniques are then employed to alternately m~e~lre and adjust loop phase to produce the 3 0 required feedback signal polarity required for negative feedback.
During the first portion of the training sequence when the baseband variable gain stages are inactivated and the loop phase has been subst~nti~lly adjusted to the value necess~ry for negative 3 5 feedb~k, the transl~ ler measures the signal voltages across the I
WO 95/01009 I'CT/US94/06268 2161901 ''' ch~nnel's variable baseband stage. This measurement represents ~e fee~lb~k open loop gain, i.e. feedback loop gain minus the gain contribution of the variable baseband gain stage, and provides an indication of the change in power amplifier gain associated with the S variable ~ e~ load when an isolator is not utilized in the tr~n~mitter. Variations in power amplifier gain result in variations in fee~lb~ck loop gain and irnpact the amount of linearity m~ntation provided by the overall tr~n~mitter. When the transmitter ~etects changes in the open loop gain, it adjusts the gain 1 0 of the variable baseband gain stages to c~ e,.~te for the gain variation incurred by the power amplifier due to changes in the 'S load.
The second portion of the training sequence is used to 1 5 determine the m~ximnm permissible data signal level which the signal source may supply. When the scalar represçnt~tion of the reflected energy is lltili7e~, this portion of the training sequence is also used as the test signal ~lllrin~ which the m~inlde of the load reflection coefficient is determinP-l Generally, the amount of gain 2 0 reduction required in the variable baseband gain stages is proportional to the magnitude of the load reflection coefficient. For reflection coefficient m~ nit~ldes less than 0.15, the transmitter may make no adjustmPnt to the variable baseband gain stages or the signal source. Alternatively, for reflection coef~lcient m~gnitlldes in the 2 5 0.15 to 0.5 range, the tr~n~mitter may reduce the gain of the variable b~eb~n-l gain stages.
Re-lllctions in the gains of the variable baseband gain stages are typically necess~ry to prevent unstable operation of the feedback 3 0 loop due to mi~m~tch~d loads present at the ~,.le"~ ntenn~ loads which are highly reactive or resonant near the frequency of operation may consume the fee~b~k loop phase margin, resulting in fee~lb~rk loop instability. When a significant reflection coefficient magnitude is m~ lred, the present invention initi~tes a reduction of 3 5 the signal source level in addition to the gain reduction of the WO 95/01009 ~ 1. 619 01 PCT/US94/06268 baseband variable gain stages. Thus, the resulting signal source level is below the ma~in~ - permissible level determine(l during the second portion of the training sequence. The signal source level reduction allows the power amplifier linearity to be m~int~ined, thus 5 inhibiting the occu~ lce of adjacent ch~nnel splatter due to the varying ~..te.~ load.
When the vector represent~tion of the refl~cted energy is incorporated into this example, the ~letel...i.~tion of the open loop 10 gain performed during the first portion of the training sequence may be lmnecess~ry. Rather, adjllctmlont~ to the variable baseband gain stages and the signal source are directly dete~ l from data stored in the tr~n~n~iue~s ROM, or supplied via DSP c~lclll~tion.
1 5 At the conrlll~ion of the one milli~econd training sequence period, the rem~ining fourteen milli.~econds of the slot is used for the tr~n.~mi.csion of user data or digitized voice illfollllation. The training process may be repeated as often as every slot in order to allow the ~ ...itter to track the effects of the time varying ~.~te...
2 0 load on loop gain and linearity.
The present invention provides a method for a transmitter to colllp~n~te for the effects of varying ~ntçnn~ loading without lltili7~tion of an isolator. With this method, the isolator is not 2 5 required between the tr~n~...iller's amplifying elem~nt, or power amplifier, and its ~lt~lllla to insure optimal ~ -nitter perform~nce.
Moreover, the method of the present invention yelllli~s elimin~tion of the isolator, thus enabling the transmitter to occupy less volume, cost less, weigh less, and be less band limited than with the isolator, 3 0 while still m~int~inin~: desired tr~n~mitter functionality.
Claims (8)
1. A method for a transmitter to compensate for varying loading without utilization of an isolator, wherein the transmitter includes closed loop feedback, at least one linear amplifying element, an antenna, and at least one gain stage, wherein the closed loop feedback substantially maintains linear operation of the at least one amplifying element, the method comprises the steps of:
a) determining, by the transmitter, effects of the varying loading on overall loop gain of the transmitter to determine overall loop gain changes; and b) adjusting, by the transmitter, gain of the at least one gain stage based on the overall loop gain changes to substantially maintain the overall loop gain at a predetermined gain level.
a) determining, by the transmitter, effects of the varying loading on overall loop gain of the transmitter to determine overall loop gain changes; and b) adjusting, by the transmitter, gain of the at least one gain stage based on the overall loop gain changes to substantially maintain the overall loop gain at a predetermined gain level.
2. In the method of claim 1, the determination of the overall loop gain changes of step (a) comprises:
a1) temporarily deactivating the at least one gain stage; and a2) determining the effects of the varying loading on the overall loop gain less the gain of the at least one gain stage to produce open loop gain changes.
a1) temporarily deactivating the at least one gain stage; and a2) determining the effects of the varying loading on the overall loop gain less the gain of the at least one gain stage to produce open loop gain changes.
3. In the method of claim 2, step (b) further comprises adjusting the gain of the at least one gain stage based on the open loop gain changes to substantially maintain the overall loop gain at the predetermined gain level.
4. In the method of claim 1, the determination of the effects of the varying loading further comprises determining reflected energy received by the antenna.
5. In the method of claim 4, step (b) further comprises determining a representation of the reflected energy and adjusting the gain of the at least one gain stage based on the representation of the reflected energy and the overall loop gain changes to substantially maintain the overall loop gain at the predetermined gain level, wherein the representation of the reflected energy includes a magnitude portion.
6. In the method of claim 4, step (b) further comprises determining a vector representation of the reflected energy and adjusting the gain of the at least one gain stage based on the vector representation of the reflected energy to substantially maintain the overall loop gain at the predetermined gain level, wherein the vector representation of the reflected energy includes a magnitude portion and a phase portion.
7. A method for a transmitter to compensate for varying loading without utilization of an isolator, wherein the transmitter includes closed loop feedback, at least one linear amplifying element, an antenna, a signal source, and at least one gain stage, wherein the closed loop feedback substantially maintains linear operation of the at least one amplifying element, wherein the signal source provides signals to the at least one linear amplifying element for amplification, and wherein the varying loading substantially occurs as a result of reflected energy being received by the antenna, the method comprises the steps of;
a) determining, by the transmitter, a representation of the reflected energy b) adjusting, by the transmitter, signal source level of the signals provided by the signal source to the at least one amplifying element based on the representation of the reflected energy;
the method further comprises the steps of:
c) determining, by the transmitter, effects of the varying loading on overall loop gain of the transmitter to determine overall loop gain changes;
d) adjusting, by the transmitter, the gain of the at least one gain stage based on the representation of the of the reflected energy and the overall loop gain changes to substantially maintain the overall loop gain at a predetermined gain level.
a) determining, by the transmitter, a representation of the reflected energy b) adjusting, by the transmitter, signal source level of the signals provided by the signal source to the at least one amplifying element based on the representation of the reflected energy;
the method further comprises the steps of:
c) determining, by the transmitter, effects of the varying loading on overall loop gain of the transmitter to determine overall loop gain changes;
d) adjusting, by the transmitter, the gain of the at least one gain stage based on the representation of the of the reflected energy and the overall loop gain changes to substantially maintain the overall loop gain at a predetermined gain level.
8. In the method of claim 7, step (d) further comprises adjusting the gain of the at least one gain stage based on a vector representation of the reflected energy to substantially maintain the overall loop gain at the predetermined gain level, wherein the vector representation of the reflected energy includes a magnitude portion and a phase portion.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002323915A CA2323915A1 (en) | 1993-06-24 | 1994-06-03 | A method for a transmitter to compensate for varying loading without an isolator |
Applications Claiming Priority (3)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| US08/080,635 US5423082A (en) | 1993-06-24 | 1993-06-24 | Method for a transmitter to compensate for varying loading without an isolator |
| US08/080,635 | 1993-06-24 | ||
| PCT/US1994/006268 WO1995001009A1 (en) | 1993-06-24 | 1994-06-03 | A method for a transmitter to compensate for varying loading without an isolator |
Related Child Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002323915A Division CA2323915A1 (en) | 1993-06-24 | 1994-06-03 | A method for a transmitter to compensate for varying loading without an isolator |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| CA2161901A1 CA2161901A1 (en) | 1995-01-05 |
| CA2161901C true CA2161901C (en) | 2001-02-20 |
Family
ID=22158623
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002161901A Expired - Lifetime CA2161901C (en) | 1993-06-24 | 1994-06-03 | A method for a transmitter to compensate for varying loading without an isolator |
Country Status (7)
| Country | Link |
|---|---|
| US (2) | US5423082A (en) |
| EP (2) | EP1239597B1 (en) |
| CN (1) | CN1046385C (en) |
| AT (2) | ATE225991T1 (en) |
| CA (1) | CA2161901C (en) |
| DE (2) | DE69434350T2 (en) |
| WO (1) | WO1995001009A1 (en) |
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- 1994-06-03 DE DE69434350T patent/DE69434350T2/en not_active Expired - Lifetime
- 1994-06-03 CN CN94192522A patent/CN1046385C/en not_active Expired - Lifetime
- 1994-06-03 EP EP02008491A patent/EP1239597B1/en not_active Expired - Lifetime
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- 1995-01-27 US US08/379,417 patent/US5542096A/en not_active Expired - Lifetime
Also Published As
| Publication number | Publication date |
|---|---|
| EP1239597A1 (en) | 2002-09-11 |
| DE69434350D1 (en) | 2005-05-25 |
| EP0710411A1 (en) | 1996-05-08 |
| CA2161901A1 (en) | 1995-01-05 |
| WO1995001009A1 (en) | 1995-01-05 |
| DE69431522T2 (en) | 2003-02-20 |
| DE69431522D1 (en) | 2002-11-14 |
| CN1046385C (en) | 1999-11-10 |
| DE69434350T2 (en) | 2005-09-22 |
| US5423082A (en) | 1995-06-06 |
| EP1239597B1 (en) | 2005-04-20 |
| ATE225991T1 (en) | 2002-10-15 |
| ATE293852T1 (en) | 2005-05-15 |
| CN1125497A (en) | 1996-06-26 |
| US5542096A (en) | 1996-07-30 |
| EP0710411B1 (en) | 2002-10-09 |
| EP0710411A4 (en) | 2000-04-12 |
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Legal Events
| Date | Code | Title | Description |
|---|---|---|---|
| EEER | Examination request | ||
| MKEX | Expiry |
Effective date: 20140603 |