DE4128962A1 - Solar or wind generator combination with rechargeable battery - has DC generator decoupled by diode(s) and protected against reverse current - Google Patents

Abstract

The d.c. generator, e.g. solar cells, are decoupled by at least one diode (D7). The current from a wind driven alternator, or three-phase generator is connected to a nodal point via a rectifier (D1-D6). The system uses a parallel control with a variable shunt of a transistor (T1), resistor (R1), capacitor (C1) and a diode (D8). The transistor and the resistor are in series, but in parallel to the nodal point of the two generator types. They are energised by an electronic circuit in a pulse width modulated or analogue manner. USE/ADVANTAGE - For combined battery charger etc., with increased precision and reduced noise. Includes overcharge and over discharge protection and r.p.m. display for rotation of wind generator blades.

Description

Fig. 1 overview circuit diagram of the power electronics, interconnection of (wind) generator, solar panel, battery and consumer with the auxiliary circuits indicated as "black boxes": overcharge protection, deep discharge protection and speed display.

Fig. 2 overload protection according to the prior art.

Fig. 3 overcharge protection electronics part of the invention with voltage and current limitation in pulse width modulated design.

Fig. 4 diagram: temperature dependence of the final charge voltage of a lead accumulator.

Fig. 5 block diagram for the introduction of sensing lines Lei.

Fig. 6 Four circuit options for generating a temperature-compensated reference voltage with a negative temperature coefficient.

Fig. 7 embodiment of electronics for a temperature-compensated voltage limitation with special power supply.

Fig. 8 deep discharge protection in the prior art.

Fig. 9 deep discharge protection, invention with transistor and electronic fuse.

Fig. 10 deep discharge protection, invention with bistable relay.

Fig. 11 overall circuit diagram of an embodiment with current and voltage limitation, deep discharge protection with bistable relay, rectification and overvoltage protection in the high-current section, measuring resistors and display instruments.

Fig. 12 invention of a speed display that converts the generator frequency into a speed-proportional voltage.

Fig. 13 control circuit for separately excited (wind) generator with the aid of the circuit of Fig. 12.

Fig. 1 overview circuit diagram

The main circuits for the generation, limitation, charging and consumption of the high-voltage current are only shown schematically in the overview circuit diagram. The control, regulation and display electronics for the charge limitation (electronics 1 ), the deep discharge protection (electronics 2 ), the speed display (electronics 3 ) and the rectification of the generator current are drawn as "black boxes" and are dealt with in more detail below. As in circuits according to the prior art, T1 and D8 are present as a parallel regulator actuator or as a reflux blocking diode. However, D8 could even be omitted in this circuit if the charge limit is set up so safely that T1 does not empty the battery. This is because a further reflux blocking diode D7 in series with the solar cells feeding on the node of rectifiers D1-D6, C1, R1 (T1), D8 prevents the wind generator current from flowing through the solar cells and destroying them. Furthermore, according to the invention, R1 and C1 are additionally contained in the circuit. C1 serves as a sieve and charging capacitor as in a normal device power supply. It causes the mean value of the DC output voltage from the wind generator with rectification to increase and that pulse currents of the pulse width modulation cannot spread as AC voltages on the connecting lines and emit high-frequency interference waves and cause the wings to resonate.

R1 is in series with the transistor T1 and limits the Current through this, because without R1 C1 turns on when T1 would be suddenly discharged via T1, with over exceeding the maximum drain or collector current T1 would be destroyed. R1 decreases by analog operation T1 strongly the power loss in T1. With pulse width modulation After T1 works, R1 reduces the power loss in the generator. Since R1 itself has a significant amount of ver must perform, it is also referred to as a Heating resistance or burning resistance called. One can actually R1 to the controller via a long cable close and in a particularly warm room, like  e.g. B. the bathroom. Since R1 with pulse width modules tion also begins to vibrate and emit sound, it was also called a (squeaking) squeak resistance net.  

Fig. 2 charging voltage limitation according to the prior art (Parallel controller)

Circuits as in FIG. 2 are currently known, which are intended for the operation of rechargeable batteries with solar cells. Since current flows through the parallel to the solar cells (solar cells connected to terminal 1 (+) and terminal 2 (-)) lying power transistor T1 (which must have a high current amplification factor, e.g. a Darlington transistor or a power MOS -FET must be (depending on the power there may be several)) when the batteries have reached their final charge voltage. The transistor T1 receives its base current from an operational amplifier OP1, which compares the battery voltage with a reference voltage that is generated with R4 and D3. It should be noted that the inverting input is swapped with the non-inverting input because the transistor rotates the phase by 180 °. The diode D1, usually a power Schottky diode, prevents the accumulator from being able to discharge through the solar cells when they are not supplying current.

The disadvantage of the circuit in Fig. 2 is that the power that T1 has to endure is very high. When the battery is full, it is roughly the product of the battery voltage and the current supplied by the solar cells, i.e. the full power that is generated. As a result, huge heat sinks are necessary for T1.

There are other circuits for solar cells after State of the art ver a switching transistor for T1 turn, which is controlled by pulse width modulation. That has the advantage that the power loss in T1 ver is reduced, but the disadvantage that now the loss loss power in the solar cells increases, which makes them hotter will. If you imagine that a wind generator over a rectifier bridge connected to the circuit this point is more important because of a generator is even more sensitive to warming. You'd have to build bigger so that it can withstand the warming. It should Wire cross-section of the winding can be increased  more temperature-resistant glue can be used, and / or tempe stronger magnetic materials are used. Furthermore would be the square wave signal from the switching transistor all over Length of the line and would be radio frequency interference send out. The wings would be in time with the switching frequency radiate unwanted sound. The power transistors T1 are more sensitive to voltage and current peaks than in circuits according to the invention.  

Fig. 3 electronics for pulse width modulation with current and Voltage limitation

Compared to conventional circuits, Fig. 3 firstly offers pulse width modulation instead of analog technology and secondly a current limitation plus voltage limitation, i.e. charging with an IU characteristic instead of a mere voltage limitation.

The combination of current and voltage limitation is achieved in terms of circuitry by an analog OR circuit with D3, D4 and R22. The control voltages are supplied to the two anodes of D3 and D4, and the output signal on the cathodes follows the higher input voltage less the forward voltage of the conductive diode. Higher voltages mean higher power deduction through T1 ( Fig. 1). The same circuit with reverse polarity meaning, reversed diodes and resistance to plus is also conceivable.

The analog OR circuit is followed by a comparator (OP3, R23, R24, R25) connected as a threshold switch, which receives a triangular voltage with a slightly superimposed square-wave voltage (through R4) at its inverting input. The delta voltage is generated by a threshold switch with high hysteresis (OP4, R1, R2, R3), the output signal of which is fed back to the inverting input with an RC element from R5, R4, C1, which means that the arrangement acts as an oscillator at the output PIN14 provides a square-wave voltage and a triangular voltage at C1. At the connection point of R4 and R5 there is a mixed form of triangular voltage plus slightly superimposed square wave voltage. The small superimposed square-wave voltage improves the Umschaltverhal th of OP3 and T1 ( Fig. 1), so that states should be avoided, in which T1 is not completely switched on or off, which would increase the power loss in T1. Other astable multivibrators or oscillators can also be used as long as they provide a triangular voltage of sufficient amplitude (peak-peak approximately half the operating voltage).

The voltage limitation around OP1 compares the battery voltage at the voltage divider R6, R7, R8 with a reference voltage that is generated with R11 and D2. OP1 amplifies the differential voltage with a gain that can be set with the negative feedback with R10 and R9. A gain factor of 100 should be sufficient. The control loop is frequency compensated with C2 and C4 so that it does not start to oscillate. C2 also smoothes the rippled battery voltage. The current limitation around OP2 converts the magnitude of the current in the current measuring resistor R18 into a proportional voltage at OP2 output PIN7, which is fed to the analog OR circuit as a second signal. R18 is a low-resistance shunt with a high load capacity that hangs in the power line to the battery. D1 in FIG. 3 corresponds to D8 in FIG. 1. With a bridge circuit consisting of resistors between the shunt in the positive line and the negative line, the voltage drop across shunt R18 is brought into the common-mode voltage range of the operational amplifier OP2 and can be amplified. R19 and R20 are part of the bridge and together with the negative feedback resistor R21 determine the gain. Since the gain should be large to reduce the control deviation and at the same time R19 and R20 should be high-impedance so that the battery is not discharged through them, R21 must of course be selected to be extremely high-impedance. The other bridge branch with either R12, R13, R14 or R15, R16, R17 is switchable and adjustable. R13 and R16 can be used to preset two different current levels at which the current into the battery is limited, and one of the two limit current levels can be selected with the changeover switch. With a corresponding number of resistors and switching contacts, more than two current values can also be preset. To adjust the current limit, R18 is expediently replaced by a higher-impedance resistor (e.g. 1 ohm instead of 0.01 ohm), so that a power supply unit with usual low currents (e.g. 0-2 A) can be used. From the same for voltage and current limitation is carried out separately, in such a way that a pulse width modulation signal with 50% duty cycle (measured z. B. with an oscilloscope) arises at the respective limit at the output to the gate. C5 in the negative feedback path of OP2 prevents the control loop from oscillating.

C6, D5 and R26 are reverse polarity protection and protection against transient overvoltages. The reverse polarity protection of the circuit in FIG. 3 is not yet complete, however, since the inputs of OP1 and OP2 are not protected against reverse polarity. Three diodes would have to be used for this purpose, with the cathode at the input and the anode at the minus.

Fig. 4 temperature response of the final charge voltage

The end-of-charge voltage of an accumulator is the voltage value to be set on the charger at which the battery can be fully charged, but can remain connected to the charger after being fully charged without overly ga sen. "Gases" means the bubbling of air out of the cells, which occurs when water is split into hydrogen and oxygen. The amount of gas produced is directly proportional to the flowing residual current and the amount of water consumed. In order to keep the batteries as maintenance-free as possible, attempts are made to keep the current as low as possible. The higher the final charge voltage, the greater the residual current, the energy of which is consumed with the help of electrolysis. Common setting values for the final charge voltage at 20 ° C are between 13.8 V and 14.4 V for a 12 V battery. The residual charge current should be between 0.2 A and 0.5 A per 100 Ah battery capacity, in other words, one five hundredth to two hundredths of the battery capacity. As can be seen from Fig. 4, the necessary end-of-charge voltage is strongly temperature-dependent. At low temperatures, much more voltage is required than at high temperatures. The temperature coefficient is approximately -46 mV / ° C for a 12 V battery or, generally speaking, -0.38% / ° C. The value of the temperature coefficient as well as the absolute value of the final charge voltage at 20 ° C is somewhat dependent on the aging condition of the battery. New batteries have a higher end-of-charge voltage and also a higher negative temperature coefficient than old batteries. What is written here, especially the numerical values given, relate exclusively to lead accumulators with sulfuric acid electrolyte. Temperatures below -10 ° C should be viewed with caution, as a deeply discharged battery can freeze and burst at -10 ° C. If you want to reach extremely low temperatures, you must ensure that the batteries are always full during these frost periods and that only a maximum of 20% of the capacity is removed.

Fig. 5 Introduction of sensing lines

When the battery is normally connected with two lines, a line L1 for plus and a line L4 for minus is that the electronics see a higher voltage than on the battery while the battery is charging, because the voltage drop across the line resistors RL1 and RL4 reduces the voltage on the battery. This will make the battery charged slower than would be theoretically possible. The reverse is the case when unloading. The electronics see us less tension in the controller housing than tension on the battery rie is. The consequence of this is that at high discharge currents the load is switched off too early.

To avoid these disadvantages, the internal Bridges Br.1 and Br.2 removed and two sensing lines L2 and L3 directly from the electronics, which is the voltage "sees", placed on the poles of the battery plus and minus. The cross section of these lines can be relative to the high power lines L1 and L4 be very small, e.g. B. 2 × 0.75 mm².

When sensing, it must also be noted that the gate from T1 to its source is not biased too positively. No state may occur in which T1 is continuously conducting. The voltage arrows for a power MOS FET which does not yet conduct at a gate voltage of 3 V have been drawn as an example in the drawing in FIG. 5. Since the low output voltage of the driver IC's (LM124) is below 0.3 V, a maximum of 2.7 V may drop at RL4. Assuming that the charging current is 27 A, the line resistance of the minus line RL4 must be less than 0.1 ohm. With a copper conductor cross-section of 2.5 qmm, this corresponds to a maximum length of 14 meters. If a longer line length is necessary, you can increase the cross-section or you have to come up with something else. This would be a different control of the T1, for example with an optocoupler or more simply with a collector resistor between the gate and source of T1, which is controlled by the collector of a pnp transistor (open collector).

Fig. 6 temperature-dependent reference voltage

The easiest way to obtain a temperature-compensated end-of-charge voltage is to use a circuit for the reference voltage which provides a voltage which has the same relative negative temperature coefficient as the end-of-charge voltage should have. Fig. 6 shows four different circuits with which one can generate a reference voltage with a negative temperature coefficient. In all four circuits Fig. 6a-d, a fairly constant reference voltage is first generated with resistor R1 and Zener diode D1 (or integrated voltage reference D1). A temperature-dependent resistor (R3, R6, R12 or R15) is directly exposed to the ambient temperature or better to the battery temperature. It is good if the temperature sensor is cast in a housing made of a material with good temperature conductivity (i.e. metal). It can be a tube, on one end of which the two connection cables are led out and on the other end a tab is attached with a hole (usually for M6), with the help of which the device can be screwed to a battery terminal.

Circuit Fig. 6a uses a PTC resistor (R3) in the negative feedback between the inverting input and ground. The reference input voltage is at the non-inverting input of the OP. When the temperature rises, the resistance of R3 increases, consequently less current flows through R4, consequently there is less voltage at R4, consequently the output voltage U is lower. A simple mathematical formula: there is no temperature coefficient of U depending on R2, R3, R4 and temperature coefficient of R3. The easiest way is to use values for R2, R3, R4 and calculate the output voltage for different temperatures until you have reached the right temperature coefficient.

Circuit Fig. 6b uses a PTC resistor R6 in the upper part of a passive voltage divider with R5, R6, R7, which is also buffered and amplified with the OP and R8 and R9. The output voltage has a definable negative temperature coefficient.

Circuit Fig. 6c uses an NTC resistor R12 in the negative feedback of an OP's between its output and inverting input. The temperature-stable reference voltage lies at the non-inverting input. When the temperature rises, the resistance of R12 plus R11 decreases, as the voltage drop decreases with a constant current and thus the total output voltage also decreases.

Circuit Fig. 6d uses an NTC resistor R15 as a temperature sensor which lies in the lower part of a passive voltage divider which is connected to the reference voltage of D1. If the temperature rises, the resistance of R15 drops and thus the output voltage drops, which can be buffered and amplified with an OP and R16 and R17.

Fig. 7 Temperature compensated pulse width modulated voltage limitation with power supply for higher supply voltages, with protection against reverse polarity and against transient overvoltages. Exemplary embodiment with values for the components.

The temperature-dependent reference voltage in FIG. 7 is generated with IC2 (LM136 Z2, 5 V) and IC1b (LM124). The circuit device corresponds in principle to Fig. 6a, but is supplemented by a few capacitors (C1, C2, C3, C11), which make interference and PWM residues harmless. In addition to the temperature-dependent reference voltage, the voltage comparator IC1a is also supplied with the battery voltage via the voltage divider R12, R13 and R14. The correct final charge voltage is compared with R13. D1 (e.g. 1N4148) clamps input voltages that are too high to the 13.5 V operating voltage and thus protects the OP inputs from overvoltages. The triangular oscillator with IC1d in Fig. 7 corresponds to OP4 in Fig. 3. More positive input voltages at pin 10 and 1 result in a PWM output signal, which has a higher duty cycle than when the DC input signal is lower.

The power supply in FIG. 7 has been developed further than that of FIG. 3. The higher the output of a generator, the more important it is to switch to higher voltages in order to keep the currents within acceptable limits. The rectifier efficiency also increases with increasing operating voltage. A doubling of the operating voltage halves the rectifier losses with the same generator power. If you use e.g. B. a three-phase compact rectifier bridge with 30 A load capacity, you can use a 300 watt generator at 12 V operating voltage, a 600 watt generator at 24 V operating voltage and a 1200 watt generator at 48 V operating voltage, each with the same rectification power loss of approx 50 watts.

The reverse polarity protection in Fig. 7 is made D2, a rectifier diode in series to the terminal of the battery in through laßrichtung. If the battery is connected with the wrong polarity, simply locks D2 and keeps the destructive negative voltage away from the power supply of the circuit and from the control inputs (IC1a Pin3). The load cut-off is connected with the same protection. A possibly changing voltage drop at D2 leads to changes in the end-of-charge voltage, switch-off voltage and switch-on voltage, but is insignificant since the voltages only change in the range of 0.1 V-0.2 V.

Protect against overvoltages D5, D1 and D4. D5 is a Z Diode or Transil diode, which after D2 parallel to the input lies. The Z voltage of D5 should be slightly above the maxima len battery voltage (approx. 20%), whereby too term is that the final charge voltage at extremely low Temperatures rise sharply. The input of IC1a (Pin3) is protected by D1 and D4. The current overflows D2, R14, D1, D4.

T1, D3, C8, R20, R21, R22 supply the electronics with a stabilized operating voltage. Series resistor R21 and Z- Diode D3 stabilize the voltage across R22 to the Base of T1 is given. T1 is a Darlington transistor medium power, its maximum collector-emitter chip voltage should be greater than the Z voltage of D5. With this Circuit it becomes possible to set the battery voltage above 24 To make volts without the IC's due to excessive voltages (The maximum voltage of the standard IC's is usually 36 V) destroyed would.

R20, a resistor in the collector line from T1, protects the transistor from current peaks. R20 must be chosen this way that the voltage at the emitter of T1 is not related breaks when the maximum consumption current flows, which would rend switching of the bistable relay becomes.

C8, C9, C10 smooth the operating voltage and provide an energy reserve during current peaks. Care must be taken that the voltage of the transil or Z diode D4 at the emitter of T1 must be at least as high or slightly higher than the voltage of the Z diode D3 at the base, since normally no current should flow through D4 . R22, the base resistance of T1, prevents the tendency to oscillate, since without the base R22 T1 would be at ground potential in terms of radio frequency, i.e. could oscillate in the base circuit ( FIG. 7).

Deep discharge protection for accumulators Fig. 8 prior art

Function of the circuit: A threshold switch as implemented in FIG. 8 with an operational amplifier or otherwise, e.g. B. with a Schmitt trigger, compares the battery voltage (divided battery voltage) with a reference voltage produced with a Zener diode with a series resistor. The output of OP1 has high potential if the battery voltage exceeds a certain value. By coupling via R5 and R6 from the operational amplifier output to the non-inverting input, a switching hysteresis is obtained, which means that the load is reduced when the battery voltage falls below z. B. 10.5 V, and only switched on again when z. B. 12.5 V are exceeded when recharging or the battery recovers. The OP output supplies the base current for T1 via R7, with which the collector coil current is applied to the relay coil current. The relay switches on the load with its contact S1.

Disadvantages of the circuit according to the prior art:

1. The relay drive current is very high with z. B. 0.24 A. This current is constantly the battery when the battery is full removed until it is empty. With a battery capacity from Z. B. 100 Ah is a full battery after 400 hours deep discharge if no load current flows. Well could someone come up with the idea instead of a work contact to use a normally closed contact for S1. That’s it Problem just postponed. Now there is almost no flow Electricity as long as the batteries are full, but they are once empty, the relay coil current discharges them even further, so that they can be recharged by deep discharge to lose. By sulfating the positive lead plates the batteries become unusable during deep discharge.

2nd disadvantage

In the adjustment work on a circuit according to Fig. 8 with means of adjusting resistors R3 for the voltage level and the switching hysteresis with R6 for the two components interfere with each other. It is therefore very difficult, at least protracted, to set the exact switch-off voltage and switch-on voltage.

3. Disadvantage

With load resistors that have a high starting current, the circuit in Fig. 8 can switch off prematurely, with tilting vibrations occurring as in an astable multivibrator, which apart from the fact that the load does not remain switched on and starts up, the relay contacts wear out too much. Such load resistors are e.g. B. motors and incandescent lamps. If switching off and multivibrating is not due to the inrush current, but to an excessive continuous current, there is nothing you can do but increase the battery capacity to ten times the value of the continuous current.

Fig. 9 load cut-off electronic fuse

The circuit in FIG. 9 shows a load cutoff in which a power MOS FET (T1) serves as a switching element. This is controlled at the gate by a comparator, which is switched as a threshold switch. This comparator (ICI) compares the battery voltage with the reference voltage of a Zener diode (D1) in such a way that if the voltage of the battery drops when it is deeply discharged, the power MOS-FET (T1) quickly takes away the gate voltage between the gate and source, so that it locks. R11 sets the switching voltages at which the load is switched on or off. The threshold switch behavior of the comparator is achieved by feedback via R1 and R14 between the output and the non-inverting input. These two resistors determine the hysteresis. Due to its voltage-retaining effect, the capacitor C1 prevents the load from being switched off in the event of short-term (approximately 1-2 s) voltage dips in the battery when switching on loads with high starting currents such as e.g. B. motors, incandescent lamps, fluorescent lamps and power supply capacitors. On the other hand, the maximum drain current of T1 must not be exceeded for a short time. The fast electronic short-circuit protection with R2, T2 and V1 is used for this. R2 acts as a low-resistance measuring resistor through which the consumer current flows. Above the voltage drop across R2 exceeds the base-emitter voltage of T2, at which this turns on, the collector current of T2 ignites the thyristor V1 via R5. V1 thus removes the gate-source voltage from T1 and T blocks quickly.

Since it is in practical operation in the event of radio interference such as this occur when switching inductive consumers, too un desired shutdowns came, the radio interference suppression R6, C4, R5, C5 inserted. As a guide to dimensioning can apply: There is a delay of the switch-off by 1 µs too long for the modern power MOS FETs. Persistent interferers are, for example: fluorescent lamps (Mains), motors, transformers in undisturbed execution when switching off. To protect the MOSFETs from voltage  Spikes are still high voltage arresters, varistors (R13) and transile diodes between source and drain and par allel required on the load side. A light emitting diode D2 with Series resistor R7 parallel to the load (same as output) shows whether the load is getting tension. For restarting a switch in parallel with the thyristor shorts it.  

To Fig. 10 load shutdown with bistable relay

Task: The load switch-off has the task, the battery (the accumulators) from damage due to deep discharge protect. The battery voltage is used as the control variable used. A temperature compensation of the switch-off chip Because of the behavior of accumulators, voltage is not necessary tig. This means that the deep discharge threshold is always on can leave the same voltage level, regardless of which temperature of the batteries. In contrast, the low Discharge threshold (discharge) dependent on current (internal resistance). With the final charge voltage limitation for the (lead ) Accumulators, on the other hand, is a temperature dependency of the Final charge voltage strongly advised.

Circuit Fig. 10

R1, R2, R3, R4, R5 and C1 form an adjustable span divider. The reclosing voltage can be switched on with R3 (For a 12 V battery, for example, to 12.2 to 12.5 V), and R4 is used to set the battery voltage the load is switched off (voltage e.g. 10.5 V to 11.5 V for a 12 V battery). The capacitor C1 gives that Voltage divider low pass characteristics and prevents brief voltage drops in the battery, for example when Switch on incandescent lamps or motors to switch off to lead. One will choose its value so that a time con constant of about one to ten seconds is reached.

Series resistor R6, Zener diode D1 and C2 generate a stable reference voltage, which is supplied to OP2 at the inverting input and OP1 at the non-inverting input. The stabilized voltage can be selected within the common-mode range of the OPs, but expediently from approximately 0.2 _* U battery to 0.7 _* U battery. In the exemplary embodiment in FIG. 10, it would be 0.5 _* U battery. The two operational amplifiers are as a threshold switch with a low hysteresis of z. B. 0.005 _* U battery switched. The output of OP1 jumps to high when the input of the battery voltage divided by the voltage divider inverts below the reference voltage, while OP2 jumps to high if the divided battery voltage supplied to its non-inverting input exceeds the reference voltage.

The operational amplifiers follow two similarly built pulse separation and amplifier stages, consisting of C4, D2, R11, R12, T1, D4 and C3, D3, R13, R14, T2, D5. The Functions are as follows: T1 and T2 are switching transistors for the relay windings, which do not continuously operate as usual but only briefly after a low to high over output of the operational amplifier outputs. T1 is over by OP1 C4 and R12 controlled and switched off the load by a current surge on the winding W1 of the bista cheap relay there. T2 is indicated by OP2 via C3 and R14 controls and gives a current pulse to the switching on Winding W2 of the bistable relay when the output of OP2 gets high. The transistors T1, T2 are in emit circuit. The freewheeling diodes D4 and D5 are located parallel to the relay windings W1 and W2 and protect the Transistors T1 and T2 before mutual induction voltages from W1 and W2. D2 and D3 limit the base reverse voltage of the Transistors T1 and T2 to safe values. If OP1 and OP2 has no internal short-circuit protection of the outputs there would have to be resistance between the operations amplifier outputs and the diodes are inserted, or the cathodes of the diodes are placed directly on the base of T1, T2. R11 and R13 leave the reverse currents of C3 and .4 and from the collector base routes from the bases against Drain mass. R12 and R14 are in series between the Bases and the OP outputs with C4 or C3 together and be limit the base current (if this is not already the case) by the short circuit protection of the operational amplifier and on the other hand they extend the pulse time. C4 and R12 or C3 and R13 determine the pulse time that the safe switching of the relay lie between 100 ms and 1 s should. In addition to blocking the operating voltage, C5 also has the task of storing so much charge that the Re  relay can still switch to "Off" if the event occurs should that the line for electronics is broken is not under while the line of the load circuit will break. Wouldn't the relay be switched to "off" there is a risk that the battery will be over-discharged becomes.

The advantages of the circuit presented in FIG. 10 are a total of three. First: The current consumption is due to the fact that no continuous currents flow through the windings of the bistable relay, but only brief pulses when actuated, very low. Second: The switch-off voltage and the switch-on voltage can be set separately from each other (with R4 and R3) and the circuit can thus be adapted well to different load cases. Thirdly, the adjustment work is easier to carry out than in the case of circuits according to the prior art, since hysteresis and absolute values of the thresholds do not influence one another.

Fig. 12 speed display

The speed display in Fig. 12 has the task of displaying the current speed of the wind generator without additional sensors and without additional lines to the wind generator. Since the speed is roughly linearly related to the wind speed via the speed factor, you can get a rough overview of the power curve as a function of speed and wind speed with additional power measurement.

The connection of the speed indicator to the generator must be mass-free, because otherwise the negative terminal of the battery would be short-circuited with a phase of the generator, which only would if the generator in a star connection with neutral conductor to ground plus three rectifier diodes for the three phases (as in Figure . 13) would be switched. In order to make a Mas connection to the generator superfluous - according to the invention a symmetrical input stage is connected upstream, the two inputs of which are connected to two phases of the generator.

The task without additional sensors and cables get along is solved by the fact that the synchronous genes alternator output frequency before the rectifier tapping and into a span proportional to the frequency voltage or electricity converted for a moving coil instrument becomes. The main components of the invention include one monostable multivibrator around IC2, a needle pulse former with C6 and IC1a to control the monostable Multivibrators with needle impulses, a threshold switch with IC1b to generate a square wave voltage for the Na delimpulsformer, and from a balanced input around IC1c and IC1d around.

Symmetrical input around ICIc and IC1d: IC1c and IC1d are voltage followers with a gain factor goal equal to one. It is important that the input stages must not be overdriven, otherwise IC2 will trigger several times and a too high speed value is displayed. That's why R1 and R3 or R2 and R4 serve as voltage  divider. C1 and C2 block DC voltage, but leave low frequency alternating currents from 0.5 Hertz through. Higher frequencies Zen over 100 Hertz are suppressed by C3 and C4, because Interference frequencies, for example the pulse width modulation frequency frequency from the charge limit (e.g. 1 kHz) The diodes D1, D2, D3, D4 clamp the input voltage to positive or negative operating voltage of IC1 in order to Protect inputs from IC1d and IC1c from overvoltage. R5 and R6 halve the operating voltage, (C5 serves as a sieve capacitor), and via R3 and R4 the inputs of IC1d and IC1c biased with half the operating voltage.

DC chip is then at the output of IC1d and IC1c voltage at half the operating voltage level (with operating voltage the output voltage of the voltage regulator IC3 is meant) plus an alternating voltage, which at the outputs (IC1 Pin8 and 14) the same height, but out of phase. The signal at IC1 Pin14 and 8 is now one using IC1b Signal merged and at the same time in a rectangle gnal transformed. For this the signal from IC1d (Pin14) is the positive input (pin5) of IC1b via R7 and the signal from IC1c (Pin8) via the voltage divider R9 and R10 to the negative input (pin6) of the operational amplifier IC1b ge leads. By coupling with R8 from the output of the ICIb (Pin7) to the non-inverting input (Pin5) Threshold switch behavior set. To achieve ei ner optimal common mode rejection, it is important that R8 / R7 = R10 / R9. Nevertheless, the function is also with one grant another arrangement or a different ratio provides, as long as R8 is greater than 100 × R7, then R9 = 0 ohm and R10 = infinite ohms. Now that becomes rectangular Differentiated output signal from IC1b, namely with C6 and R11 parallel to R13. It is a pulse duration at the output of Aim for IC1a from 1 µs to 100 µs. The pulse duration must be un below the period of the frequency of the wind generator lie. Since the frequency of the wind generator is usually low (below 200 Hz), the pulse duration can be relatively long to get voted. IC1a amplified, switched as a comparator, the differentiated square wave signal to a square wave pulse,  with which the monostable multivibrator (IC2) is controlled becomes. When the output idle signal from IC1a (Pin1) is positive should be with negative impulses (like the example IC "555" you would need), the non-inverting input biased more positively by IC1a than the inverting one Entrance. Conversely, if a negative resting potential with positive impulses for the monostable multivibrator the inverting input of IC1a would have to be posi be more biased than the non-inverting input. C7, the capacitor at the non-inverting input, holds Distant interference pulses, but may not be necessary under certain circumstances. The DC voltage presetting of the inputs is done by the two voltage dividers, R11 and R13, and R12 and R14.

The following circuit section, the monostable multivibra tor around IC2, in principle transforms the input pulses into longer-lasting output pulses, which, however, have a constant duration and level. As a result, the mean value of the output voltage increases in proportion to the input frequency. This pul sizing output voltage is smoothed with voltage dividers and capacitors and is given to the moving-coil instrument M1 or to the output terminal 6 for a recorder, which then records or plots the speed as a function of time or another variable. R17 and R21 at the output of IC2 are the adjustable series resistor for the current measuring device M1, which should preferably be a rotary instrument with a linear scale so that the speed can be read linearly. C10 smoothes the pulsating current, whereby a small time constant of, for example, 0.2 seconds is sufficient, since the moving coil instrument will normally react very slowly. It is different with the recorder output, here the time constant should be higher, for example 0.5 to 2 seconds, since the recorder pens any voltage ripple, especially at low frequencies, i.e. speeds less than 300 rpm. C11 is the smoothing capacitor for the recorder output (Kl.6) and works together with the voltage divider R18, R22 and R19. With R22 you can fine tune the output voltage. The rough adjustment of the output voltage is carried out by adjusting the pulse length with R20 and C9. The pulse length must be less than the minimum period of the wind generator frequency. R16 protects the IC2 internal discharge transistor from excessive current peaks. You can find out more about how the 555-IC works from a data book from Raytheon. C8 decouples the IC2 internal reference voltage before voltage peaks. R15 increases the maximum output voltage swing of IC2. In this context I would like to point out that the output voltage swing of IC's of the type "555" and that of the C-MOS version "7555" are different, so that a new adjustment is necessary when replacing. In addition, the IC's of type "555" or "7555" by the IC-in internal collector-emitter residual voltages also stood at rest, with the wind generator stopped, a slight output voltage, which can falsify the measurement result. There are the following remedies: The zero point can be mechanically corrected on the M1 moving instrument, and there is certainly a button on the recorder for zero point correction. However, if you do not have these correction options, you can also compensate electrically, for example by using a voltage divider between the operating voltage and ground, whose minus-side trim potentiometer provides a voltage that is equal to the residual voltage of the IC2. The measuring devices are then connected between the two equally high voltages, thus avoiding the zero point error. In addition, an operational amplifier would be conceivable as an output stage, which could take over zero correction and smoothing at the same time.

Fig. 13 control circuit for separately excited wind generator (Type: car alternator)

A three-phase alternator with its three phases U, V, W in a star connection with a star point Mp was shown as an example in the circuit in FIG. 13. The positive terminal of the excitation winding is J, while the negative is K. T1 and the circuit around IC1b is the usual regulation, as used for example in motor vehicles. The diodes D4, D5, D6 rectify the three-phase current and thus charge the battery or supply connected consumers. D1, D2 and D3 also rectify the three-phase current, but thereby supply the control circuit and the field winding with current. So that there is power for the control circuit after the (wind) generator has come to a standstill or when the excitation current is switched off, D11 supplies the control circuit with current from the battery.

So that no excitation current flows when the wind generator stands still or runs so slowly that it has its own er The exciter cannot apply the excitation current itself current below the economic speed be tested. In alternative literature, there is a wind about this described pressure switch, which in the simplest case from egg ner metal plate that is perpendicular to the wind direction is in contact and at higher wind speeds closes, which then allows the excitation current to flow. The After parts of this process are its inaccuracy and Noise development of the striking metal plate, especially al lem with gusty wind. There are also speed and windge speed in gusty wind not synchronized in time, because the wind generator ver runs up late.

The invention solves the problem in an elegant manner. The speed is determined using the tachometer. The output voltage of the tachometer ( FIG. 12), which is proportional to the speed, is compared with a reference voltage (D10) using a threshold switch (IC1a). The output signal of the threshold switch is high if the speed is too low and low if the speed is high enough. The base of a switching transistor (T2), which takes away the base current for T1 when the speed is too low (IC1a PIN1 = High), is thus controlled via R3 and Zener diode D9. The excitation current then drops to zero because T1 no longer conducts.

The speed proportional output voltage of the speed knife is well screened (C1) the inverting input a comparator or operational amplifier (IC1a). This comparator compares this voltage with that Voltage at the non-inverting input (PIN3), the on is with the voltage divider R6, R7, R8, its Receives reference voltage from D10. With R7 the pathogen can current switch-on can be set exactly. Because the windge nerator initially runs empty when the exciter is switched on but is loaded and therefore slows down is one Hysteresis in switching behavior required. This hysteresis is with a positive feedback between output and noninver ting input of IC1a with R5 and R4 reached with R4 is adjustable.

One more word about the speed display: your function is at sufficient input sensitivity ensured, because the residual magnetism in the rotor a sufficiently high unbela provides constant output voltage. if the output voltage generated by the residual magnetism for control of the speed display is not sufficient, (Fig. 12), the input voltage divider ratio of R1: R3 and R2: R4 are lowered and possibly additionally a voltage gain of the IC1d and IC1c with a Ge gene coupling with 3-4 resistors are introduced.

And one more word about the generator: Because wind turbines are smaller Have speeds as internal combustion engines, must in the generator accordingly the number of turns can be increased or it is install a gearbox between the wind turbine and the generator.

Claims (17)

The invention includes circuits for generating and storing generation of electrical energy with the help of accumulators, Solar, wind, and other powered generators. 1. Claim: Interconnection of generators, in particular solar and wind generators for the common processing of energy, characterized in that the direct current generator (solar cells for example) is decoupled by one or more diodes D7 and protected against reverse current, during which by rotating one AC or three-phase generator generated current via a rectifier D1-D6 (three-phase or alternating current rectifier) is connected to the node (see Fig. 1). 2. Claim: A parallel control with a variable shunt from T1, R1, C1 and D8, characterized in that T1 and R1 are located one behind the other, but are parallel to the node with the energy generators and are pulse-width modulated or controlled by electronics so that they withdraw electricity from energy suppliers when the battery threatens to be overcharged. C1 is also parallel to the node, which serves as an energy storage device, in order to clean the voltages on the lines from AC voltages of the PWM ( FIG. 1). Furthermore, surge protection components are located between the drain and source (or collector and emitter) of T1, which prevent T1 from being destroyed by voltage peaks when switched off, implemented with an RC element, where R is a low-inductance resistor with the same value as R1, but with a lower load capacity , and / or with other overvoltage protection components in parallel, such as Transil diodes or VDR resistors. 3. Claim: Electronics for current and voltage limitation with pulse width modulation, consisting of a triangular oscillator (OP4), a pulse width modulator (OP3), a current limit with current measuring resistor (OP2), a voltage limit (OP1) and an analog OR circuit with D3 , D4, R22 ( Fig. 3), characterized in that both a charging current which is too large and a charging voltage which is too high lead to a pulse-width-modulated output signal whose "on" time increases, so that more current via R1 and T1 ( Fig. 1) drains away. 4. Claim: Circuits according to the preamble, characterized in that the small-power electronics boards that "see" the battery voltage, are connected directly to the battery terminals via two additional "sensing" lines (L2, L3) so that the voltage drop the high-current lines (L1, L4) does not lead to falsification of the voltages to be measured and regulated (end-of-charge voltage, switch-off voltage, switch-on voltage) ( FIG. 5). 5. Claim: Introduction of a temperature dependency of the final charge voltage according to the needs of the battery mulators, solved according to the invention in that the reference voltage for the final charge voltage limiting stage has a corresponding temperature response which is generated by four different circuits according to Fig. 6a-d, hot or PTC thermistors with preferably linear temperature response can be used, which measure the temperature of the batteries by exposing them to their ambient temperature or by attaching them directly to the battery in a thermally conductive manner. The change in resistance of the temperature-dependent resistors is evaluated using one of the four methods and the reference voltage is generated therefrom with the correct (negative) temperature coefficient, as described below in patent claims 5a-d. Claim 5a: Circuit for generating a temperature-dependent reference voltage, characterized in that a PTC resistor (R3) lies in the negative feedback between the inverting input and ground of an operational amplifier, which amplifies the reference voltage (D1) ( Fig. 6a). Claim 5b: Circuit for generating a temperature-dependent reference voltage, characterized in that a PTC resistor (R6) is connected in the upper part of a passive voltage divider which depends on the reference voltage (D1) ( Fig. 6b). Claim 5c: Circuit for generating a temperature-dependent Reference voltage, characterized in that a NTC resistor (R12) in the negative feedback between OP-off gang and inverting input of an operational amplifier lies, which amplifies the reference voltage (D1). Claim 5d: Circuit for generating a temperature-dependent reference voltage, characterized in that an NTC resistor (R15) is in the lower part of a passive voltage divider, which is connected to the reference voltage (D1) ( Fig. 6d). 6. Claim: Four circuits according to claim 5a-d, but with the difference that the output voltages no negati ven, but have a positive temperature coefficient, characterized in that the NTC resistors R12 and R15 against PTC resistors and the PTC resistors R3 and R6 ge NTC resistors are replaced. 7. Claim: reverse polarity protection and protection against transients over voltages and voltage stabilization for the operation of electronic circuits according to the invention on battery voltages that exceed 12 V, characterized in that

• a) reverse polarity protection is taken over by a diode (D2) in the forward direction with correctly polarized operation, which is the foremost component to which the operating voltage comes, and in the event of a failure of D2 (high reverse current), the transil diode D5 in the forward direction maximum negative voltage limited to 0.7 V ( Fig. 7).
• b) the protection against (transient) overvoltages by the Transil diode D5 is arranged parallel to the input in the reverse direction, while permanent overvoltages may be present due to a high collector-emitter voltage of T1 and the input of IC1a (voltage detector) from a combina tion of D1 (diode) and D4 (Z diode or Transil diode above the stabilized supply voltage, with a higher Z voltage than this) is protected ( Fig. 7).
• c) the voltage stabilization consists of series resistor R21 and Zener diode D3, base resistance to oscillation tendency R22, (Darlington) transistor T1 and current limiting resistor R20 as well as blocking capacitors C8, C9, C10 ( Fig. 7), which the battery voltage to a for the Lower the voltage of the semiconductor devices.

8. Claim: Invention of a load switch-off in which a Po-MOS-FET (T1) serves as a switching element which is controlled by a threshold switch (IC1), characterized in that an electronic overcurrent protection with R2, T2, R5, V1, S1 and other auxiliary components, the switching transistor T1 suddenly and memory blocks when the maximum current is exceeded (which is determined with R2), and the shutdown can be reset with S1. ( Fig. 9). 9. Claim: Invention of a load switch-off with bistable relay with two windings, characterized in that the relay per winding with its own threshold switch stage (OP1 or OP2), pulse isolating stage (C3 or C4) and switching amplifier stage in C mode (T1 or T2) , in which no quiescent current flows, is controlled so that the shutdown voltage and reclosure voltage can be set separately with R4 or R3. ( Fig. 10). 10. Claim: Tachometer with analog output voltage or current, consisting of a monostable multivibrator (IC2), which is controlled by a needle pulse generator (IC1a, C6), which in turn is controlled by a threshold switch (IC1b) with a symmetrical input (IC1c, IC1d ) is driven, characterized in that the two connections of the symmetrical input two phases of the three-phase or alternating current generator can be supplied without the ground potential of the tachometer circuit having to be connected to the generator ( Fig. 12). 11. Claim: Tachometer according to claim 11, however ge characterized by the fact that on symmetrical input stage and threshold switch (IC1b-d) a microprocessor or Frequency counter follows, which counts the pulses digitally and on shows. 12. Claim: Control circuit for separately excited (wind) generators, characterized in that using a speed indicator ( Fig. 12, claim 10) and a threshold switch (IC1a) and a switching transistor (T2) the current through the excitation winding at uneconomical use (too low) speeds is switched off. ( Fig. 13).