EP0127101B1 - Inverter for feeding discharge lamps - Google Patents

Inverter for feeding discharge lamps Download PDF

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Publication number
EP0127101B1
EP0127101B1 EP84105786A EP84105786A EP0127101B1 EP 0127101 B1 EP0127101 B1 EP 0127101B1 EP 84105786 A EP84105786 A EP 84105786A EP 84105786 A EP84105786 A EP 84105786A EP 0127101 B1 EP0127101 B1 EP 0127101B1
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EP
European Patent Office
Prior art keywords
voltage
inverter
capacitor
control
lamp
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EP84105786A
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German (de)
French (fr)
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EP0127101A1 (en
Inventor
Peter Ing. Grad. Krummel
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Siemens AG
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Siemens AG
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Priority to AT84105786T priority Critical patent/ATE25800T1/en
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    • H—ELECTRICITY
    • H05—ELECTRIC TECHNIQUES NOT OTHERWISE PROVIDED FOR
    • H05B—ELECTRIC HEATING; ELECTRIC LIGHT SOURCES NOT OTHERWISE PROVIDED FOR; CIRCUIT ARRANGEMENTS FOR ELECTRIC LIGHT SOURCES, IN GENERAL
    • H05B41/00—Circuit arrangements or apparatus for igniting or operating discharge lamps
    • H05B41/14—Circuit arrangements
    • H05B41/26—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc
    • H05B41/28—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters
    • H05B41/295—Circuit arrangements in which the lamp is fed by power derived from dc by means of a converter, e.g. by high-voltage dc using static converters with semiconductor devices and specially adapted for lamps with preheating electrodes, e.g. for fluorescent lamps
    • H05B41/298—Arrangements for protecting lamps or circuits against abnormal operating conditions
    • H05B41/2981—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions
    • H05B41/2985—Arrangements for protecting lamps or circuits against abnormal operating conditions for protecting the circuit against abnormal operating conditions against abnormal lamp operating conditions

Definitions

  • the invention relates to an inverter according to the preamble of claim 1.
  • the inverter is adapted to a certain lamp power, the operating frequency of the inverter and the inductance of the inductor of the series resonant circuit being dimensioned at this operating frequency so that the discharge lamp receives the current required for its nominal power ; a very specific operating voltage is assumed.
  • Discharge lamps with the same nominal power have been on the market for some time now, which differ in their gas filling (previously only krypton, now also argon) and thus also in their burning voltage: Discharge lamps with argon filling have a significantly higher burning voltage. You would therefore consume an impermissibly high power with an inverter designed for a Krypton lamp.
  • the invention is therefore based on the object of designing the inverter so that it can be used for supplying lamps of the same nominal power regardless of their gas filling and thus of their operating voltage.
  • the inventive solution to this problem is characterized in claim 1.
  • the headset can have a pulse generator, the frequency of which increases with an input variable derived from the operating voltage of the discharge lamp: if this characteristic curve is designed accordingly, the frequency matching the respective operating voltage and thus the required lamp current are automatically set.
  • the frequency of the pulse generator can also be switchable, depending on whether certain threshold values of the operating voltage have been reached or fallen below.
  • Sufficient preheating of its electrodes is usually desirable before the discharge lamp is ignited.
  • it is known to increase the operating frequency of the inverter during a preheating time to such an extent that it is at a sufficient distance from the resonance frequency of the resonance circuit so that the voltage at the lamp is not sufficient for ignition. After this preheating time has elapsed, the operating frequency is reduced and approximated to the resonance frequency until the voltage at the lamp is sufficient for ignition.
  • an evaluator is provided between the headset and the combustion voltage sensor, to which the voltage of a timing element located at the input of the inverter is also fed and which during a preheating time beginning when the inverter is switched on and a subsequent ignition time determines the frequency of the control voltages and which then allows the output signal of the internal voltage sensor for the headset to take effect.
  • the operating frequency of the inverter is therefore not determined by the operating voltage but by the timing element, which in the simplest case is an RC element or a monostable multivibrator.
  • the switches of the inverter are field-effect power transistors T1, T2, which are connected in series to the DC voltage source Q.
  • the load circuit with a reversing capacitor C10, the discharge lamp LP and the series resonance circuit with the capacitor C9 and the choke L1 is parallel to T1, the capacitor C9 of the series resonance circuit being located between the heatable electrodes of the lamp.
  • T2 When T2 is activated, the load circuit is located at source Q and reloads in the following half-wave via activated T1.
  • a control set S with a pulse generator g, which generates control pulses with an adjustable frequency, is used to alternately control the transistors T1 and T2; these are alternately applied to the two transistors via a pulse distributor i.
  • the frequency of the pulses of the generator g is dependent on the output signal of the evaluator A, which is connected on the input side to a combustion voltage sensor B and to a timing element Z with two outputs z1, z2, and which essentially has the function of one of these inputs, depending on the operating phase to connect to the pulse generator g.
  • the evaluator A After switching on the DC voltage source Q, the evaluator A activates the first output z1 of the timing element Z, the voltage of which ensures the maximum operating frequency f v between the times t o and t 1 (preheating time): see FIG. 2. At this frequency, the Do not ignite the lamp. At time t 1 , the evaluator A - controlled by Z - switches on the second output z2 of the timing element, which supplies a smaller or a continuously decreasing voltage up to time t 2 : the associated lowering of the operating frequency leads to the vicinity of the resonance frequency of the resonance circuit and thus to the ignition of the lamp.
  • the evaluator After the time t 2 , the evaluator finally switches over to the operating voltage sensor B, so that an output signal dependent on the operating voltage of the lamp is then present at the pulse generator, which has an operating frequency f BA higher for an argon lamp and a lower operating frequency for a krypton lamp F BK (dotted line in Fig. 2) results.
  • a saturation transformer L2 is essentially used as the control set, the primary winding L21 of which is arranged in the load circuit and the secondary windings L22, L23 of which are connected to the control paths of the transistors T1, T2.
  • the load circuit and the saturation transformer are dimensioned in such a way that the lowest possible operating frequency, that is to say that for supplying a krypton lamp, is obtained without the interventions according to the invention described below.
  • the turn-through time of T2 is shortened with the aid of blocking transistors T3, T4, by which the control path of T2 is short-circuited.
  • the emitter of T4 is connected to a synchronizing capacitor C3, which is connected via a series resistor R2 to the DC voltage source Q and via a series capacitor C2 to the switching path of the transistor T2 to be blocked prematurely.
  • the course of the sawtooth voltage Des at the emitter of the blocking transistor T4 is shown in FIG. 4.
  • T4 lies on a voltage divider with the resistors R3, R4 and a comparison capacitor C4, on which a reference voltage which can be controlled in size occurs.
  • C4 is connected in parallel to a control transistor T5 and is also connected via R5 to an auxiliary voltage source which is formed by a capacitor C5 which is connected in parallel with the synchronization capacitor C3 via a charging diode D4.
  • the current of the load circuit continues to flow in the same direction, namely via the reverse current diode of transistor T1 (integrated in MOS-FET transistors).
  • the saturation transformer at L22 then generates a control voltage for T1, via which the discharge current of the reversing capacitor C10 can flow through the load circuit in the opposite direction.
  • the duration of this half-wave depends on the saturation transformer and is essentially constant; in this case, the change in the resulting operating frequency is only achieved by shortening the one half-wave determined by T2.
  • the magnitude of the reference voltage U c4 at the comparison capacitor C4 can be set differently by the control transistor T5 connected in parallel.
  • the control path of this transistor is connected to a voltage divider with resistors R6 and R7 and a Zener diode D8 and a voltage divider R8. R9 on a capacitor C8, which charges via D9 or D10, C7 and R12 to a value dependent on the lamp voltage.
  • R6 and R7 are connected in parallel to a capacitor C6, which is connected via a decoupling diode D6 to an RC element R1, C1 and is charged by the DC voltage source Q via resistors R11, R10 and an electrode of the lamp LP.
  • the voltage at C1 controls the transistor T2 via the switching diode D2 when the inverter is switched on, but then disappears because C1 is discharged via D1 and T2. Therefore, the decoupling diode D6 is blocked after the inverter starts to oscillate:
  • the control transistor T5 is therefore only activated after the inverter has started to oscillate during a short discharge time of the capacitor C6, which determines the preheating time with the resistors R6 and R7.
  • the blocking transistor T4 becomes conductive in each period due to the low reference voltage U c4v in time t S1 (FIG. 4) and thus the half-wave flowing via T2 is shortened.
  • the resulting increase in the operating frequency results in a correspondingly low voltage across the capacitor C4 in parallel with the discharge lamp LP, so that it cannot ignite.
  • T5 is increasingly controlled and thus the reference voltage at C4 increases, so that the half-waves flowing over T2 become longer and longer until the resulting operating frequency has approached the resonance frequency of the load circuit to such an extent that the lamp ignites.
  • T3, T4 and T5 are then blocked and the operating frequency is the same long half-waves, which is determined exclusively by the saturation transformer and is dimensioned for the operation of a Krypton lamp.
  • the voltage occurring at the capacitor C8 is too low to be able to again control the control transistor T5 via the zener diode D8.
  • the voltage at the capacitor C8 is also used to switch off the inverter when the lamp is permanently unable to ignite: If this voltage reaches a limit value specified by the monitoring device Ü, then this responds and short-circuits the starting capacitor C1 and the control electrode from T1 to D5.
  • the response threshold of this monitoring device is above the voltage that arises at C8 when the inverter is operated with an argon lamp.
  • the charging time constant of C8 is so great that the monitoring device cannot respond during the preheating and ignition phases, unless the lamp would not ignite in the specified time (t 1 / t 2 ).

Abstract

1. An inverter comprising two switches (T1, T2) which can be alternately rendered conductive, and comprising a load circuit arranged parallel to the first switch (T1) and connected via the second switch (T2) to a d. c. voltage source (Q), and consisting of the series arrangement of a reversible oscillatory capacitor (C10), a series resonant circuit composed of a choke (L1) and a capacitor (C9), and a discharge lamp (LP) equipped with heatable electrodes, where these electrodes are arranged in the load circuit and are connected to one another via the capacitor (C9) of the series resonant circuit, and comprising a control set (S) which supplies control voltages which alternately render the switches conductive, characterised by a burning voltage sensor (B) which monitors the burning voltage of the lamp (LP) and supplies an output signal dependent upon the burning voltage and fed to the control set (S) to determine the frequency of the control voltages which it supplies.

Description

Die Erfindung betrifft einen Wechselrichter gemäß Oberbegriff von Anspruch 1. Der Wechselrichter ist dabei einer bestimmten Lampenleistung angepaßt, wobei die Betriebsfrequenz des Wechselrichters und die Induktivität der Drossel des Serienresonanzkreises bei dieser Betriebsfrequenz so bemessen sind, daß die Entladungslampe gerade den für ihre Nennleistung erforderlichen Strom erhält ; dabei wird von einer ganz bestimmten Brennspannung ausgegangen.The invention relates to an inverter according to the preamble of claim 1. The inverter is adapted to a certain lamp power, the operating frequency of the inverter and the inductance of the inductor of the series resonant circuit being dimensioned at this operating frequency so that the discharge lamp receives the current required for its nominal power ; a very specific operating voltage is assumed.

Seit einiger Zeit sind nun Entladungslampen gleicher Nennleistung auf dem Markt, die sich durch ihre Gasfüllung (früher nur Krypton, jetzt auch Argon) und damit auch durch ihre Brennspannung unterscheiden : Entladungslampen mit Argon-Füllung haben eine deutlich höhere Brennspannung. Sie würden daher mit einem für eine Krypton-Lampe ausgelegten Wechselrichter eine unzulässig hohe Leistung aufnehmen.Discharge lamps with the same nominal power have been on the market for some time now, which differ in their gas filling (previously only krypton, now also argon) and thus also in their burning voltage: Discharge lamps with argon filling have a significantly higher burning voltage. You would therefore consume an impermissibly high power with an inverter designed for a Krypton lamp.

Der Erfindung liegt daher die Aufgabe zugrunde, den Wechselrichter so auszulegen, daß er zur Speisung von Lampen gleicher Nennleistung unabhängig von ihrer Gasfüllung und damit von ihrer Brennspannung brauchbar ist.The invention is therefore based on the object of designing the inverter so that it can be used for supplying lamps of the same nominal power regardless of their gas filling and thus of their operating voltage.

Die erfindungsgemäße Lösung dieser Aufgabe ist in Anspruch 1 gekennzeichnet. Sie sorgt abhängig von der jeweils festgestellten Brennspannung für eine solche Betriebsfrequenz des Wechselrichters, daß sich in Verbindung mit der konstanten Induktivität der Drossel des Serienresonanzkreises gerade der erforderliche Lampenstrom einstellt. Der Steuersatz kann dabei einen Impulsgenerator haben, dessen Frequenz mit einer von der Brennspannung der Entladungslampe abgeleiteten Eingangsgröße ansteigt : Bei entsprechender Auslegung dieser Kennlinie stellt sich dann automatisch die zur jeweiligen Brennspannung passende Frequenz und damit der erforderliche Lampenstrom ein. Abweichend davon kann aber auch die Frequenz des Impulsgenerators umschaltbar sein, abhängig von dem Erreichen bzw. Unterschreiten bestimmter Schwellwerte der Brennspannung.The inventive solution to this problem is characterized in claim 1. Depending on the operating voltage determined in each case, it ensures such an operating frequency of the inverter that, in conjunction with the constant inductance of the choke of the series resonant circuit, the lamp current required is established. The headset can have a pulse generator, the frequency of which increases with an input variable derived from the operating voltage of the discharge lamp: if this characteristic curve is designed accordingly, the frequency matching the respective operating voltage and thus the required lamp current are automatically set. Deviating from this, however, the frequency of the pulse generator can also be switchable, depending on whether certain threshold values of the operating voltage have been reached or fallen below.

Meist ist vor Zündung der Entladungslampe eine ausreichende Vorheizung ihrer Elektroden erwünscht. Zu diesem Zweck ist es bekannt, die Betriebsfrequenz des Wechselrichters während einer Vorheizzeit soweit heraufzusetzen, daß sie einen genügenden Abstand von der Resonanzfrequenz des Resonanzkreises hat, so daß die Spannung an der Lampe zur Zündung nicht ausreicht. Nach Ablauf dieser Vorheizzeit wird dann die Betriebsfrequenz herabgesetzt und soweit an die Resonanzfrequenz angenähert, bis die Spannung an der Lampe zur Zündung ausreicht. Zur Verbindung dieser bekannten Funktionen mit der Erfindung ist gemäß einer vorteilhaften Weiterbildung vorgesehen, zwischen dem Steuersatz und dem Brennspannungsfühler einen Auswerter anzuordnen, dem zusätzlich die Spannung eines am Eingang des Wechselrichters liegenden Zeitgliedes zugeführt wird und der während einer mit der Einschaltung des Wechselrichters beginnenden Vorheizzeit und einer daran anschließenden Zündzeit die Frequenz der Steuerspannungen bestimmt und der danach das Ausgangssignal des Brennspannungsfühlers für den Steuersatz wirksam werden läßt. Während der durch das Zeitglied bestimmten Vorheiz- und Zündphase ist somit die Betriebsfrequenz des Wechselrichters nicht durch die Brennspannung sondern durch das Zeitglied bestimmt, das im einfachsten Fall ein RC-Glied oder eine monostabile Kippstufe ist.Sufficient preheating of its electrodes is usually desirable before the discharge lamp is ignited. For this purpose, it is known to increase the operating frequency of the inverter during a preheating time to such an extent that it is at a sufficient distance from the resonance frequency of the resonance circuit so that the voltage at the lamp is not sufficient for ignition. After this preheating time has elapsed, the operating frequency is reduced and approximated to the resonance frequency until the voltage at the lamp is sufficient for ignition. To connect these known functions with the invention, according to an advantageous further development, an evaluator is provided between the headset and the combustion voltage sensor, to which the voltage of a timing element located at the input of the inverter is also fed and which during a preheating time beginning when the inverter is switched on and a subsequent ignition time determines the frequency of the control voltages and which then allows the output signal of the internal voltage sensor for the headset to take effect. During the preheating and ignition phase determined by the timing element, the operating frequency of the inverter is therefore not determined by the operating voltage but by the timing element, which in the simplest case is an RC element or a monostable multivibrator.

Weitere vorteilhafte Ausgestaltungen der Erfindung sind in den Unteransprüchen gekennzeichnet.Further advantageous embodiments of the invention are characterized in the subclaims.

Die Erfindung wird anhand der Figuren näher erläutert ; es zeigen

  • Figur 1 ein Blockschaltbild der Erfindung,
  • Figur 2 die Abhängigkeit der Betriebsfrequenz des Wechselrichters während der einzelnen Betriebsphasen,
  • Figur 3 ein besonders einfaches Ausführungsbeispiel der Erfindung und
  • Figur 4 den Verlauf der Sägezahnspannung und die davon abhängige Verkürzung der Durchsteuerzeit - Erhöhung der Frequenz - abhängig von der Bezugsspannung.
The invention is illustrated by the figures; show it
  • FIG. 1 shows a block diagram of the invention,
  • FIG. 2 shows the dependency of the operating frequency of the inverter during the individual operating phases,
  • Figure 3 shows a particularly simple embodiment of the invention and
  • FIG. 4 shows the course of the sawtooth voltage and the shortening of the control time dependent on it - increase in frequency - depending on the reference voltage.

Bei dem Ausführungsbeispiel nach Fig. 1 sind die Schalter des Wechselrichters Feldeffektleistungstransistoren T1, T2, die in Reihe an der Gleichspannungsquelle Q liegen. Der Lastkreis mit einem Umschwingkondensator C10, der Entladungslampe LP und dem Serienresonanzkreis mit dem Kondensator C9 und der Drossel L1 liegt parallel zu T1, wobei der Kondensator C9 des Serienresonanzkreises zwischen den heizbaren Elektroden der Lampe liegt. Bei durchgesteuertem T2 liegt der Lastkreis an der Quelle Q und lädt sich in der folgenden Halbwelle über den durchgesteuerten T1 um.In the exemplary embodiment according to FIG. 1, the switches of the inverter are field-effect power transistors T1, T2, which are connected in series to the DC voltage source Q. The load circuit with a reversing capacitor C10, the discharge lamp LP and the series resonance circuit with the capacitor C9 and the choke L1 is parallel to T1, the capacitor C9 of the series resonance circuit being located between the heatable electrodes of the lamp. When T2 is activated, the load circuit is located at source Q and reloads in the following half-wave via activated T1.

Zur abwechselnden Durchsteuerung der Transistoren T1 und T2 dient ein Steuersatz S mit einem Impulsgenerator g, der Steuerimpulse mit einstellbarer Frequenz erzeugt ; diese werden über einen Impulsverteiler i abwechselnd auf die beiden Transistoren gegeben.A control set S with a pulse generator g, which generates control pulses with an adjustable frequency, is used to alternately control the transistors T1 and T2; these are alternately applied to the two transistors via a pulse distributor i.

Die Frequenz der Impulse des Generators g ist abhängig von dem Ausgangssignal des Auswerters A, der eingangsseitig an einen Brennspannungsfühler B und an ein Zeitglied Z mit zwei Ausgängen z1, z2 angeschlossen ist, und der im wesentlichen die Funktion hat, je nach Betriebsphase einen dieser Eingänge auf den Impulsgenerator g durchzuschalten.The frequency of the pulses of the generator g is dependent on the output signal of the evaluator A, which is connected on the input side to a combustion voltage sensor B and to a timing element Z with two outputs z1, z2, and which essentially has the function of one of these inputs, depending on the operating phase to connect to the pulse generator g.

Nach dem Einschalten der Gleichspannungsquelle Q macht der Auswerter A den ersten Ausgang z1 des Zeitgliedes Z wirksam, dessen Spannung zwischen den Zeitpunkt to und t1 (Vorheizzeit) für die maximale Betriebsfrequenz fv sorgt : Siehe Fig. 2. Bei dieser Frequenz kann die Lampe nicht zünden. Im Zeitpunkt t1 schaltet dann der Auswerter A - gesteuert durch Z - auf den zweiten Ausgang z2 des Zeitgliedes um, der eine kleinere oder eine bis zum Zeitpunkt t2 stetig sinkende Spannung liefert : Die damit verbundene Absenkung der Betriebsfrequenz führt in die Nähe der Resonanzfrequenz des Resonanzkreises und damit zur Zündung der Lampe. Nach dem Zeitpunkt t2 schließlich schaltet der Auswerter auf den Brennspannungsfühler B um, so daß dann ein von der Brennspannung der Lampe abhängiges Ausgangssignal am Impulsgenerator liegt, das bei einer Argon-Lampe eine höhere Betriebsfrequenz fBA und bei einer Krypton-Lampe eine niedrigere Betriebsfrequenz FBK (punktierte Linie in Fig. 2) zur Folge hat.After switching on the DC voltage source Q, the evaluator A activates the first output z1 of the timing element Z, the voltage of which ensures the maximum operating frequency f v between the times t o and t 1 (preheating time): see FIG. 2. At this frequency, the Do not ignite the lamp. At time t 1 , the evaluator A - controlled by Z - switches on the second output z2 of the timing element, which supplies a smaller or a continuously decreasing voltage up to time t 2 : the associated lowering of the operating frequency leads to the vicinity of the resonance frequency of the resonance circuit and thus to the ignition of the lamp. After the time t 2 , the evaluator finally switches over to the operating voltage sensor B, so that an output signal dependent on the operating voltage of the lamp is then present at the pulse generator, which has an operating frequency f BA higher for an argon lamp and a lower operating frequency for a krypton lamp F BK (dotted line in Fig. 2) results.

Bei dem detaillierten Ausführungsbeispiel nach Fig. 3 dient als Steuersatz im wesentlichen ein Sättigungstransformator L2, dessen Primärwicklung L21 in dem Lastkreis angeordnet ist und dessen Sekundärwicklungen L22, L23 an die Steuerstrecken der Transistoren T1, T2 angeschlossen sind. Der Lastkreis und der Sättigungstransformator sind so bemessen, daß sich ohne die nachfolgend beschriebenen erfindungsgemäßen Eingriffe die niedrigste in Betracht kommende Betriebsfrequenz, also die zur Speisung einer Krypton-Lampe, einstellt.In the detailed exemplary embodiment according to FIG. 3, a saturation transformer L2 is essentially used as the control set, the primary winding L21 of which is arranged in the load circuit and the secondary windings L22, L23 of which are connected to the control paths of the transistors T1, T2. The load circuit and the saturation transformer are dimensioned in such a way that the lowest possible operating frequency, that is to say that for supplying a krypton lamp, is obtained without the interventions according to the invention described below.

Bei Betrieb dieses Wechselrichters mit einer Argon-Lampe (mit einer höheren Brennspannung) wird die Durchsteuerzeit von T2 mit Hilfe von Sperrtransistoren T3, T4 verkürzt, durch die die Steuerstrecke von T2 kurzgeschlossen wird. Hierzu liegt der Emitter von T4 an einem Synchronisierkondensator C3, der über einen Vorwiderstand R2 an die Gleichspannungsquelle Q und über einen Vorschaltkondensator C2 an die Schaltstrecke des vorzeitig zu sperrenden Transistors T2 angeschlossen ist. Der Verlauf der Sägezahnspannung Des am Emitter des Sperrtransistors T4 ist in Fig. 4 dargestellt.When this inverter is operated with an argon lamp (with a higher operating voltage), the turn-through time of T2 is shortened with the aid of blocking transistors T3, T4, by which the control path of T2 is short-circuited. For this purpose, the emitter of T4 is connected to a synchronizing capacitor C3, which is connected via a series resistor R2 to the DC voltage source Q and via a series capacitor C2 to the switching path of the transistor T2 to be blocked prematurely. The course of the sawtooth voltage Des at the emitter of the blocking transistor T4 is shown in FIG. 4.

Die Basis von T4 liegt an einem Spannungsteiler mit den Widerständen R3, R4 und einem Vergleichskondensator C4, an dem eine in ihrer Größe steuerbare Bezugsspannung auftritt. Hierzu ist C4 einem Steuertransistor T5 parallel geschaltet und ferner über R5 an eine Hilfsspannungsquelle angeschlossen, die von einem Kondensator C5 gebildet wird, der über eine Ladediode D4 dem Synchronisierkondensator C3 parallel liegt.The basis of T4 lies on a voltage divider with the resistors R3, R4 and a comparison capacitor C4, on which a reference voltage which can be controlled in size occurs. For this purpose, C4 is connected in parallel to a control transistor T5 and is also connected via R5 to an auxiliary voltage source which is formed by a capacitor C5 which is connected in parallel with the synchronization capacitor C3 via a charging diode D4.

Sperrtransistor T4 wird immer durchgesteuert, wenn die Sägezahnspannung Uc3 größer als die Bezugsspannung Uc4 wird (Fig. 2) :

  • Bis zum Zeitpunkt t2 ist Transistor T2 gesperrt und T1 leitend ; letzterer wird zu diesem Zeitpunkt durch den Sättigungstransformator L2 zugesteuert. Die Induktivitäten des Lastkreises treiben dann den Laststrom in gleicher Richtung weiter über die Gleichspannungsquelle Q und die beiden Kondensatoren C3 und C2, die damit umgeladen werden. Dabei wird der Emitter von T4 negativer als seine Basis und dieser Transistor und damit T3 gesperrt : Infolge dessen kann der von L23 gelieferte Steuerimpuls an T2 wirksam werden und diesen Transistor im Zeitpunkt t4 durchsteuern. Von da an werden die Kondensatoren C2 und C3 in Parallelschaltung (C2 über T2) über den Vorwiderstand R2 aufgeladen bis die Sägezahnspannung am Emitter von T4 größer als die Bezugsspannung an dessen Basis geworden ist. Dann werden T3 und T4 durchgesteuert und Transistor T2 vorzeitig gesperrt.
Blocking transistor T4 is always turned on when the sawtooth voltage U c3 becomes greater than the reference voltage U c4 (FIG. 2):
  • Until time t 2 , transistor T2 is blocked and T1 is conducting; the latter is controlled by the saturation transformer L2 at this time. The inductances of the load circuit then drive the load current in the same direction on via the DC voltage source Q and the two capacitors C3 and C2, which are thus reloaded. The emitter of T4 is more negative than its base and this transistor and thus T3 is blocked: As a result, the control pulse supplied by L23 can take effect at T2 and control this transistor at time t 4 . From then on, the capacitors C2 and C3 are charged in parallel (C2 via T2) via the series resistor R2 until the sawtooth voltage at the emitter of T4 has become greater than the reference voltage at its base. Then T3 and T4 are turned on and transistor T2 is blocked prematurely.

Der Strom des Lastkreises fließt jedoch noch in gleicher Richtung weiter und zwar über die Rückstromdiode des Transistors T1 (bei MOS-FET-Transistoren integriert). Beim Nulldurchgang des Laststromes erzeugt dann der Sättigungstransformator an L22 eine Durchsteuerspannung für T1, über den dann der Entladestrom des Umschwingkondensators C10 in entgegengesetzter Richtung durch den Lastkreis fließen kann. Die Zeitdauer dieser Halbwelle ist dabei von dem Sättigungstransformator abhängig und im wesentlichen konstant ; die Änderung der resultierenden Betriebsfrequenz wird in diesem Fall also nur durch Verkürzen der einen durch T2 bestimmten Halbwelle erreicht.However, the current of the load circuit continues to flow in the same direction, namely via the reverse current diode of transistor T1 (integrated in MOS-FET transistors). When the load current crosses zero, the saturation transformer at L22 then generates a control voltage for T1, via which the discharge current of the reversing capacitor C10 can flow through the load circuit in the opposite direction. The duration of this half-wave depends on the saturation transformer and is essentially constant; in this case, the change in the resulting operating frequency is only achieved by shortening the one half-wave determined by T2.

Für diese Verkürzung ist die Größe der Bezugsspannung Uc4 an dem Vergleichskondensator C4 durch den parallel geschalteten Steuertransistor T5 unterschiedlich einstellbar. Die Steuerstrecke dieses Transistors liegt hierzu einerseits an einem Spannungsteiler mit den Widerständen R6 und R7 und über eine Zenerdiode D8 und einen Spannungsteiler R8. R9 an einem Kondensator C8, der sich über D9 bzw. D10, C7 und R12 auf einen von der Lampenspannung abhängigen Wert auflädt.For this shortening, the magnitude of the reference voltage U c4 at the comparison capacitor C4 can be set differently by the control transistor T5 connected in parallel. The control path of this transistor is connected to a voltage divider with resistors R6 and R7 and a Zener diode D8 and a voltage divider R8. R9 on a capacitor C8, which charges via D9 or D10, C7 and R12 to a value dependent on the lamp voltage.

R6 und R7 liegen parallel zu einem Kondensator C6, der über eine Entkopplungsdiode D6 an ein RC-Glied R1, C1 angeschlossen ist und mit diesem über Widerstände R11, R10 und eine Elektrode der Lampe LP von der Gleichspannungsquelle Q aufgeladen wird. Die Spannung an C1 steuert dabei über die Schaltdiode D2 den Transistor T2 beim Einschalten des Wechselrichters an, verschwindet dann aber, da C1 über D1 und T2 entladen wird. Daher ist die Entkopplungsdiode D6 nach dem Anschwingen des Wechselrichters gesperrt : Der Steuertransistor T5 ist daher nach dem Anschwingen des Wechselrichters nur während einer kurzen Entladezeit des Kondensators C6 durchgesteuert, der mit den Widerständen R6 und R7 die Vorheizzeit bestimmt. Solange T5 auf diesem Wege durchgesteuert ist wird der Sperrtransistor T4 infolge der niedrigen Bezugsspannung Uc4v in jeder Periode bereits im Zeitpunkt tS1 (Fig. 4) leitend und damit die über T2 fließende Halbwelle verkürzt. Die daraus resultierende Erhöhung der Betriebsfrequenz hat eine entsprechend niedrige Spannung am Kondensator C4 parallel zu der Entladungslampe LP zur Folge, so daß diese nicht zünden kann.R6 and R7 are connected in parallel to a capacitor C6, which is connected via a decoupling diode D6 to an RC element R1, C1 and is charged by the DC voltage source Q via resistors R11, R10 and an electrode of the lamp LP. The voltage at C1 controls the transistor T2 via the switching diode D2 when the inverter is switched on, but then disappears because C1 is discharged via D1 and T2. Therefore, the decoupling diode D6 is blocked after the inverter starts to oscillate: The control transistor T5 is therefore only activated after the inverter has started to oscillate during a short discharge time of the capacitor C6, which determines the preheating time with the resistors R6 and R7. As long as T5 is controlled in this way, the blocking transistor T4 becomes conductive in each period due to the low reference voltage U c4v in time t S1 (FIG. 4) and thus the half-wave flowing via T2 is shortened. The resulting increase in the operating frequency results in a correspondingly low voltage across the capacitor C4 in parallel with the discharge lamp LP, so that it cannot ignite.

Im Zeitpunkt t1 in Fig. 2 wird T5 zunehmend zugesteuert und damit steigt die Bezugsspannung an C4, so daß die über T2 fließenden Halbwellen stetig länger werden, bis sich die resultierende Betriebsfrequenz soweit der Resonanzfrequenz des Lastkreises genähert hat, daß die Lampe zündet. T3, T4 und T5 sind dann gesperrt und es stellt sich eine Betriebsfrequenz mit gleich langen Halbwellen ein, die ausschließlich durch den Sättigungstransformator bestimmt ist und für den Betrieb einer Krypton-Lampe bemessen ist. In diesem Fall ist die an dem Kondensator C8 auftretende Spannung zu klein, um über die Zenerdiode D8 den Steuertransistor T5 wieder leitend steuern zu können.At time t 1 in Fig. 2, T5 is increasingly controlled and thus the reference voltage at C4 increases, so that the half-waves flowing over T2 become longer and longer until the resulting operating frequency has approached the resonance frequency of the load circuit to such an extent that the lamp ignites. T3, T4 and T5 are then blocked and the operating frequency is the same long half-waves, which is determined exclusively by the saturation transformer and is dimensioned for the operation of a Krypton lamp. In this case, the voltage occurring at the capacitor C8 is too low to be able to again control the control transistor T5 via the zener diode D8.

Wird dagegen an dem so bemessenen Wechselrichter eine Lampe gleicher Nennleistung mit Argon-Füllung angeschlossen, so stellt sich - nach gleichartigem Ablauf der Vorheiz- und Zündphase - im Brennbetrieb eine höhere Brennspannung ein, die den Steuertransistor T5 über Zenerdiode D8 definiert aufsteuert, so daß sich an dem Vergleichskondensator C4 (infolge der Gegenkopplungsdiode D7) eine für diesen Betriebsfall charakteristische Bezugsspannung Uc4A einstellt, bei der die Sperrtransistoren T3, T4 bereits im Zeitpunkt t62 (Fig. 4) durchgesteuert werden, was zu einer entsprechenden Verkürzung der über T2 fließenden Halbwelle und damit zu einer Erhöhung der resultierenden Betriebsfrequenz führt. Diese Erhöhung ist so bemessen, daß die Lampe mit Argon-Füllung gerade den richtigen Betriebsstrom erhält.If, on the other hand, a lamp of the same nominal power with argon filling is connected to the inverter rated in this way, after the preheating and ignition phase has proceeded in the same way, a higher operating voltage is set in the burner mode, which turns on the control transistor T5 via zener diode D8, so that on the comparison capacitor C4 (as a result of the negative feedback diode D7) sets a reference voltage U c4A which is characteristic of this operating case and at which the blocking transistors T3, T4 are already activated at time t 62 (FIG. 4), which leads to a corresponding shortening of the half-wave flowing via T2 and thus leads to an increase in the resulting operating frequency. This increase is such that the lamp with argon filling receives the correct operating current.

Die Spannung an dem Kondensator C8 wird zugleich zur Abschaltung des Wechselrichters bei dauernd zündunwilliger Lampe herangezogen : Erreicht diese Spannung einen durch die Überwachungseinrichtung Ü vorgegebenen Grenzwert, dann spricht diese an und schließt den Anschwingkondensator C1 und die Steuerelektrode von T1 über D5 kurz. Die Ansprechschwelle dieser Überwachungseinrichtung liegt über der Spannung, die sich an C8 bei Betrieb des Wechselrichters mit einer Argon-Lampe einstellt. Ferner ist die Ladezeitkonstante von C8 so groß, daß die Überwachungseinrichtung auch nicht während der Vorheiz- und der Zündphase ansprechen kann, es sei denn, die Lampe würde in der vorgegebenen Zeit (t1/t2) nicht zünden.The voltage at the capacitor C8 is also used to switch off the inverter when the lamp is permanently unable to ignite: If this voltage reaches a limit value specified by the monitoring device Ü, then this responds and short-circuits the starting capacitor C1 and the control electrode from T1 to D5. The response threshold of this monitoring device is above the voltage that arises at C8 when the inverter is operated with an argon lamp. Furthermore, the charging time constant of C8 is so great that the monitoring device cannot respond during the preheating and ignition phases, unless the lamp would not ignite in the specified time (t 1 / t 2 ).

Claims (6)

1. An inverter comprising two switches (T1, T2) which can be alternately rendered conductive, and comprising a load circuit arranged parallel to the first switch (T1) and connected via the second switch (T2) to a d. c. voltage source (Q), and consisting of the series arrangement of a reversible oscillatory capacitor (C10), a series resonant circuit composed of a choke (L1) and a capacitor (C9), and a discharge lamp (LP) equipped with heatable electrodes, where these electrodes are arranged in the load circuit and are connected to one another via the capacitor (C9) of the series resonant circuit, and comprising a control set (S) which supplies control voltages which alternately render the switches conductive, characterised by a burning voltage sensor (B) which monitors the burning voltage of the lamp (LP) and supplies an output signal dependent upon the burning voltage and fed to the control set (S) to determine the frequency of the control voltages which it supplies.
2. An inverter as claimed in Claim 1, characterised by an analyser (A) arranged between the control set (S) and the burning voltage sensor (B) and additionally supplied with the voltage of a timer (Z) connected to the input (PN) of the inverter to determine the frequency of the control voltages during a preliminary heating period (tD/tl) which commences when the d. c. voltage source is switched on and during a following ignition time (t1/t2), and which subsequently actuates the output signal of the burning voltage sensor (B) for the control set (S).
3. An inverter as claimed in Claim 2, characterised in that the analyser (A) compares a saw-tooth voltage (VC3) with a reference voltage (Vc4) and controls a blocking transistor (T3, T4) in dependence thereon to terminate the control voltage of one of the switches (T2) of the inverter, where the saw-tooth voltage is tapped from a synchronising capacitor (C3) connected via a series resistor (R2) to the d. c. voltage source (Q) and connected in parallel with the switch (T2) which is to be blocked via a series capacitor (C2).
4. An inverter as claimed in Claim 3, characterised in that the reference voltage (Vc4) is tapped from a comparator capacitor (C4) arranged in parallel with a control transistor (T5) whose control path is connected via a Zener diode (D8) to a voltage source (R8, R9, C8) which supplies a voltage proportional to the voltage connected to the lamp (LP).
5. An inverter as claimed in Claim 4, characterised in that the control path of the control transistor (T5) is connected to an RC-component (R6, R7, C6) charged via a decoupling diode (D6) which is conductive only between the switching on of the inverter and its start-up.
6. An inverter as claimed in Claim 5, with a monitoring device which disconnects the inverter when the lamp is permanently unwilling to ignite, characterised in that the monitoring device (0) is likewise connected at its input to the voltage source (C8) which represents an RC-component (R8, R9, C8) whose time constant is such that the voltage connected to the capacitor (C8) does not reach the response threshold of the monitoring device if the lamp ignites within a predetermined length of time.
EP84105786A 1983-05-27 1984-05-21 Inverter for feeding discharge lamps Expired EP0127101B1 (en)

Priority Applications (1)

Application Number Priority Date Filing Date Title
AT84105786T ATE25800T1 (en) 1983-05-27 1984-05-21 INVERTER FOR POWERING DISCHARGE LAMPS.

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
DE19833319352 DE3319352A1 (en) 1983-05-27 1983-05-27 INVERTER FOR POWERING DISCHARGE LAMPS
DE3319352 1983-05-27

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EP0127101A1 EP0127101A1 (en) 1984-12-05
EP0127101B1 true EP0127101B1 (en) 1987-03-04

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EP (1) EP0127101B1 (en)
AT (1) ATE25800T1 (en)
DE (2) DE3319352A1 (en)
FI (1) FI76906C (en)

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GB2180418A (en) * 1985-09-14 1987-03-25 Contrology Limited Fluorescent lamp supply circuit
EP0241279A1 (en) * 1986-04-08 1987-10-14 Actronic Lighting Cc Controller for gas discharge lamps
EP0359860A1 (en) * 1988-09-23 1990-03-28 Siemens Aktiengesellschaft Device and method for operating at least one discharge lamp
GB2226463A (en) * 1988-12-21 1990-06-27 Sirous Yazdanian Control of fluorescent lights
EP0396621A1 (en) * 1988-01-19 1990-11-14 Etta Industries, Inc. Fluorescent dimming ballast utilizing a resonant sine wave power converter
EP0413991A1 (en) * 1989-07-28 1991-02-27 Toshiba Lighting & Technology Corporation Discharge lamp lighting apparatus for driving discharge lamp according to rating thereof
EP0422255A1 (en) * 1989-10-09 1991-04-17 Siemens Aktiengesellschaft Electronic ballast
WO1991007863A1 (en) * 1989-11-18 1991-05-30 Ac/Dc Lighting Limited Inverters and cathode lamp arrangements
EP0435228A2 (en) * 1989-12-29 1991-07-03 Zumtobel Aktiengesellschaft Circuit and process for operating (and igniting) a discharge lamp
US5798617A (en) 1996-12-18 1998-08-25 Pacific Scientific Company Magnetic feedback ballast circuit for fluorescent lamp
US5866993A (en) 1996-11-14 1999-02-02 Pacific Scientific Company Three-way dimming ballast circuit with passive power factor correction
US5925986A (en) 1996-05-09 1999-07-20 Pacific Scientific Company Method and apparatus for controlling power delivered to a fluorescent lamp
US5955841A (en) 1994-09-30 1999-09-21 Pacific Scientific Company Ballast circuit for fluorescent lamp
US5982111A (en) 1994-09-30 1999-11-09 Pacific Scientific Company Fluorescent lamp ballast having a resonant output stage using a split resonating inductor
US6037722A (en) 1994-09-30 2000-03-14 Pacific Scientific Dimmable ballast apparatus and method for controlling power delivered to a fluorescent lamp
WO2000024233A2 (en) * 1998-10-16 2000-04-27 Electro-Mag International, Inc. Ballast circuit
US6169375B1 (en) 1998-10-16 2001-01-02 Electro-Mag International, Inc. Lamp adaptable ballast circuit
WO2001097574A2 (en) * 2000-06-14 2001-12-20 Brenex Electrics Pty. Limited Control circuits for fluorescent tubes
DE102008031409A1 (en) * 2008-07-02 2010-01-07 Tridonicatco Gmbh & Co. Kg Detection of the type of a gas discharge lamp connected to an operating device
DE102008047440A1 (en) * 2008-09-16 2010-03-25 Tridonicatco Gmbh & Co. Kg Determination of the type of bulb or the topology of several bulbs

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DE3521496A1 (en) * 1985-04-22 1986-10-23 Sorbios Verfahrenstech Method and device for medium-frequency and high-frequency high-voltage supply for impedance-like loads, especially in the case of gas discharges
DE3628989A1 (en) * 1986-08-26 1988-03-03 Ceag Licht & Strom ELECTRONIC BALLAST
US6160361A (en) * 1998-07-29 2000-12-12 Philips Electronics North America Corporation For improvements in a lamp type recognition scheme
US6677719B2 (en) * 2002-06-03 2004-01-13 Stmicroelectronics, Inc. Ballast circuit
US20050168171A1 (en) 2004-01-29 2005-08-04 Poehlman Thomas M. Method for controlling striations in a lamp powered by an electronic ballast

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US4060752A (en) * 1976-03-01 1977-11-29 General Electric Company Discharge lamp auxiliary circuit with dI/dt switching control
DE2924331A1 (en) * 1979-06-15 1980-12-18 Siemens Ag Supply and starting circuit for discharge lamp - has function generator producing rising control signal for current in lamp with arc discharge and incandescent cathode
DE2928490A1 (en) * 1979-07-14 1981-01-29 Frei Hans Joachim Solar lamp constant control circuit - has series resonant start and current control with feedback thermistor to pulse width modulation power supply
GB2095930A (en) * 1981-03-27 1982-10-06 Stevens Carlile R Constant power ballast

Cited By (25)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
GB2180418A (en) * 1985-09-14 1987-03-25 Contrology Limited Fluorescent lamp supply circuit
EP0241279A1 (en) * 1986-04-08 1987-10-14 Actronic Lighting Cc Controller for gas discharge lamps
EP0396621A1 (en) * 1988-01-19 1990-11-14 Etta Industries, Inc. Fluorescent dimming ballast utilizing a resonant sine wave power converter
EP0396621A4 (en) * 1988-01-19 1992-01-15 Etta Industries, Inc. Fluorescent dimming ballast utilizing a resonant sine wave power converter
EP0359860A1 (en) * 1988-09-23 1990-03-28 Siemens Aktiengesellschaft Device and method for operating at least one discharge lamp
US5049790A (en) * 1988-09-23 1991-09-17 Siemens Aktiengesellschaft Method and apparatus for operating at least one gas discharge lamp
GB2226463A (en) * 1988-12-21 1990-06-27 Sirous Yazdanian Control of fluorescent lights
US5039921A (en) * 1989-07-28 1991-08-13 Toshiba Lighting And Technology Corporation Discharge lamp lighting apparatus for driving discharge lamp according to rating thereof
EP0413991A1 (en) * 1989-07-28 1991-02-27 Toshiba Lighting & Technology Corporation Discharge lamp lighting apparatus for driving discharge lamp according to rating thereof
EP0422255A1 (en) * 1989-10-09 1991-04-17 Siemens Aktiengesellschaft Electronic ballast
US5066894A (en) * 1989-10-09 1991-11-19 Siemens Aktiengesellschaft Electronic ballast
WO1991007863A1 (en) * 1989-11-18 1991-05-30 Ac/Dc Lighting Limited Inverters and cathode lamp arrangements
EP0435228A2 (en) * 1989-12-29 1991-07-03 Zumtobel Aktiengesellschaft Circuit and process for operating (and igniting) a discharge lamp
EP0435228A3 (en) * 1989-12-29 1992-07-08 Zumtobel Aktiengesellschaft Circuit and process for operating (and igniting) a discharge lamp
US6037722A (en) 1994-09-30 2000-03-14 Pacific Scientific Dimmable ballast apparatus and method for controlling power delivered to a fluorescent lamp
US5955841A (en) 1994-09-30 1999-09-21 Pacific Scientific Company Ballast circuit for fluorescent lamp
US5982111A (en) 1994-09-30 1999-11-09 Pacific Scientific Company Fluorescent lamp ballast having a resonant output stage using a split resonating inductor
US5925986A (en) 1996-05-09 1999-07-20 Pacific Scientific Company Method and apparatus for controlling power delivered to a fluorescent lamp
US5866993A (en) 1996-11-14 1999-02-02 Pacific Scientific Company Three-way dimming ballast circuit with passive power factor correction
US5798617A (en) 1996-12-18 1998-08-25 Pacific Scientific Company Magnetic feedback ballast circuit for fluorescent lamp
WO2000024233A2 (en) * 1998-10-16 2000-04-27 Electro-Mag International, Inc. Ballast circuit
US6169375B1 (en) 1998-10-16 2001-01-02 Electro-Mag International, Inc. Lamp adaptable ballast circuit
WO2001097574A2 (en) * 2000-06-14 2001-12-20 Brenex Electrics Pty. Limited Control circuits for fluorescent tubes
DE102008031409A1 (en) * 2008-07-02 2010-01-07 Tridonicatco Gmbh & Co. Kg Detection of the type of a gas discharge lamp connected to an operating device
DE102008047440A1 (en) * 2008-09-16 2010-03-25 Tridonicatco Gmbh & Co. Kg Determination of the type of bulb or the topology of several bulbs

Also Published As

Publication number Publication date
DE3319352A1 (en) 1984-11-29
FI76906C (en) 1988-12-12
FI76906B (en) 1988-08-31
ATE25800T1 (en) 1987-03-15
EP0127101A1 (en) 1984-12-05
FI841794A (en) 1984-11-28
DE3462574D1 (en) 1987-04-09
FI841794A0 (en) 1984-05-04

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