EP0451909A2 - Predistortion linearizer for microwave power amplifiers - Google Patents
Predistortion linearizer for microwave power amplifiers Download PDFInfo
- Publication number
- EP0451909A2 EP0451909A2 EP91200799A EP91200799A EP0451909A2 EP 0451909 A2 EP0451909 A2 EP 0451909A2 EP 91200799 A EP91200799 A EP 91200799A EP 91200799 A EP91200799 A EP 91200799A EP 0451909 A2 EP0451909 A2 EP 0451909A2
- Authority
- EP
- European Patent Office
- Prior art keywords
- transistor
- linearizer
- amplifier
- output
- phase
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
Images
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
- H03F1/3276—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits using the nonlinearity inherent to components, e.g. a diode
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
Definitions
- the present invention relates to improvements in microwave power amplifiers.
- Quadrature Amplitude Modulation impose very stringent linearity requirements for the radiofrequency power amplifier of the transmitter on which depends not a little the degradation of the modulated signal.
- the power output from the final amplifier devices must be considerably lower than their saturation power so that the nonlinear distortions thereof introduced satisfy the specifications of the transmitter. Said distortions are due to compression of gain at high power found in the trend of the Amplitude Modulation (AM/AM) distortion curve at high power and the amplitude/modulation phase modulation (AM/PM) conversion curve, again at high power.
- AM/AM Amplitude Modulation
- AM/PM amplitude/modulation phase modulation
- linearizing network in the transmitting section permits use of power devices with lower saturation for given distortion produced with a resulting increase of efficiency, e.g. for application in the transmitters of on-board repeaters in satellite communication systems or, for a given saturation power of final devices, allows higher linearity of the amplifier, e.g. for applications in transmitters for earth stations in said satellite communication systems.
- a linearization technique presently well known is termed 'feed forward error control' and includes all the linearizers which use an auxiliary microwave amplifier which amplifies an error signal obtained by determining the difference between the input signal and the distorted one appropriately attenuated output from the main amplifier.
- the error signal is proportional to the distortions generated by the main amplifier so that, again added with appropriate phase and amplitude at the output of the main amplifier, it reduces the distortions affecting the output signal.
- RF radio frequency
- predistorters i.e. nonlinear networks inserted upstream of the final microwave amplifier, which distort the input signal by means of networks embodied with components which work in nonlinear state in order to compensate for the AM/AM distortion curve and the amplitude/phase conversion curve AM/PM of the final power amplifier and guaranteeing better linearity of the transmitting section.
- the main drawback of said known predistorters consists however of the excessive complexity of said networks and thus of their still excessive cost.
- a predistortion linearizer applicable upstream of the power amplifier characterized in that it comprises a main network including a phase modulator and an amplitude modulator arranged in cascade and a secondary network with base band frequency including means for detecting in amplitude and filtering part of the input signal so as to produce a detected signal being a function of the instantaneous input power and a pair of adjustable gain amplifiers supplied with said detected signal and acting on said modulators in such a manner as to give them nonlinear response curves such as to compensate independently for both amplitude and phase nonlinearity of the power amplifier.
- Said linearizer although it has all the advantages compared with the known art listed in the application, is still of costly and cumbersome embodiment, principally because of the presence therein of derived branches.
- the object of the present invention is to overcome the above drawbacks and indicate a predistortion linearizer for microwave power amplifiers extremely simplified in its circuital structure. Indeed it comprises essentially a single transistor which carries out the functions both of the gain expander amplifier for recovery of the amplitude distortion of the power amplifier and generator of a command signal for a phase shift element for recovery of the phase distortion of the power amplifier.
- said single transistor is placed upstream of the final power amplifier and followed by the phase shift element.
- its function is fulfilled by the same final power amplifier.
- the object of the present invention is a predistortion linearizer for microwave power amplifiers as described in claim 1.
- FIG. 1 shows a possible form of circuit embodiment of the linearizer which is the object of the invention and is applied upstream of a power amplifier for the purpose of compensating for the amplitude and phase distortions.
- the amplitude and phase response curves of a microwave power amplifier are typically as shown in FIG. 2 wherein is shown the qualitative shape of the output power Pu (AM/AM distortion curve) and the input-output phase variation VF (AM/PM conversion curve) as a function of input power Pi.
- the linearizer has thus a dual purpose, i.e. first to expand its own gain at the Pi values so that in FIG.
- the knee of the curve Pu occurs to compensate for said knee and continue the linear shape of Pu even in the immediate proximity of saturation, this latter zone being indicated by a horizontal shape which however cannot be offset;
- the second purpose is to vary its own input-output phase in a manner opposite to the shape of VF in FIG. 2.
- RFin indicate the input and output connectors respectively of a radiofrequency signal.
- C1, C2 ...C5 indicate capacitances and L1, L2 inductances of known type.
- FT1 indicates a GaAsFET transistor (with gallium arsenide field effect) equipped as known with three connectors, source S, drain D and gate G1, used in a common source configuration.
- R1 indicates a polarization resistance of FT1.
- D1 indicates a varactor diode.
- RA1, RA2, RA3 indicate common impedance matching networks for input, interstage and output respectively embodied for example in microstrip.
- VG, VP and V+ indicate the supply voltages of the gate of FT1, the anode of D1 and fixed positive respectively.
- C1 and C2 are used as block capacitances of the continuous component at the input RFin and the output RFout respectively.
- C4 and the networks C3-L2, C5-L1 form low-pass decoupling filters between signal and supplies VG, VP and V+;
- C3, C4 and C5 have a grounded end.
- the input signal is applied to the gate of FT1 through C1 and RA1; the voltage Vg is also applied to the gate of FT1 through the filter C3-L2.
- the source of FT1 is connected to ground while the voltage V+ is brought to the drain through the filter C5-L1 and the resistance R1.
- the cathode of the diode D1 is connected also continuously to the drain of FT1 through the network RA2 and to the output RFout through the network RA3 and C2.
- the transistor FT1 amplifies the RF signal applied at the input RFin; the values of fixed supply voltage V+, resistance R1 and voltage VG are chosen in such a manner as to keep FT1 in underpolarization conditions, i.e. with the working point near the pinch-off region (low values of drain-source current Ids and gate-source voltage Vgs).
- an increase in the power of the RF input signal is capable of changing the working point of the device, i.e. increasing the continuous component of the current Ids. Since the gain of FT1 depends on said continuous component of Ids, the increase of the latter causes an increase in gain and supplies the desired expansion effect.
- FIG. 3 shows the qualitative shape of the gain curve Gi obtained as a function of the power Pi of the signal RF at the input RFin.
- the parameters of the polarization network of FT1, i.e. VG, R1 and V+ it is possible to obtain a shape of Gi optimized from the point of view of compensation of the AM/AM distortion curve.
- the figures also show as a function of Pi the corresponding qualitative shape of the continuous component of Ids.
- a detected voltage of the modulating signal proportional to the continuous component of Ids is localized and used as a control signal of the phase variation introduced by the downstream device which in the nonlimiting example described is the varactor diode D1.
- the effect obtained is thus a modulation of said voltage FL which, as is known, influences the internal capacitance value of the varactor and hence the phase shift introduced therefrom onto the RF signal.
- This effect can be used with advantage to offset the conversion curve AM/PM.
- FIG. 4 shows the qualitative shape of the phase variation FL as a function of the power Pi of the input signal.
- the shape of FL can be made specular in relation to that of VF (shown with broken line in FIG. 4) by appropriate sizing of the parameters which influence it, i.e. R1 and VP.
- the transistor FT1 fulfils a dual function: that of gain expander to offset the AM/AM distortion curve and that of detector of a modulation voltage which directly pilots a phase shifter to offset the AM/PM conversion curve.
- the gain expansion characteristic is adjusted by varying VG and the phase characteristic is adjusted by varying VP.
- the device of FIG. 1 follows very rapidly the dynamics of the input signal RF, i.e. its response as gain expander and phase shifter is a wide band for the modulating signal because it consists of a single RF branch. This is in effect very important because it makes the device usable even when the multicarrier modulating signal is wide band, e.g. in transponders for satellites.
- the linearization function can be performed by said final power stage consisting for example of a transistor of the FT1 type of FIG.1 brought to a similar working condition.
- phase variations can be offset by a circuit of known type upstream piloted by the voltage Vds of the final transistor as described with reference to FIG. 1.
- the variant now described can be used to advantage in QAM multi-level modulation systems where the RF output power is not constant but varies considerably in relation to the average value depending on the point of the constellation of symbols to be transmitted; for example in the 64QAM system there is a difference of approximately 8dB between average and maximum RF output power.
- the current Ids stays around 0,7A and reaches 2A only for the higher levels for which Pu is high. Since the points of the constellation are all equally probable, the average Ids current will be approximately 1.1-1.2A, and not 2A, with a clear consumption savings. Decrease of gain in the linear zone from 10 to 7dB does not involve problems because the 3dB difference is necessary only at the power peaks.
- a second variant calls for the embodiment of the phase shifter as in FIG. 6 wherein the same symbols as in FIG. 1 indicate the same components interconnected in the same manner.
- FIG. 6 there has been added a second varicap diode D2 equal and in antiparallel to D1 for the signals.
- the anode of D2 is connected to the cathode of D1 while the cathode of D2 is polarized by the direct voltage VP2 and is connected to ground for the signals through the filter capacitance C6.
- FIG. 6 serves in case it desired to offset a phase variation which may be either leading or delaying.
- One diode offsets the delaying phase variations and the other the leading ones; by appropriately sizing the voltages VP and VP2 it is possible to make one of the two diodes work while excluding the other, thus obtaining a leading phase shift with D2 inserted or delaying with D1 inserted.
- a third variant calls for embodiment of the phase shifter as in FIG. 7 wherein the same symbols as in FIG. 1 indicate the same components interconnected in the same manner and performing the same functions.
- phase shifter is embodied by a GaAsFET transistor indicated by FT2 and polarized in such a manner as to function as a normal amplifier in a common source configuration with the continuous component of the drain-source current Ids2 ⁇ 1 ⁇ 2Idss.
- R2 indicates the load resistance of FT2.
- C7, C8 and C9 indicate capacitances and L3, L4 indicate inductances of known type.
- C9 and the networks C7-L3, C8-L4 form low-pass decoupling filters between signals and supplies for output RFout, gate FT2 and drain FT2;
- C7, C8 and C9 have one end grounded.
- RA4 indicates an output matching network with the same function as RA1, RA2 and RA3.
- the voltage detected in the modulation of the modulating signal (which is proportional to the continuous component of Ids of FT1 located at the ends of R1) is used as the control signal of the phase variation introduced by FT2.
- the control signal is used as the control signal of the phase variation introduced by FT2.
- AMP receives and amplifies the continuous component of the drain voltage of FT1 at one end of R1 and supplies it with changed sign to the gate of FT2 through the filter C7-L3 to obtain a modulation effect of the continuous component of the gate voltage of FT2. This effect is translated into a variation of the capacitance Cgs of FT2.
- AMP can be embodied by a reversing stage of any known type. If embodied by a GaAsFET transistor the entire circuit of FIG. 7 can be integrated in MMIC technology of simple and economical embodiment.
- the amplifier AMP will have the same characteristics as above except that it will not be reversing.
- the transistors called for in the various forms of embodiment described can also be of another type, e.g. bipolar.
Abstract
Description
- The present invention relates to improvements in microwave power amplifiers.
- Modulations presently in use in radio links, essentially Quadrature Amplitude Modulation (QAM), impose very stringent linearity requirements for the radiofrequency power amplifier of the transmitter on which depends not a little the degradation of the modulated signal.
- The power output from the final amplifier devices must be considerably lower than their saturation power so that the nonlinear distortions thereof introduced satisfy the specifications of the transmitter. Said distortions are due to compression of gain at high power found in the trend of the Amplitude Modulation (AM/AM) distortion curve at high power and the amplitude/modulation phase modulation (AM/PM) conversion curve, again at high power.
- Obviating these distortions usually involves oversizing said final amplifiers and hence high cost of the power amplifying section.
- As known in itself use of a linearizing network in the transmitting section permits use of power devices with lower saturation for given distortion produced with a resulting increase of efficiency, e.g. for application in the transmitters of on-board repeaters in satellite communication systems or, for a given saturation power of final devices, allows higher linearity of the amplifier, e.g. for applications in transmitters for earth stations in said satellite communication systems.
- A linearization technique presently well known is termed 'feed forward error control' and includes all the linearizers which use an auxiliary microwave amplifier which amplifies an error signal obtained by determining the difference between the input signal and the distorted one appropriately attenuated output from the main amplifier. The error signal is proportional to the distortions generated by the main amplifier so that, again added with appropriate phase and amplitude at the output of the main amplifier, it reduces the distortions affecting the output signal.
- It is clear that the merit figure or degree of quality of this linearization system depends almost exclusively on the balancing of the final adder or coupler, which subtracts the error signal from the output signal of the amplifier. A balancing regulation circuitry (in amplitude and phase) of said coupler is therefore necessary and is quite complex. In addition it is a true amplifier-linearizer complex in itself, not an addition to improve a known amplifier.
- Another present linearization technique calls for the use of RF (radio frequency) predistorters, i.e. nonlinear networks inserted upstream of the final microwave amplifier, which distort the input signal by means of networks embodied with components which work in nonlinear state in order to compensate for the AM/AM distortion curve and the amplitude/phase conversion curve AM/PM of the final power amplifier and guaranteeing better linearity of the transmitting section. The main drawback of said known predistorters consists however of the excessive complexity of said networks and thus of their still excessive cost.
- An example of predistortion linearizer for microwave power amplifiers is described in Italian patent application no. 19497-A/87 filed by the same applicant 26 February 1987.
- In said previous Italian patent application there is described a predistortion linearizer applicable upstream of the power amplifier characterized in that it comprises a main network including a phase modulator and an amplitude modulator arranged in cascade and a secondary network with base band frequency including means for detecting in amplitude and filtering part of the input signal so as to produce a detected signal being a function of the instantaneous input power and a pair of adjustable gain amplifiers supplied with said detected signal and acting on said modulators in such a manner as to give them nonlinear response curves such as to compensate independently for both amplitude and phase nonlinearity of the power amplifier.
- Said linearizer, although it has all the advantages compared with the known art listed in the application, is still of costly and cumbersome embodiment, principally because of the presence therein of derived branches.
- Accordingly the object of the present invention is to overcome the above drawbacks and indicate a predistortion linearizer for microwave power amplifiers extremely simplified in its circuital structure. Indeed it comprises essentially a single transistor which carries out the functions both of the gain expander amplifier for recovery of the amplitude distortion of the power amplifier and generator of a command signal for a phase shift element for recovery of the phase distortion of the power amplifier. In a first form of embodiment said single transistor is placed upstream of the final power amplifier and followed by the phase shift element. In a second form of embodiment its function is fulfilled by the same final power amplifier.
- To achieve said purposes the object of the present invention is a predistortion linearizer for microwave power amplifiers as described in claim 1.
- The variant embodiments described in the dependent claims comprise a further object of the present invention.
- Further objects and advantages of the present invention will be made clear by the following detailed description of an embodiment thereof and the annexed drawings given for purely nonlimiting explanatory purposes and wherein -
- FIG. 1 shows the circuit diagram of a first example of realization of the linearizer which is the object of the present invention,
- FIGS. 2, 3, 4 and 5 show the trends of some characteristic parameters of the linearizer and final power amplifier as a function of the power Pi of the input signal,
- FIG. 6 shows a first embodiment variant of the phase shift element D1 of FIG. 1, and
- FIG. 7 shows a second embodiment variant of said phase shift element.
- FIG. 1 shows a possible form of circuit embodiment of the linearizer which is the object of the invention and is applied upstream of a power amplifier for the purpose of compensating for the amplitude and phase distortions. Indeed the amplitude and phase response curves of a microwave power amplifier are typically as shown in FIG. 2 wherein is shown the qualitative shape of the output power Pu (AM/AM distortion curve) and the input-output phase variation VF (AM/PM conversion curve) as a function of input power Pi. The linearizer has thus a dual purpose, i.e. first to expand its own gain at the Pi values so that in FIG. 2 the knee of the curve Pu occurs to compensate for said knee and continue the linear shape of Pu even in the immediate proximity of saturation, this latter zone being indicated by a horizontal shape which however cannot be offset; the second purpose is to vary its own input-output phase in a manner opposite to the shape of VF in FIG. 2.
- In FIG. 1 RFin, RFout indicate the input and output connectors respectively of a radiofrequency signal.
- C1, C2 ...C5 indicate capacitances and L1, L2 inductances of known type.
- FT1 indicates a GaAsFET transistor (with gallium arsenide field effect) equipped as known with three connectors, source S, drain D and gate G1, used in a common source configuration.
- R1 indicates a polarization resistance of FT1.
- D1 indicates a varactor diode.
- RA1, RA2, RA3 indicate common impedance matching networks for input, interstage and output respectively embodied for example in microstrip.
- VG, VP and V+ indicate the supply voltages of the gate of FT1, the anode of D1 and fixed positive respectively.
- C1 and C2 are used as block capacitances of the continuous component at the input RFin and the output RFout respectively. C4 and the networks C3-L2, C5-L1 form low-pass decoupling filters between signal and supplies VG, VP and V+; C3, C4 and C5 have a grounded end.
- The input signal is applied to the gate of FT1 through C1 and RA1; the voltage Vg is also applied to the gate of FT1 through the filter C3-L2. The source of FT1 is connected to ground while the voltage V+ is brought to the drain through the filter C5-L1 and the resistance R1. The cathode of the diode D1 is connected also continuously to the drain of FT1 through the network RA2 and to the output RFout through the network RA3 and C2.
- The transistor FT1 amplifies the RF signal applied at the input RFin; the values of fixed supply voltage V+, resistance R1 and voltage VG are chosen in such a manner as to keep FT1 in underpolarization conditions, i.e. with the working point near the pinch-off region (low values of drain-source current Ids and gate-source voltage Vgs).
- In this manner an increase in the power of the RF input signal is capable of changing the working point of the device, i.e. increasing the continuous component of the current Ids. Since the gain of FT1 depends on said continuous component of Ids, the increase of the latter causes an increase in gain and supplies the desired expansion effect.
- FIG. 3 shows the qualitative shape of the gain curve Gi obtained as a function of the power Pi of the signal RF at the input RFin. By varying the parameters of the polarization network of FT1, i.e. VG, R1 and V+, it is possible to obtain a shape of Gi optimized from the point of view of compensation of the AM/AM distortion curve. The figures also show as a function of Pi the corresponding qualitative shape of the continuous component of Ids.
- Due to the presence of the resistance R1, as the continuous component of Ids increases the continuous component of the drain-source voltage Vds of FT1 decreases and hence that of the polarization voltage VL of the varactor diode D1 (indeed D1 is connected continuously to the drain of FT1).
- Thus a detected voltage of the modulating signal proportional to the continuous component of Ids is localized and used as a control signal of the phase variation introduced by the downstream device which in the nonlimiting example described is the varactor diode D1.
- The effect obtained is thus a modulation of said voltage FL which, as is known, influences the internal capacitance value of the varactor and hence the phase shift introduced therefrom onto the RF signal. This effect can be used with advantage to offset the conversion curve AM/PM.
- FIG. 4 shows the qualitative shape of the phase variation FL as a function of the power Pi of the input signal. The shape of FL can be made specular in relation to that of VF (shown with broken line in FIG. 4) by appropriate sizing of the parameters which influence it, i.e. R1 and VP.
- To sum up, the transistor FT1 fulfils a dual function: that of gain expander to offset the AM/AM distortion curve and that of detector of a modulation voltage which directly pilots a phase shifter to offset the AM/PM conversion curve.
- More specifically, the gain expansion characteristic is adjusted by varying VG and the phase characteristic is adjusted by varying VP.
- The device of FIG. 1 follows very rapidly the dynamics of the input signal RF, i.e. its response as gain expander and phase shifter is a wide band for the modulating signal because it consists of a single RF branch. This is in effect very important because it makes the device usable even when the multicarrier modulating signal is wide band, e.g. in transponders for satellites.
- In addition there is an undoubted advantage in terms of size and cost reduction of the components of the linearizer which can be embodied with microstrip or Microwave Integrated Circuit (MIC) technology using both discrete packaged components and chip-and-wire or integrated in Monolithic MIC (MMIC) technology.
- Numerous variants on the embodiment described as an example in FIG. 1 are possible without going beyond the scope of the innovative principles contained in the inventive idea.
- The linearization function can be performed by said final power stage consisting for example of a transistor of the FT1 type of FIG.1 brought to a similar working condition.
- This may be explained by a digital example also with reference to FIG. 5 which shows the shapes of the output power Pu of the final power stage in two working conditions, normal (Ids=2A) and modified in accordance with the invention (Ids=0,7A) respectively, and the shape of the continuous component Ids of the drain current of the transistor of said final stage, as a function of the input power Pi.
- Assume a final power stage consisting of a GaAsFET transistor which in linear conditions gains 10dB with working point Fds=10V, Ids=2A and output saturation power Pusat=40dBm at modulated signal frequencies of approximately 6-7GHz. Under these conditions the shape of output power Pu as a function of input power Pi is as indicated in FIG. 5 by parameter Ids=2A with the knee zone to be linearized.
- If we now decrease the gate-source voltage Vgs of the transistor to obtain Ids=0,7A the gain of the linear zone decreases to approximately 7dB. Under these conditions when the input power Pi increases Ids also tends to increase to a nominal value of 2A in conformity with the shape of Ids shown in the figure. The increase of Ids causes an increase in the gain Gi of the transistor just in the zone where the curve Pu(Ids=2A) had the knee, introducing in this case a self-linearization effect: the amplitude in the knee zone of the curve Pu(Ids=0,7A) decreases greatly, offsetting the distortion curve AM/AM.
- As concerns offsetting the conversion curve AM/PM the phase variations can be offset by a circuit of known type upstream piloted by the voltage Vds of the final transistor as described with reference to FIG. 1.
- The variant now described can be used to advantage in QAM multi-level modulation systems where the RF output power is not constant but varies considerably in relation to the average value depending on the point of the constellation of symbols to be transmitted; for example in the 64QAM system there is a difference of approximately 8dB between average and maximum RF output power. There is a considerable current savings; for the low levels of the constellation for which the output power Pu is low the current Ids stays around 0,7A and reaches 2A only for the higher levels for which Pu is high. Since the points of the constellation are all equally probable, the average Ids current will be approximately 1.1-1.2A, and not 2A, with a clear consumption savings. Decrease of gain in the linear zone from 10 to 7dB does not involve problems because the 3dB difference is necessary only at the power peaks.
- A second variant calls for the embodiment of the phase shifter as in FIG. 6 wherein the same symbols as in FIG. 1 indicate the same components interconnected in the same manner.
- In FIG. 6 there has been added a second varicap diode D2 equal and in antiparallel to D1 for the signals. The anode of D2 is connected to the cathode of D1 while the cathode of D2 is polarized by the direct voltage VP2 and is connected to ground for the signals through the filter capacitance C6.
- The embodiment of FIG. 6 serves in case it desired to offset a phase variation which may be either leading or delaying. One diode offsets the delaying phase variations and the other the leading ones; by appropriately sizing the voltages VP and VP2 it is possible to make one of the two diodes work while excluding the other, thus obtaining a leading phase shift with D2 inserted or delaying with D1 inserted.
- A third variant calls for embodiment of the phase shifter as in FIG. 7 wherein the same symbols as in FIG. 1 indicate the same components interconnected in the same manner and performing the same functions.
- In FIG. 7 the phase shifter is embodied by a GaAsFET transistor indicated by FT2 and polarized in such a manner as to function as a normal amplifier in a common source configuration with the continuous component of the drain-source current Ids2≃½Idss. The latter is the maximum value of the drain current for Vgs=0.
- R2 indicates the load resistance of FT2.
- C7, C8 and C9 indicate capacitances and L3, L4 indicate inductances of known type. C9 and the networks C7-L3, C8-L4 form low-pass decoupling filters between signals and supplies for output RFout, gate FT2 and drain FT2; C7, C8 and C9 have one end grounded.
- RA4 indicates an output matching network with the same function as RA1, RA2 and RA3.
- By varying the continuous component of the gate-source voltage of FT2 there is obtained a variation of the continuous component of the drain current Ids which does not cause appreciable variation in the gain of FT2 since FT2 is in linear operation conditions, but the value of gate-source capacitance Cgs of FT2. The effect obtained is equivalent to that supplied by the varactor diode D1 of FIG. 1, i.e. variation of the phase FL of the output signal RFout which offsets the AM/PM conversion curve (see FIG. 4).
- In this case also the voltage detected in the modulation of the modulating signal (which is proportional to the continuous component of Ids of FT1 located at the ends of R1) is used as the control signal of the phase variation introduced by FT2. To bring said control signal to FT2 it is necessary to pair continuously the two transistors FT1 and FT2, bringing to the input of FT2 the voltage taken from R1 either in phase or phase inverted depending on whether the phase variation desired is leading or delaying respectively.
- To introduce a phase delay there can be used the component indicated by AMP in FIG. 7, which is a continuous inverting amplifier with variable gain and a passing band having a width at least double that of the modulating input signal band. AMP receives and amplifies the continuous component of the drain voltage of FT1 at one end of R1 and supplies it with changed sign to the gate of FT2 through the filter C7-L3 to obtain a modulation effect of the continuous component of the gate voltage of FT2. This effect is translated into a variation of the capacitance Cgs of FT2.
- AMP can be embodied by a reversing stage of any known type. If embodied by a GaAsFET transistor the entire circuit of FIG. 7 can be integrated in MMIC technology of simple and economical embodiment.
- If FT2 is not to lead but delay in phase, the amplifier AMP will have the same characteristics as above except that it will not be reversing.
- As another variant, the transistors called for in the various forms of embodiment described can also be of another type, e.g. bipolar.
Claims (11)
- Predistortion linearizer for a final microwave power amplifier subject to gain compression and phase distortion characterized in that it comprises:- a first subpolarized transistor (FT1) which constitutes a gain expander stage with increase in the power of the signal at its input; and- a phase shifter (D1; D1, D2; FT2) controlled by the continuous component of the voltage present at the output of said first transistor, which introduces a phase distortion which offsets that of said final power amplifier.
- Linearizer as in claim 1 characterized in that said first transistor (FT1) is placed upstream of said final power amplifier.
- Linearizer as in claim 1 characterized in that said first transistor (FT1) is said final power amplifier.
- Linearizer as in claim 2 or 3 characterized in that said first transistor (FT1) is a GaAsFET in common source configuration whose gate polarization voltage (VG) is adjustable to obtain said subpolarization.
- Linearizer as in claim 2 or 3 characterized in that said first transistor (FT1) is bipolar in common emitter configuration whose base polarization voltage (VG) is adjustable to obtain said subpolarization.
- Linearizer as in claim 2 characterized in that said phase shifter consists of a varactor diode (D1) placed in parallel for the signals and coupled continuously to the output of said first transistor (FT1), the polarization voltage (VP) of said diode being adjustable.
- Linearizer as in claim 2 characterized in that said phase shifter consists of two varactor diodes (D1, D2) placed in antiparallel for the signals and coupled continuously at the output of said first transistor (FT1), the polarization voltage (VP, VP2) of each of said diodes being independently adjustable.
- Linearizer as in claim 2 characterized in that said phase shifter is an amplifier stage placed downstream of said gain expander stage consisting of a second GaAsFET transistor (FT2) in common source configuration whose input is coupled continuously with the output of said first transistor (FT1).
- Linearizer as in claim 2 characterized in that said phase shifter is an amplifier stage placed downstream of said gain expander stage consisting of a third bipolar transistor (FT2) in common emitter configuration whose input is coupled continuously with the output of said first transistor (FT1).
- Linearizer as in claims 8 or 9 characterized in that said continuous coupling between the first and second transistors is embodied by a wide band inverting continuous amplifier (AMP).
- Linearizer as in claims 8 or 9 characterized in that said continuous coupling between the first and second transistors is embodied by a wide band noninverting continuous amplifier (AMP).
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
IT19975A IT1239472B (en) | 1990-04-09 | 1990-04-09 | LINEARIZER OF THE PRE-DISTORTION TYPE FOR MICROWAVE POWER AMPLIFIERS |
IT1997590 | 1990-04-09 |
Publications (3)
Publication Number | Publication Date |
---|---|
EP0451909A2 true EP0451909A2 (en) | 1991-10-16 |
EP0451909A3 EP0451909A3 (en) | 1992-09-02 |
EP0451909B1 EP0451909B1 (en) | 1994-12-07 |
Family
ID=11162779
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
EP91200799A Expired - Lifetime EP0451909B1 (en) | 1990-04-09 | 1991-04-05 | Predistortion linearizer for microwave power amplifiers |
Country Status (9)
Country | Link |
---|---|
US (1) | US5138275A (en) |
EP (1) | EP0451909B1 (en) |
JP (1) | JP2825993B2 (en) |
AU (1) | AU637209B2 (en) |
DE (1) | DE69105587T2 (en) |
ES (1) | ES2064880T3 (en) |
IT (1) | IT1239472B (en) |
NO (1) | NO177479C (en) |
ZA (1) | ZA912619B (en) |
Cited By (14)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
WO1994015395A1 (en) * | 1992-12-23 | 1994-07-07 | Siemens Telecomunicazioni S.P.A. | Procedure and circuit for adaptive compensation of the gain distortions of a microwave amplifier with linearizer |
DE19616803A1 (en) * | 1995-04-28 | 1996-10-31 | Mitsubishi Electric Corp | Nonlinear distortion compensation unit |
EP0760184A1 (en) * | 1994-05-19 | 1997-03-05 | Ortel Corporation | In-line predistorter for linearization of electronic and optical signals |
ES2098185A1 (en) * | 1992-11-02 | 1997-04-16 | Motorola Inc | Power amplifier linearization in a tdma mobile radio system |
EP0774833A1 (en) * | 1995-11-16 | 1997-05-21 | Ntt Mobile Communications Network Inc. | Amplifying device |
EP0823778A1 (en) * | 1996-08-09 | 1998-02-11 | Trw Inc. | Active feedback pre-distortion linearization |
EP0854569A1 (en) * | 1997-01-21 | 1998-07-22 | Matsushita Electric Industrial Co., Ltd. | High-frequency power amplifier |
EP0883243A2 (en) * | 1997-06-04 | 1998-12-09 | Nec Corporation | A matching circuit and a method for matching a transistor circuit |
WO1999010987A2 (en) * | 1997-08-27 | 1999-03-04 | Ylinen Electronics Oy | Predistortion linearizer circuit |
WO2002015391A1 (en) * | 2000-08-11 | 2002-02-21 | The National University Of Singapore | Programmable radio frequency predistortion linearizer and method thereof |
WO2002043241A1 (en) * | 2000-11-24 | 2002-05-30 | Nokia Corporation | Lineariser |
EP1271765A2 (en) * | 2001-06-08 | 2003-01-02 | TRW Inc. | Application of the doherty amplifier as a predistortion circuit for linearizing microwave amplifiers |
EP1315286A1 (en) * | 2000-08-28 | 2003-05-28 | Mitsubishi Denki Kabushiki Kaisha | Multistage amplifier |
EP1164694B1 (en) * | 2000-06-16 | 2014-05-14 | Nokia Siemens Networks Italia S.p.A. | Microwave amplifier |
Families Citing this family (18)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5280931A (en) * | 1992-11-20 | 1994-01-25 | Thistle Sports Enterprises, Inc. | Roller brake |
US5523716A (en) * | 1994-10-13 | 1996-06-04 | Hughes Aircraft Company | Microwave predistortion linearizer |
US5673001A (en) * | 1995-06-07 | 1997-09-30 | Motorola, Inc. | Method and apparatus for amplifying a signal |
US6223019B1 (en) | 1996-03-14 | 2001-04-24 | Sirius Satellite Radio Inc. | Efficient high latitude service area satellite mobile broadcasting systems |
US6023616A (en) * | 1998-03-10 | 2000-02-08 | Cd Radio Inc. | Satellite broadcast receiver system |
US6255908B1 (en) | 1999-09-03 | 2001-07-03 | Amplix | Temperature compensated and digitally controlled amplitude and phase channel amplifier linearizer for multi-carrier amplification systems |
US6717463B2 (en) * | 2000-11-03 | 2004-04-06 | Qualcomm Incorporated | Circuit for linearizing electronic devices |
JP2002368546A (en) * | 2001-06-06 | 2002-12-20 | Nec Corp | Distortion pre-compensator and linear amplifier using the same |
KR100457785B1 (en) * | 2002-06-05 | 2004-11-18 | 주식회사 웨이브아이씨스 | Apparatus For Pre-distorter With Electronic Tuning |
JP4868433B2 (en) * | 2004-02-09 | 2012-02-01 | ソニー・エリクソン・モバイルコミュニケーションズ株式会社 | Distortion compensation apparatus and power amplification apparatus with distortion compensation function |
US7596326B2 (en) * | 2005-10-27 | 2009-09-29 | Emcore Corporation | Distortion cancellation circuitry for optical receivers |
US7729667B2 (en) * | 2006-02-16 | 2010-06-01 | Raytheon Company | System and method for intermodulation distortion cancellation |
US7634198B2 (en) * | 2006-06-21 | 2009-12-15 | Emcore Corporation | In-line distortion cancellation circuits for linearization of electronic and optical signals with phase and frequency adjustment |
US8391811B2 (en) * | 2009-12-30 | 2013-03-05 | Triquint Semiconductor, Inc. | Input-power overload-protection circuit |
US20120242405A1 (en) * | 2011-03-25 | 2012-09-27 | Younkyu Chung | Frequency-Desensitizer for Broadband Predistortion Linearizers |
WO2016179541A1 (en) | 2015-05-06 | 2016-11-10 | Rfaxis, Inc. | Rf power amplifiers with diode linearizer |
CN113242023B (en) * | 2021-03-30 | 2022-07-26 | 宁波大学 | Analog predistorter of field effect transistor series connection reflection type Schottky diode |
CN114285381B (en) * | 2021-12-20 | 2024-02-09 | 电子科技大学 | Dual-channel broadband linearizer based on common source structure |
Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
GB1039830A (en) * | 1963-11-28 | 1966-08-24 | Standard Telephones Cables Ltd | A radio-frequency transistorised amplifier |
DE2140851A1 (en) * | 1971-08-14 | 1973-02-15 | Rohde & Schwarz | PRE-DISTORTION CIRCUIT FOR NONLINEAR AMPLIFIER |
CA981762A (en) * | 1972-10-27 | 1976-01-13 | Jan J. Spisar | Signal-processing method and device |
US4465980A (en) * | 1982-09-23 | 1984-08-14 | Rca Corporation | Predistortion circuit for a power amplifier |
US4532477A (en) * | 1983-12-23 | 1985-07-30 | At&T Bell Laboratories | Distortion compensation for a microwave amplifier |
Family Cites Families (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JPS5326362B2 (en) * | 1972-09-01 | 1978-08-02 | ||
US3996524A (en) * | 1975-11-05 | 1976-12-07 | Rca Corporation | Linear amplifier utilizing adaptive biasing |
JPS5828766B2 (en) * | 1977-01-22 | 1983-06-17 | 日本電信電話株式会社 | Nonlinear phase distortion compensation circuit |
JPS586327A (en) * | 1981-07-03 | 1983-01-13 | Jidosha Kiki Co Ltd | Glow plug for diesel engine |
IT1215370B (en) * | 1987-02-26 | 1990-02-08 | Gtg Ora Siemens Telecomunicazi | LINEARIZER FOR MICROWAVE POWER AMPLIFIER. |
JP2546279B2 (en) * | 1987-08-05 | 1996-10-23 | ソニー株式会社 | Variable amplitude equalizer |
-
1990
- 1990-04-09 IT IT19975A patent/IT1239472B/en active IP Right Grant
-
1991
- 1991-04-05 ES ES91200799T patent/ES2064880T3/en not_active Expired - Lifetime
- 1991-04-05 DE DE69105587T patent/DE69105587T2/en not_active Expired - Fee Related
- 1991-04-05 EP EP91200799A patent/EP0451909B1/en not_active Expired - Lifetime
- 1991-04-08 AU AU74126/91A patent/AU637209B2/en not_active Ceased
- 1991-04-08 US US07/681,739 patent/US5138275A/en not_active Expired - Fee Related
- 1991-04-08 NO NO911360A patent/NO177479C/en unknown
- 1991-04-09 ZA ZA912619A patent/ZA912619B/en unknown
- 1991-04-09 JP JP3103450A patent/JP2825993B2/en not_active Expired - Fee Related
Patent Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
GB1039830A (en) * | 1963-11-28 | 1966-08-24 | Standard Telephones Cables Ltd | A radio-frequency transistorised amplifier |
DE2140851A1 (en) * | 1971-08-14 | 1973-02-15 | Rohde & Schwarz | PRE-DISTORTION CIRCUIT FOR NONLINEAR AMPLIFIER |
CA981762A (en) * | 1972-10-27 | 1976-01-13 | Jan J. Spisar | Signal-processing method and device |
US4465980A (en) * | 1982-09-23 | 1984-08-14 | Rca Corporation | Predistortion circuit for a power amplifier |
US4532477A (en) * | 1983-12-23 | 1985-07-30 | At&T Bell Laboratories | Distortion compensation for a microwave amplifier |
Non-Patent Citations (1)
Title |
---|
PATENT ABSTRACTS OF JAPAN, vol. 2, no. 120 (E-78)[6900] 06 October 1978; & JP-A-53 085 142 (NIPPON DENSHIN DENWA KOSHA) 27 July 1978 * |
Cited By (29)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
ES2098185A1 (en) * | 1992-11-02 | 1997-04-16 | Motorola Inc | Power amplifier linearization in a tdma mobile radio system |
WO1994015395A1 (en) * | 1992-12-23 | 1994-07-07 | Siemens Telecomunicazioni S.P.A. | Procedure and circuit for adaptive compensation of the gain distortions of a microwave amplifier with linearizer |
AU675479B2 (en) * | 1992-12-23 | 1997-02-06 | Siemens Telecomunicazioni S.P.A. | Procedure and circuit for adaptive compensation of the gain distortions of a microwave amplifier with linearizer |
EP0760184A1 (en) * | 1994-05-19 | 1997-03-05 | Ortel Corporation | In-line predistorter for linearization of electronic and optical signals |
EP0760184A4 (en) * | 1994-05-19 | 2001-08-08 | Ortel Corp | In-line predistorter for linearization of electronic and optical signals |
DE19616803A1 (en) * | 1995-04-28 | 1996-10-31 | Mitsubishi Electric Corp | Nonlinear distortion compensation unit |
FR2733646A1 (en) * | 1995-04-28 | 1996-10-31 | Mitsubishi Electric Corp | DISTORTION COMPENSATION CIRCUIT |
US5815038A (en) * | 1995-04-28 | 1998-09-29 | Mitsubishi Denki Kabushiki Kaisha | Distortion compensation circuit |
EP0774833A1 (en) * | 1995-11-16 | 1997-05-21 | Ntt Mobile Communications Network Inc. | Amplifying device |
US5796307A (en) * | 1995-11-16 | 1998-08-18 | Ntt Mobile Communications Network Inc. | Amplifying device having input and output nonlinear phase shifters of opposite phase-frequency characteristics |
EP0823778A1 (en) * | 1996-08-09 | 1998-02-11 | Trw Inc. | Active feedback pre-distortion linearization |
US5808511A (en) * | 1996-08-09 | 1998-09-15 | Trw Inc. | Active feedback pre-distortion linearization |
EP0854569A1 (en) * | 1997-01-21 | 1998-07-22 | Matsushita Electric Industrial Co., Ltd. | High-frequency power amplifier |
US5982236A (en) * | 1997-01-21 | 1999-11-09 | Matsushita Electric Industrial Co., Ltd. | High-frequency power amplifier |
KR100310623B1 (en) * | 1997-06-04 | 2002-08-21 | 닛본 덴기 가부시끼가이샤 | Matching method of matching circuit and transistor circuit |
EP0883243A2 (en) * | 1997-06-04 | 1998-12-09 | Nec Corporation | A matching circuit and a method for matching a transistor circuit |
EP0883243A3 (en) * | 1997-06-04 | 1999-09-08 | Nec Corporation | A matching circuit and a method for matching a transistor circuit |
US6130589A (en) * | 1997-06-04 | 2000-10-10 | Nec Corporation | Matching circuit and a method for matching a transistor circuit |
WO1999010987A3 (en) * | 1997-08-27 | 1999-05-20 | Insinoeoeritoimisto Juhana Yli | Predistortion linearizer circuit |
US6346853B1 (en) | 1997-08-27 | 2002-02-12 | Ylinen Electronics Oy | Predistortion linearizer circuit |
WO1999010987A2 (en) * | 1997-08-27 | 1999-03-04 | Ylinen Electronics Oy | Predistortion linearizer circuit |
EP1164694B1 (en) * | 2000-06-16 | 2014-05-14 | Nokia Siemens Networks Italia S.p.A. | Microwave amplifier |
WO2002015391A1 (en) * | 2000-08-11 | 2002-02-21 | The National University Of Singapore | Programmable radio frequency predistortion linearizer and method thereof |
EP1315286A1 (en) * | 2000-08-28 | 2003-05-28 | Mitsubishi Denki Kabushiki Kaisha | Multistage amplifier |
EP1315286A4 (en) * | 2000-08-28 | 2005-08-10 | Mitsubishi Electric Corp | Multistage amplifier |
WO2002043241A1 (en) * | 2000-11-24 | 2002-05-30 | Nokia Corporation | Lineariser |
EP1271765A2 (en) * | 2001-06-08 | 2003-01-02 | TRW Inc. | Application of the doherty amplifier as a predistortion circuit for linearizing microwave amplifiers |
EP1271765A3 (en) * | 2001-06-08 | 2004-06-16 | Northrop Grumman Corporation | Application of the doherty amplifier as a predistortion circuit for linearizing microwave amplifiers |
US6864742B2 (en) | 2001-06-08 | 2005-03-08 | Northrop Grumman Corporation | Application of the doherty amplifier as a predistortion circuit for linearizing microwave amplifiers |
Also Published As
Publication number | Publication date |
---|---|
ZA912619B (en) | 1992-01-29 |
AU637209B2 (en) | 1993-05-20 |
JP2825993B2 (en) | 1998-11-18 |
AU7412691A (en) | 1991-10-10 |
EP0451909B1 (en) | 1994-12-07 |
IT9019975A0 (en) | 1990-04-09 |
DE69105587D1 (en) | 1995-01-19 |
NO911360L (en) | 1991-10-10 |
NO911360D0 (en) | 1991-04-08 |
DE69105587T2 (en) | 1995-05-18 |
JPH04252506A (en) | 1992-09-08 |
EP0451909A3 (en) | 1992-09-02 |
US5138275A (en) | 1992-08-11 |
NO177479B (en) | 1995-06-12 |
NO177479C (en) | 1995-09-20 |
ES2064880T3 (en) | 1995-02-01 |
IT1239472B (en) | 1993-11-02 |
IT9019975A1 (en) | 1991-10-09 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US5138275A (en) | Predistortion linearizer for microwave power amplifiers | |
CA1191215A (en) | Active predistorter for linearity compensation | |
US9030255B2 (en) | Linearization circuit and related techniques | |
US5915213A (en) | Transmitter apparatus with linearizing circuit to compensate for distortion in power amplifier | |
US20020125947A1 (en) | Linearizer for a power amplifier | |
US6043707A (en) | Method and apparatus for operating a radio-frequency power amplifier as a variable-class linear amplifier | |
US3755754A (en) | Predistortion compensation for a microwave amplifier | |
US6255908B1 (en) | Temperature compensated and digitally controlled amplitude and phase channel amplifier linearizer for multi-carrier amplification systems | |
US5644268A (en) | Feed forward RF amplifier for combined signal and error amplification | |
US7332961B2 (en) | Predistortion linearizing | |
US5576660A (en) | Broadband predistortion linearizer with automatic temperature compensation for microwave amplifiers | |
EP1724917A1 (en) | Feedforward amplifier | |
KR20010064900A (en) | adaptive predistortion linearizer | |
US4752743A (en) | Linearizer for TWT amplifiers | |
US4532477A (en) | Distortion compensation for a microwave amplifier | |
GB2158625A (en) | Predistortion circuit | |
US5568087A (en) | Pre-distorting linearization circuit | |
US5247542A (en) | QPSK power amplifier distortion correction system | |
US4772855A (en) | Linearizer for microwave power amplifiers | |
US7221221B2 (en) | Power amplifier with pre-distorter | |
US6191652B1 (en) | Amplifier distortion correction using cross-modulation | |
CN113659938B (en) | Analog predistorter | |
KR950003277B1 (en) | Feedback linearizating circuit for power amplifier | |
CN117411443A (en) | Reflection type analog predistortion linearizer | |
CN117439553A (en) | Analog predistortion structure with independently adjustable amplitude and phase |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
PUAI | Public reference made under article 153(3) epc to a published international application that has entered the european phase |
Free format text: ORIGINAL CODE: 0009012 |
|
AK | Designated contracting states |
Kind code of ref document: A2 Designated state(s): CH DE ES FR GB GR IT LI SE |
|
PUAL | Search report despatched |
Free format text: ORIGINAL CODE: 0009013 |
|
AK | Designated contracting states |
Kind code of ref document: A3 Designated state(s): CH DE ES FR GB GR IT LI SE |
|
17P | Request for examination filed |
Effective date: 19930122 |
|
17Q | First examination report despatched |
Effective date: 19930607 |
|
GRAA | (expected) grant |
Free format text: ORIGINAL CODE: 0009210 |
|
AK | Designated contracting states |
Kind code of ref document: B1 Designated state(s): CH DE ES FR GB GR IT LI SE |
|
ITF | It: translation for a ep patent filed |
Owner name: SIEMENS - ING. CORRADO BORSANO |
|
REF | Corresponds to: |
Ref document number: 69105587 Country of ref document: DE Date of ref document: 19950119 |
|
REG | Reference to a national code |
Ref country code: ES Ref legal event code: FG2A Ref document number: 2064880 Country of ref document: ES Kind code of ref document: T3 |
|
ET | Fr: translation filed | ||
REG | Reference to a national code |
Ref country code: GR Ref legal event code: FG4A Free format text: 3014498 |
|
PLBE | No opposition filed within time limit |
Free format text: ORIGINAL CODE: 0009261 |
|
STAA | Information on the status of an ep patent application or granted ep patent |
Free format text: STATUS: NO OPPOSITION FILED WITHIN TIME LIMIT |
|
26N | No opposition filed | ||
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: GB Payment date: 19970321 Year of fee payment: 7 |
|
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: ES Payment date: 19970418 Year of fee payment: 7 Ref country code: GR Payment date: 19970418 Year of fee payment: 7 |
|
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: SE Payment date: 19970421 Year of fee payment: 7 |
|
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: CH Payment date: 19970715 Year of fee payment: 7 |
|
PG25 | Lapsed in a contracting state [announced via postgrant information from national office to epo] |
Ref country code: GB Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 19980405 |
|
PG25 | Lapsed in a contracting state [announced via postgrant information from national office to epo] |
Ref country code: ES Free format text: LAPSE BECAUSE OF THE APPLICANT RENOUNCES Effective date: 19980406 Ref country code: SE Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 19980406 |
|
PG25 | Lapsed in a contracting state [announced via postgrant information from national office to epo] |
Ref country code: CH Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 19980430 Ref country code: GR Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 19980430 Ref country code: LI Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 19980430 |
|
GBPC | Gb: european patent ceased through non-payment of renewal fee |
Effective date: 19980405 |
|
REG | Reference to a national code |
Ref country code: CH Ref legal event code: PL |
|
EUG | Se: european patent has lapsed |
Ref document number: 91200799.4 |
|
REG | Reference to a national code |
Ref country code: ES Ref legal event code: FD2A Effective date: 20001009 |
|
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: DE Payment date: 20070430 Year of fee payment: 17 |
|
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: IT Payment date: 20070626 Year of fee payment: 17 |
|
PGFP | Annual fee paid to national office [announced via postgrant information from national office to epo] |
Ref country code: FR Payment date: 20070416 Year of fee payment: 17 |
|
PG25 | Lapsed in a contracting state [announced via postgrant information from national office to epo] |
Ref country code: DE Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 20081101 |
|
REG | Reference to a national code |
Ref country code: FR Ref legal event code: ST Effective date: 20081231 |
|
PG25 | Lapsed in a contracting state [announced via postgrant information from national office to epo] |
Ref country code: FR Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 20080430 |
|
PG25 | Lapsed in a contracting state [announced via postgrant information from national office to epo] |
Ref country code: IT Free format text: LAPSE BECAUSE OF NON-PAYMENT OF DUE FEES Effective date: 20080405 |