EP2082478A2 - System and method for on-chip im3 reduction over a broad range of operating powers - Google Patents
System and method for on-chip im3 reduction over a broad range of operating powersInfo
- Publication number
- EP2082478A2 EP2082478A2 EP07843972A EP07843972A EP2082478A2 EP 2082478 A2 EP2082478 A2 EP 2082478A2 EP 07843972 A EP07843972 A EP 07843972A EP 07843972 A EP07843972 A EP 07843972A EP 2082478 A2 EP2082478 A2 EP 2082478A2
- Authority
- EP
- European Patent Office
- Prior art keywords
- branch
- biasing
- mode
- amplifier
- stage
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Withdrawn
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Classifications
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/26—Modifications of amplifiers to reduce influence of noise generated by amplifying elements
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/02—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation
- H03F1/0205—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers
- H03F1/0261—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the polarisation voltage or current, e.g. gliding Class A
- H03F1/0266—Modifications of amplifiers to raise the efficiency, e.g. gliding Class A stages, use of an auxiliary oscillation in transistor amplifiers with control of the polarisation voltage or current, e.g. gliding Class A by using a signal derived from the input signal
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/04—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements with semiconductor devices only
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/189—High frequency amplifiers, e.g. radio frequency amplifiers
- H03F3/19—High frequency amplifiers, e.g. radio frequency amplifiers with semiconductor devices only
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/20—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers
- H03F3/21—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only
- H03F3/211—Power amplifiers, e.g. Class B amplifiers, Class C amplifiers with semiconductor devices only using a combination of several amplifiers
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/60—Amplifiers in which coupling networks have distributed constants, e.g. with waveguide resonators
- H03F3/602—Combinations of several amplifiers
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/432—Two or more amplifiers of different type are coupled in parallel at the input or output, e.g. a class D and a linear amplifier, a class B and a class A amplifier
Definitions
- This invention generally relates to radio frequency/microwave amplifiers for communication devices, and in particular, to a system and method for automatically reducing third order intermodulation product levels to reduce signal distortion and
- Amplifiers are used in all cell phones to amplify a modulated signal from a transceiver prior to passing the signal through the front end of the cell phone, which typically consists of passive components and a switch into an antenna where the signal is transmitted to a base station.
- Fig. 1 illustrates a high level topography of the circuits
- a baseband integrated chip 1 provides a signal to a transceiver 2.
- the transceiver transmits an RF signal to a power amplifier 3, which outputs an RF signal to a front end module 4 and out into the atmosphere by an
- antenna 5 AU of these components are housed within a typical cell phone 6.
- linearity refers to the device's ability to amplify without distortion
- efficiency refers to the device's ability to convert DC power to RF/microwave power with as little wasted energy as possible.
- improvement in one area typically causes degradation in the other.
- An RF/microwave amplifier has two regions of operation: linear and nonlinear. In the linear region, the input signal envelope is amplified and no distortion is present at the output. For large peak to peak input signal levels, the amplifier enters the non-linear region and the output signal becomes distorted.
- Distortion in RF/microwave amplifiers is generally caused by amplitude clipping, phase variations as a function of signal amplitude, and intermodulation products.
- Amplitude clipping occurs when the peak to peak input signal envelope amplitude extends beyond the linear region of the amplifier.
- Phase variations with signal amplitude also result when the peak to peak input signal envelope amplitude extends beyond the linear region of the amplifier.
- Intermodulation distortion occurs as a result of nonlinearities in the amplifier transfer function resulting in mixing products being generated at the sum and difference frequencies of the input signals.
- the third order intermodulation product (IMD3) is of great interest since this product is very close to the carrier signal on the frequency spectrum. Being located so close to the carrier signal, IMD3 is very difficult to eliminate or even reduce and is often the limiting factor in the linearity of the RF/microwave amplifier.
- the output third-order intercept point (OIP3), also known as OTOI (Output Third Order Intercept), is defined as the intersection point between the extrapolated 1:1 slope of the fundamental output power and the extrapolated 3:1 slope of the third-order intermodulation products. If the extrapolations are done well within the linear region, the OIP3 (OTOI) becomes a useful specification for predicting the linearity of the power amplifier. Thus, the higher the OIP3 (OTOI) point, the more linear the power amplifier. As mentioned above, reducing the IMD of the power amplifier improves its linearity and thus improves the OIP3 (OTOI).
- power amplifier 3 comprises one or more stages 7a, 7b . . .
- Each stage 7a, 7b . . . 7n consists of a number of branches 18a, 18b . . . 18n connected in parallel.
- Each branch 18 has of one or more unit cells 20 connected in parallel.
- a unit cell is composed of one or more transistors in a circuit topology of zero or more resistors, capacitors, and inductors.
- Each power amplifier stage 7 is typically biased with a bias circuit (not shown) that provides the appropriate current or voltage for the branches to operate in a single class of operation.
- a class of operation is determined by the percentage of an input sinusoidal signal during which the unit cells in each branch is on and conducting current.
- the branches are all biased such that they are on and conducting current for 360 degrees of the input sinusoid signal.
- the branches are on and conducting for 180 degrees of the input signal.
- the near-Class B biasing condition is for the case where the branches are on and the conduction angle is close to but above 180 degrees.
- Class AB the branches are on and conducting typically at or around 270 degrees but may vary between 180 degrees and 360 degrees.
- the limits for each class of operation are not rigorously set and are used herein for purposes of general understanding of the operating condition of an amplifier.
- One method is to operate the RF/microwave amplifier at lower power levels to ensure that the device remains in the linear region.
- a drawback to this method is that when the device is operated at a lower power level it is operating less efficiently than it does at higher power levels.
- the present invention seeks to improve the output response of RF/microwave power amplifiers by reducing IM3 levels and improving the OIP3 (OTOI) of the power amplifier.
- One solution to this problem is to reduce the distortion present in the linear and nonlinear region so that the operation of the amplifier can be extended into the nonlinear region where higher power levels are possible.
- a discrete amplifier for canceling at least one distortion component at the output of the discrete amplifier comprising a first stage having at least a first branch and a second branch in parallel with one another, wherein each of the first and second branches is formed of one or more transistor legs having one or more unit cells in parallel with one another and a second stage having at least a third branch and a fourth branch in parallel with one another, wherein each of the third and fourth branches is formed of one or more transistor legs having one or more unit cells in parallel with one another.
- the discrete amplifier further comprises a first biasing circuit having a first impedance and a first biasing level, the first biasing circuit being operatively connected to the first stage first branch for biasing the first branch into a first mode of operation, a second biasing circuit having a second impedance and a second biasing level, the second biasing circuit being operatively connected to the first stage second branch for biasing the second branch into a second mode of operation, a third biasing circuit having a third impedance and a third biasing level, the third biasing circuit being operatively connected to the second stage third branch for biasing the third branch into a third mode of operation, and a fourth biasing circuit having a fourth impedance and a fourth biasing level, the fourth biasing circuit being operatively connected to the second stage fourth branch for biasing the fourth branch into a fourth mode of operation. At least one of the first mode of operation and the second mode of operation, and the third mode of operation and the fourth mode of operation are chosen so that at least one distortion component is substantially canceled at
- the first biasing circuit first impedance and first biasing level differs from the second biasing circuit second impedance and second biasing level.
- the impedance of the first and second biasing circuits and a biasing level of the first and second branches are different, and the impedance of the third and fourth biasing circuits and a biasing level of the third and fourth branches is different and are chosen so as to maximize the cancellation of the at least one distortion component at the output of at least one of the first stage and the second stage of the discrete amplifier.
- the first, the second, the third and the fourth braches are formed on a single integrated circuit chip and the first biasing circuit, the second biasing circuit, the third biasing circuit and the fourth biasing circuit are physically located on the same single integrated circuit chip to minimize electrical parasitics so as to maximize the cancellation of the at least one distortion component.
- Other embodiments further comprise a sensing circuit that senses the input power to the discrete amplifier and causes at least one of the first, the second, the third and the fourth biasing circuits to adjust the operating mode of its respective branch by adjusting at least one of the bias circuit current and impedance so as to maximize the cancellation of the at least one distortion component.
- the first mode, the second mode, the third mode and the fourth mode are chosen from one of Class A, Class B, Class C, and Class AB modes of operation, and the first, the second, the third and the fourth biasing circuits are adjusted to change the impedance of the respective biasing circuits to maximize the cancellation of the at least one distortion component.
- a circuit for canceling at least one distortion component at the output of the circuit comprises a power splitter having a first input port, a first output port, and a second output port, wherein a first signal is input on the power splitter first input port and is split into a second signal at the power splitter first output port and a third signal at the power splitter second output port, and a power combiner having a first input port, a second input port, and a first output port.
- a first amplifier has an input port coupled to the power splitter first output port, and an output port coupled to the power combiner first input port.
- a second amplifier is in parallel with the first amplifier, the second amplifier has an input port coupled to the power splitter second output port, and an output port coupled to the power combiner second input port.
- At least one biasing circuit is operatively connected with one of the first and the second amplifiers for biasing the one of the first and the second amplifiers in a different class of operation than the other of the first and the second amplifiers, and at least one control logic circuit senses the first signal and causes at least one of a size of the one of the first and the second amplifiers to change based on the first signal, and a magnitude of a biasing level of the one of the first and the second amplifiers to change based on the first signal.
- a fourth signal is output from the power combiner first output port having at least one distortion component reduced.
- a second biasing circuit is operatively connected to the other of the first and the second amplifiers for biasing the other of the first and the second amplifiers into a class of operation different than the one of the first and the second amplifiers so that the at least one distortion component is reduced in the fourth signal.
- the at least one distortion component is a third harmonic of the first signal.
- the modes of operation are chosen from one of Class A, Class B, Class C, and Class AB modes of operation.
- the power splitter and the power combiner are quadrature hybrids.
- FIG. 1 illustrates a schematic of a typical RF system in a prior art cell phone
- FIG. 2 illustrates a schematic of a prior art multi-stage RF/microwave power amplifier for use in the system of FIG. 1;
- FIG. 3 illustrates a schematic of power combined amplifiers in accordance with an embodiment of the present invention
- FIG. 4 illustrates the large-signal K3 curve versus bias level for the amplifiers in
- FIG. 5 illustrates the dependence of OIP3 on the bias point of the amplifiers of
- FIG. 6 illustrates the improvement in OIP3 seen when the bias point of the two amplifiers of FIG. 3 is chosen such that K3 is equal to -K3'.
- FIG. 7 illustrates a schematic for a balanced amplifier with control loop in accordance with an embodiment of the present invention.
- FIG. 8 shows the flowchart for control logic of FIG. 7.
- FIG. 9 illustrates a schematic of power combined branches in one stage of a power amplifier in accordance with an embodiment of the present invention.
- FIG. 10 illustrates a schematic of power combined branches in one stage of a power amplifier with control logic in accordance with an embodiment of the present invention.
- FIG. 11 illustrates a flowchart for the control logic of Fig. 10.
- a component level quadrature balanced amplifier topology generally consists of a pre- amplifier quadrature hybrid 110, an amplifier stage consisting of two amplifiers 114 and 116 in parallel and biased in different classes of operation, and a post-amplifier quadrature hybrid 112.
- the input, V 1n consists of a sum of sinusoids at two different frequencies expressed as:
- Vin A Sm(W 1 1) + Bs ⁇ n(w 2 t)
- the input signal V 1n is passed to quadrature hybrid 110 positioned before the amplifier stage.
- quadrature hybrid is the SHY550 90 degree hybrid coupler manufactured by Mid- Atlantic RF Systems, Inc.
- the input is divided into two signals with the same frequency components, but with a fraction of the power of the input and where one, Vj, is shifted by ⁇ radians and the other signal, V Q , is shifted by - ⁇ /2 radians resulting in a relative phase difference of 90 degrees:
- Vi sin(w 2 / - ⁇ )
- amplifiers 114 and 116 are then passed to amplifiers 114 and 116, respectively.
- suitable amplifiers are Model EIC5359-8 10- Volt internally matched power FET manufactured by Excelics Semiconductor, Inc.
- the signal amplification performed by each amplifier can be modeled respectively using the following approximate equations:
- VAX K 1 * Vi + Ki * Vi 2 + Ki * Vi'
- V ⁇ 2 Ki * VQ + KI * VQ 2 + Ki * VQ 3
- K 1 represents the coefficients of the third-order voltage transfer function of amplifier 114
- K 1 ' represents the coefficients of the third-order voltage transfer function of amplifier 116.
- K 3 term in amplifier 114 and the K 3 ' term in amplifier 116 are equal and opposite.
- the phase of the fundamental frequency components are substantially equal resulting in maximum power combining of the fundamental components.
- Second hybrid 112 which is the same type as the first hybrid and is used to combine the RF signals present at the output of the two amplifiers into one RP output signal.
- Hybrid 112 has two inputs (ports 1 and 4) and two outputs (ports 2 and 3).
- the "top" output port, port 2 consists of the addition of a signal from port 1 with a fraction of the power and shifted by ⁇ radians and a signal from port 4 with a fraction of the power and shifted by ⁇ /2 radians.
- the "bottom" output port, port 3, consists of the addition of a signal from
- the bottom output port is taken to be the output of the device and, under ideal conditions, the IMD3 products will be substantially removed from the frequency spectrum and the fundamentals still remain.
- the following equations show what happens to the EV1D3 terms present at the "bottom" output port, VO2.
- V ⁇ 2 K,
- Figure 4 shows the dependence of K 3 on the bias level of the amplifier.
- the phase of K 3 can be changed by operating the amplifier in either the gain compression mode or the gain expansion mode.
- the gain compression mode the gain of the amplifier decreases with increasing input power whereas in the gain expansion mode, the amplifier gain increases with increasing input power.
- the output power at the fundamental frequency can be modeled using the following terms: Ki * A + K 3 * A 3 + K 5 * A 5 where A is the amplitude of the input voltage signal and the K terms represent the Taylor series expansion coefficients that describe the amplifier's output voltage to input voltage transfer characteristics. The even order K terms do not contribute to the IMD3 components and hence have not been included in the equations.
- the K 3 and K 5 terms are positive provided Ki is positive and for gain compression, the K 3 and K 5 terms are negative.
- the third order IMD3 components for a two tone excitation also contain the K 3 and K 5 terms. This implies that the IMD3 components are in phase during gain expansion and 180 degrees out of phase with respect to the fundamental for gain compression. In RF amplifiers, this can be achieved by adjusting the biasing of the amplifier by varying the bias levels and/or using a biasing circuit block with specific input impedance values.
- the correct size must be chosen for each amplifier.
- the size of the amplifier is directly related to how much power the amplifier can produce.
- the amplifier itself is made of many single transistors connected in parallel. Each individual transistor has a standard size depending on the fabrication process. Each transistor also contributes a portion of the total output power when the transistors are connected in parallel. As the number of transistors connected in parallel increases, so does the total output power provided by the amplifier. Therefore, size, in this case, refers to the number of transistors multiplied by the amount of active area for each transistor.
- the near-class B mode amplifier must be sized so that its IMD3 magnitudes are substantially equal to the IMD3 magnitudes of the class AB mode amplifier at the same input power levels.
- One method for determining the size is by testing a number of near-class B mode amplifiers of differing sizes for the appropriate input power levels to determine the appropriate IMD3 magnitudes. That is, as the IMD3 magnitudes of the two parallel amplifiers diverge, the improvement in OIP3 seen at the output of the output quadrature hybrid 112 will not be as great had the IMD3 levels been substantially equal in magnitude and opposite in phase. Moreover, the IMD3 magnitudes of amplifiers 114 and 116 may be substantially equal at one power level, but unequal at another. [0044] As shown in Figure 7, one method of overcoming this problem is by using a correction signal that is generated by monitoring the RF input signal at hybrid 110 to control the size and biasing level of amplifier 116.
- the RF input signal Vj n is sampled and fed into a control logic block 118 that generates a digital word or correction signal by comparing input power level Vj n with levels stored in a look-up table, which are associated with the appropriate size and bias level for near-class B amplifier 116.
- the correction signal controls the size of near-class B amplifier 116 by controlling the number of parallel transistors that are active in the near-class B amplifier by turning on and off certain portions of the near-class B amplifier.
- control logic 118 determines that certain portions of the near-class B amplifier should be turned off, then some of the transistors in the near-class B amplifier are shut off by reducing their bias level to zero. If, on the other hand, control logic 118 determines that certain portions of the near-class B amplifier should be turned on, then some transistors that had their bias level at zero will be raised to the near-class B level. Bias control logic 118 is located in the bias circuitry for near-class B amplifier 116. As the near-class B amplifier decreases in size, so does its IMD3 components. Likewise, as the near-class B amplifier increases in size, its resultant IMD3 components increase as well. As a result of the size adjustment capability, it possible to see OIP3 improvement across a wide range of input power levels.
- Figure 8 illustrates the steps for the control logic to handle changes made to the near-class B amplifier.
- the input power level is first sensed at step 120, and the information is used to decide what changes should be made to the near-class B amplifier at step 124.
- a decision will be made whether to increase or decrease the size of the amplifier and to increase or decrease the bias of the amplifier, respectively, to enhance IMD3 cancellation.
- the appropriate changes are made at step 130 to the near-class B amplifier to make the magnitudes of the IMD3 components of the two amplifiers substantially equal, the phases of the IMD3 components 180 degrees out of phase with respect to one another, and the phases of the fundamentals equal.
- Figure 6 further illustrates the improvement shown in Figure 5 by illustrating that for a small range of input power levels there is improvement in OIP3 of up to 4dB.
- the OIP3 improvement would be over a wider power range had the experiment incorporated control loop 118 shown in Figure 7. Therefore, while an improvement can be obtained by choosing amplifiers of similar size, variable sizing and biasing of the near-class B amplifier will result in 0EP3 improvement over a broad power range.
- the size and/or bias level of the near-class B amplifier its IMD3 magnitudes could be adjusted to better match the IMD3 magnitudes of the class AB amplifier over a wide range of input powers resulting in an improvement in 0EP3 over a wider input power range.
- EVI3 levels can be significantly reduced or canceled at the sub-component level. More particularly, the IM3 levels at each stage 7 a , 7 b . . . 7 n of power amplifier 3 may be significantly reduced or canceled by biasing two sets of branches in a power amplifier stage in different conduction angles of operation as described by class of operation (i.e. Class A, Class B, Class AB) with two separates bias circuits. Each biasing circuit would have distinct output impedances, with the impedance at baseband frequencies being of particular interest. Baseband frequencies are defined as the range of modulation frequencies for the input signal of interest, which typically ranges from the kHz to tens of MHz.
- Biasing parallel branches allows the phase difference between the IM3 components of each set of branches to approach 180 degrees. Additionally, the difference in the output impedance at baseband frequencies for the bias circuits allows for additional phase shift for the IM3 components, enabling a 180 degree difference between the IM3 components to be achieved.
- the number of branches in each set is adjusted such that the magnitudes of the IM3 components are substantially equal. That is, because the amplification of the sets of branches must be substantially equal to maximize performance, the number of branches in each set is adjusted to compensate for amplification differences.
- the conduction angle of operation (i.e. Class A, Class AB, Class B, near-Class B) of the power amplifier branches is adjusted by varying the dc operating current in the branches with the bias circuit.
- control voltages and/or control currents are varied in the bias circuit to change the dc operating current and hence the conduction angle of operation.
- the impedance of the bias circuit can be modified, in one implementation, by using a resistor in series with the bias circuit. The value of the resistor can be varied to obtain different impedance values.
- one stage of a power amplifier 7a consists of three components: an on-chip power divider 10, two sets of branches 12 and 14 belonging to one stage of an amplifier with separate bias circuits (not shown) biasing each set of branches, followed by an on-chip power combiner 16.
- Each set of branches 12 and 14 comprises one or more branches 18 a , 18 b . . . 18 n of cells 20.
- Cells 20 can consist of one or more transistors and other passive circuit elements.
- the input, V 1 consists of a sum of sinusoids at two different frequencies:
- V x A Sm(W 1 /) + B sin(w 2 /) [0053]
- the input signal V 1 is passed to on-chip power divider 10.
- One example of a suitable on-chip power divider is a microstrip line with a characteristic impedance of Z 0 split into two quarter wave length lines with a characteristic impedance of Z 0 *sqrt(2).
- the input is divided into two signals with the same frequency components, but with a fraction of the power of the input and where one, V 2 , and the other signal, V 3 , have no
- V 4 K lA x V 6 +K 2A x V 6 2 +K 3A x V 6 3
- V 5 K w x V 7 +K 2B x V 7 2 +K iB x V 7 3
- K 1A represents the coefficients of the third-order voltage transfer function of branch set 12
- K, B represents the coefficients of the third-order voltage transfer function of branch set 14.
- branch sets 12 and 14 are passed to second on-chip power combiner 16 that may or may not be the same type as first on-chip power divider 10 and used to combine the RF signals present at the output of the two branch sets into one RF
- V 6 ( K IA + KU> ) [ A . sin ( Wj/ ) + B . sin(w 2 r)]
- K 1A is substantially equal to K IB , then the fundamental term will be present at the output of on-chip power combiner 16. Having K 1A substantially equal K IB also ensures that the phases of the fundamental components for both sets of branches are equal.
- each set of branches 12 and 14 the correct size must be chosen for each set of branches.
- the size of each set of branches is directly related to how much power it can produce. Two branches with the same number of unit cells biased at different points (i.e. different conduction angles of operation for each branch) will generate different IMD3 levels. For example, a branch with a certain number of unit cells 20 biased in the class AB mode of operation will generate significantly less IMD3 terms then a branch of the same size biased in the near-class B mode of operation at the same input power level.
- the near-class B mode amplifier must be sized so that its IMD3 magnitudes are substantially equal to the IMD3 magnitudes of the class AB mode amplifier at the same input power levels.
- One method for determining the size is by testing a number of near- class B mode branch sets to determine the appropriate IMD3 magnitudes. That is, as the IMD3 magnitudes of the two parallel set of branches diverge, then the improvement in OIP3 seen at the output of the output power combiner 16 will not be as great had the IMD3 levels been substantially equal in magnitude and opposite in phase.
- the IMD3 magnitudes for branch sets 12 and 14 may be substantially equal at one power level, but unequal at another.
- the number of branch sets in a particular amplifier stage may also change based on the application of the amplifier.
- the number of branch sets in any one amplifier stage 7 may vary depending on the application of power amplifier 3.
- the phase of K 3 can be changed by operating the amplifier in either the gain compression mode or the gain expansion mode.
- the gain compression mode the gain of the branch or set of branches decreases with increasing input power whereas in the gain expansion mode, the branch gain increases with increasing input power.
- the output power at the fundamental frequency can be modeled using the following terms: Ki * A + K 3 * A 3 + K 5 * A 5 where A is the amplitude of the input voltage signal and the K terms represent the Taylor series expansion coefficients that describe the branch's output voltage to input voltage transfer characteristics. The even order K terms do not contribute to the IMD3 components and hence have not been included in the equations.
- the K 3 and K 5 terms are positive provided Ki is positive, and for gain compression, the K 3 and K 5 terms are negative.
- the third order DVID components for a two tone excitation also contain the K 3 and K 5 terms. This implies that the IMD3 components are in phase with respect to the fundamental during gain expansion and anti-phase with respect to the fundamental for gain compression. In RF amplifiers, this can be achieved by adjusting the biasing of one or more branches by varying the bias levels and using biasing circuits with different impedances.
- the range of power levels where improvement is observed may vary as this technique is used in different semiconductor IC processes, and the incorporation of a control loop may assure the improvement over a broad power range.
- K 3 term as related to the mathematical derivations, may vary as a function of RP input power.
- a control loop may be used to compensate for the changes in K 3 by adjusting the bias level (i.e. conduction angle) and the impedance of the bias circuit of both power amplifier branch sets. This dynamic compensation for the changes in K 3 as a function of RF power allows the IMD3 magnitudes and phases to be adjusted for optimal IM3 cancellation/reduction over a wide range of input powers.
- a control loop 22 is shown connected from input V 1 to the biasing circuit (not shown) for branch sets 12 and 14 of amplifier stage 7a.
- a correction signal 24 is generated by monitoring the RF input signal Vi to control the biasing level and impedance of the bias circuit (not shown) of branch sets 12 and 14 as a function of power. That is, the RF input signal Vi is sampled and fed into control logic block 22, which generates the appropriate bias levels for the applied RF input power.
- These bias levels may be generated by any number of means, such as (but not limited to) comparing the input power level Vi with levels stored in a look-up table or by using a circuit block to directly generate the respective bias levels for the specific input power level.
- Correction signal 24 controls the bias levels and bias circuit impedance of branch sets 12 and 14 by varying the control voltages for the bias circuit (not shown). More specifically, if control logic 22 determines that branch set 14 should be biased more towards a Class AB mode, then the control voltages of the bias circuits will be adjusted accordingly. Bias control logic 22, in one implementation, can be located in the bias circuitry for branch set 14 and also in branch set 12. Also, the impedance of the bias circuit can be adjusted with switches that turn on or off segments of a parallel resistor array (not shown). As a result of the bias/impedance adjustment capability of the bias circuit (not shown), OIP3 improvement across a wide range of power levels can be obtained.
- Vi is first sensed at step 26, and the information is used to determine the changes that should be made to branch sets 12 and 14 by comparing the sensed signal to, for example, (1) values stored in a look-up table, in step 28 or (2) converting the sensed signal to the respective bias/impedance levels for the bias circuits.
- the possible changes to the bias circuits are: (1) change the impedance of the bias circuit, step 34 or (2) increase/decrease the bias of the branches, step 32.
- the above described amplifier improvements for EVB reduction results in very wide band solutions and can be implemented completely on-chip.
- the techniques are based on a unique power amplifier topology combined with a novel biasing technique at the amplifier branch level.
- the techniques are very flexible and can be applied to any IC process (i.e. Si CMOS, SiGe BiCMOS, GaAs HBT, etc..)
Abstract
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US85014606P | 2006-10-06 | 2006-10-06 | |
PCT/US2007/080699 WO2008043098A2 (en) | 2006-10-06 | 2007-10-08 | System and method for on-chip im3 reduction over a broad range of operating powers |
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EP (1) | EP2082478A4 (en) |
JP (1) | JP2010506514A (en) |
KR (1) | KR20090084843A (en) |
CN (1) | CN101589547A (en) |
WO (1) | WO2008043098A2 (en) |
Families Citing this family (6)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP2884659B1 (en) | 2013-12-10 | 2016-08-24 | Stichting IMEC Nederland | Harmonics Cancellation Circuit for a Power Amplifier |
KR102491944B1 (en) * | 2017-12-27 | 2023-01-25 | 삼성전기주식회사 | Power amplifier with asymmetric amplification structure for improving linearity |
KR102029558B1 (en) | 2017-12-27 | 2019-10-07 | 삼성전기주식회사 | Power amplifier with improved wideband linearity |
CN108400774B (en) * | 2018-03-22 | 2020-06-02 | 上海唯捷创芯电子技术有限公司 | Balanced radio frequency power amplifier, chip and communication terminal |
KR102457874B1 (en) | 2020-03-09 | 2022-10-25 | 충남대학교산학협력단 | Power amplifier using multi-stage linearization |
WO2023162173A1 (en) * | 2022-02-25 | 2023-08-31 | 三菱電機株式会社 | Low distortion amplifier |
Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20030179041A1 (en) * | 2002-03-19 | 2003-09-25 | The University Of North Carolina At Charlotte | Method and apparatus for cancellation of third order intermodulation distortion and other nonlinearities |
US20040085132A1 (en) * | 2002-11-06 | 2004-05-06 | Cree Microwave, Inc. | RF transistor amplifier linearity using suppressed third order transconductance |
US20040150473A1 (en) * | 2003-02-05 | 2004-08-05 | Sirenza Microdevices, Inc | Distortion cancellation for RF amplifiers using complementary biasing circuitry |
Family Cites Families (1)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
EP1520396A4 (en) * | 2002-06-27 | 2006-05-17 | Broadband Innovations Inc | Even order distortion elimination in push-pull or differential amplifiers and circuits |
-
2007
- 2007-10-08 EP EP07843972A patent/EP2082478A4/en not_active Withdrawn
- 2007-10-08 JP JP2009531641A patent/JP2010506514A/en not_active Withdrawn
- 2007-10-08 CN CNA2007800449305A patent/CN101589547A/en active Pending
- 2007-10-08 KR KR1020097009077A patent/KR20090084843A/en not_active Application Discontinuation
- 2007-10-08 WO PCT/US2007/080699 patent/WO2008043098A2/en active Application Filing
Patent Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20030179041A1 (en) * | 2002-03-19 | 2003-09-25 | The University Of North Carolina At Charlotte | Method and apparatus for cancellation of third order intermodulation distortion and other nonlinearities |
US20040085132A1 (en) * | 2002-11-06 | 2004-05-06 | Cree Microwave, Inc. | RF transistor amplifier linearity using suppressed third order transconductance |
US20040150473A1 (en) * | 2003-02-05 | 2004-08-05 | Sirenza Microdevices, Inc | Distortion cancellation for RF amplifiers using complementary biasing circuitry |
Non-Patent Citations (2)
Title |
---|
MARK P VAN DER HEIJDEN ET AL: "Theory and Design of an Ultra-Linear Square-Law Approximated LDMOS Power Amplifier inClass-AB Operation" IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, IEEE SERVICE CENTER, PISCATAWAY, NJ, US, vol. 50, no. 9, 1 September 2002 (2002-09-01), XP011076697 ISSN: 0018-9480 * |
See also references of WO2008043098A2 * |
Also Published As
Publication number | Publication date |
---|---|
JP2010506514A (en) | 2010-02-25 |
CN101589547A (en) | 2009-11-25 |
WO2008043098A2 (en) | 2008-04-10 |
WO2008043098A3 (en) | 2008-08-28 |
EP2082478A4 (en) | 2010-01-27 |
KR20090084843A (en) | 2009-08-05 |
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