US20050206412A1 - High frequency differential power amplifier - Google Patents
High frequency differential power amplifier Download PDFInfo
- Publication number
- US20050206412A1 US20050206412A1 US10/802,894 US80289404A US2005206412A1 US 20050206412 A1 US20050206412 A1 US 20050206412A1 US 80289404 A US80289404 A US 80289404A US 2005206412 A1 US2005206412 A1 US 2005206412A1
- Authority
- US
- United States
- Prior art keywords
- cmos inverter
- input
- output
- coupled
- supply voltage
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Granted
Links
- 230000003321 amplification Effects 0.000 claims abstract description 7
- 238000003199 nucleic acid amplification method Methods 0.000 claims abstract description 7
- 230000001105 regulatory effect Effects 0.000 claims abstract 7
- 239000003990 capacitor Substances 0.000 claims description 44
- 238000000034 method Methods 0.000 claims description 11
- 239000004065 semiconductor Substances 0.000 claims description 10
- 230000000295 complement effect Effects 0.000 claims description 6
- 238000001914 filtration Methods 0.000 claims description 5
- 230000008878 coupling Effects 0.000 claims 8
- 238000010168 coupling process Methods 0.000 claims 8
- 238000005859 coupling reaction Methods 0.000 claims 8
- 239000013256 coordination polymer Substances 0.000 description 8
- 230000007423 decrease Effects 0.000 description 5
- 238000010586 diagram Methods 0.000 description 4
- 229920006395 saturated elastomer Polymers 0.000 description 3
- 230000001413 cellular effect Effects 0.000 description 2
- 230000003247 decreasing effect Effects 0.000 description 2
- 230000000903 blocking effect Effects 0.000 description 1
- 230000015556 catabolic process Effects 0.000 description 1
- 238000006731 degradation reaction Methods 0.000 description 1
- 230000000694 effects Effects 0.000 description 1
- 238000002955 isolation Methods 0.000 description 1
- 230000010355 oscillation Effects 0.000 description 1
Images
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3205—Modifications of amplifiers to reduce non-linear distortion in field-effect transistor amplifiers
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/30—Modifications of amplifiers to reduce influence of variations of temperature or supply voltage or other physical parameters
- H03F1/301—Modifications of amplifiers to reduce influence of variations of temperature or supply voltage or other physical parameters in MOSFET amplifiers
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/42—Indexing scheme relating to amplifiers the input to the amplifier being made by capacitive coupling means
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/54—Two or more capacitor coupled amplifier stages in cascade
Definitions
- the invention relates to the field of high frequency communications, an in particular to a complementary metal-oxide-semiconductor high frequency amplifier.
- a differential amplifier is a fundamental electronic circuit that generates an output signal based on the difference between two input signals (a differential input signal). The output signal is therefore representative of the magnitude of the difference between the two input signals.
- differential amplifiers are often implemented using a metal-oxide-semiconductor (MOS) or complementary MOS (CMOS) process instead of the more expensive bipolar process.
- MOS metal-oxide-semiconductor
- CMOS complementary MOS
- FIG. 1 shows a conventional RF MOS differential amplifier 100 .
- MOS differential amplifier 100 includes input terminals 101 - 1 and 101 - 2 , capacitors C 1 _IN, C 2 _IN, and C_GND, NMOS resistors R 1 _SET, R 2 _SET, R 1 _DN, R 2 _DN, and R_BIAS, transistors 111 and 112 , output terminals 102 - 1 and 102 - 2 , and a current source CS 1 .
- Capacitor C 1 _IN is coupled between input terminal 101 - 1 and the gate of transistor 111
- capacitor C 2 _IN is coupled between input terminal 101 - 2 and the gate of transistor 112 .
- Capacitors C 1 _IN and C 2 _IN therefore provide DC filtering of input RF signals V_IN + and V_IN ⁇ , respectively, which are applied to input terminals 101 - 1 and 101 - 2 , respectively.
- resistors R 1 _SET and R 2 _SET (which typically are the same resistance) couple the drains of transistors 111 and 112 , respectively, to an upper supply voltage VDD, while current source CS 1 couples the sources of transistors 111 and 112 to a lower supply voltage VSS.
- output terminals 102 - 1 and 102 - 2 are connected to the drains of transistors 111 and 112 , respectively.
- transistors 111 and 112 are configured as a differential pair.
- Resistors R 1 _DN and R 2 _DN in conjunction with resistor R_BIAS, provide a desired bias voltage V_BIAS to the gates of transistors 111 and 112 , respectively.
- capacitor C_GND provides an AC short between resistors R 1 _DN and R 2 _DN and lower supply voltage VSS, thereby setting the input impedances seen at the sources of transistors 111 and 112 equal to the values of resistors R 1 _DN and R 2 _DN, respectively.
- the differential input signal V_DIFF(IN) (equal to V_IN + minus V_IN ⁇ ) provided to differential amplifier 100 during balanced operations is equal to zero, and a bias current I_BIAS provided by current source CS 1 is equally divided between transistors 111 and 112 (if resistors R 1 _SET and R 2 _SET have equal resistances).
- a differential current I_DIFF flows across transistors 111 and 112 .
- I_DIFF V — DIFF ( IN )/(1 /g m111 +1/ g m112 ) (1) where g m111 and g m112 are the transconductances of transistors 111 and 112 , respectively.
- output signals V_OUT + and V_OUT ⁇ are then determined by the magnitude of differential current I_DIFF and resistors R 1 _SET and R 2 _SET, respectively.
- the gain provided by differential amplifier 100 can be increased by either increasing resistance R_SET (i.e., the resistances of resistors R 1 _SET and R 2 _SET), or by increasing transconductance gm (i.e., by increasing transconductances g m111 and g m112 ).
- Another problematic issue relates to the fact that increasing the size of resistors R 1 _SET and R 2 _SET and/or increasing current I_BIAS can significantly increase the power consumption of differential amplifier 100 .
- This power inefficiency is generally undesirable, and can be particularly problematic in devices that run off of a self-contained power supply (a battery). For example, using amplifier 100 in a cellular telephone to reduce the overall cost of the phone may result in an unacceptable decrease in talk time for that phone.
- a high-frequency differential amplifier includes two CMOS inverters and biasing circuitry.
- the CMOS inverters apply a desired gain to a differential input signal based on the transconductance and output impedance values of the transistors making up the inverters.
- the biasing circuitry applies linear biasing to the CMOS inverters without consuming excessive power.
- the biasing circuitry provides a DC feedback loop that forces a DC bias voltage to appear at the outputs of the inverters.
- the inverters can be forced to operate in their linear region. AC signals at the inputs of the inverters will then be amplified by the inverters without distortion (clipping), so long as the amplitudes of the AC signals are not large enough to drive either inverter out of its linear mode of operation.
- the biasing circuitry includes a reference voltage source and a separate bias circuit for each inverter, with each bias circuit including an operational amplifier (op-amp).
- the op-amp in each bias circuit is connected in a feedback loop between the output and input of one of the inverters, while the reference voltage source provides a reference voltage to the non-inverting input of the op-amp.
- the op-amp therefore adjusts the input voltage of its associated inverter to regulate the output of that inverter to be equal to the reference voltage.
- This DC control provided by each op-amp ensures that the inverters will operate in their linear regions as long as the input signals are not large enough to push the transistors of the inverters into saturation.
- the reference voltage equal to half of the voltage difference between the upper and lower supply voltages provided to the amplifier, the output range of the amplifier can be maximized.
- FIG. 1 is a schematic diagram of a conventional CMOS RF differential amplifier.
- FIG. 2A is a schematic diagram of a CMOS high-frequency differential amplifier circuit in accordance with an embodiment of the invention.
- FIG. 2B is a sample graph of the response curve of an inverter, depicting the linear and saturated regions of operation of the inverter.
- FIG. 3 is a schematic diagram of a branch of the CMOS high-frequency differential amplifier circuit of FIG. 2A that includes a detail view of a schematic for an operational amplifier in accordance with an embodiment of the invention.
- FIG. 2A shows a high-frequency amplifier circuit 200 in accordance with an embodiment of the invention.
- Amplifier circuit 200 is formed from two branches 200 (A) and 200 (B).
- Branch 200 (A) includes an input terminal 201 (A), an output terminal 202 (A), a CMOS inverter 210 (A), a capacitor C_IN(A), and a bias circuit 220 (A).
- Capacitor C_IN(A) is coupled between input terminal 201 (A) and the input of inverter 210 (A) and provides DC filtering at the input of inverter 210 (A).
- Bias circuit 220 (A) is connected between the output and input of inverter 210 (A).
- Inverter 210 (A) includes a PMOS transistor M 1 (A) and an NMOS transistor M 2 (A) that are serially coupled between an upper supply voltage VDD and a lower supply voltage (e.g., ground).
- the gate terminals of transistors M 1 (A) and M 2 (A) are connected to form the input of inverter 210 (A), while the drain terminals of transistors M 1 (A) and M 2 (A) are connected to form the output of inverter 210 (A).
- Branch 200 (B) is substantially similar to branch 200 (A), and includes an input terminal 201 (B), an output terminal 202 (B), a CMOS inverter 210 (B), a capacitor C_IN(B), and a bias circuit 220 (B).
- Capacitor C_IN(B) is coupled between input terminal 201 (B) and the input of inverter 210 (B) and provides DC filtering at the input of inverter 210 (B).
- Bias circuit 220 (B) is connected between the output and input of inverter 210 (B).
- Inverter 210 (B) includes a PMOS transistor M 1 (B) and an NMOS transistor M 2 (B) that are serially coupled between upper supply voltage VDD and lower supply voltage VSS.
- the gate terminals of transistors M 1 (B) and M 2 (B) are connected to form the input of inverter 210 (B), while the drain terminals of transistors M 1 (B) and M 2 (B) are connected to form the output of inverter 210 (B).
- Amplifier circuit 200 is coupled to receive a high-frequency input signal V_IN + at input terminal 201 (A) and a high-frequency input signal V_IN ⁇ at input terminal 201 (B).
- High-frequency signals V_IN + and V_IN ⁇ can, for example, comprise RF signals.
- bias circuits 220 (A) and 220 (B) provide linear biasing feedback loops between the outputs and inputs of inverters 210 (A) and 210 (B), respectively.
- bias circuit 220 (A) provides a DC bias voltage to the input of inverter 210 (A) that forces the nominal output of inverter 210 (A) to a level between upper supply voltage VDD and lower supply voltage VSS, which in turn causes inverter 210 (A) to operate in its linear region.
- bias circuit 220 (B) provides a DC bias voltage to the input of inverter 210 (B) that forces the nominal output of inverter 210 (B) to a level between upper supply voltage VDD and lower supply voltage VSS, which in turn causes inverter 210 (B) to operate in its linear region.
- the outputs of both inverters 210 (A) and 210 (B) are forced to midway between upper supply voltage VDD and lower supply voltage VSS to allow for maximum output swing.
- FIG. 2B shows an exemplary response curve C for inverters 210 (A) and 210 (B).
- Response curve C consists of two main regions—a saturated region that corresponds to all input voltages less than a lower limit voltage V_DN or greater than an upper limit voltage V_UP, and a linear region that corresponds to all input voltages between voltages V_DN and V_UP. Because the normal use of an inverter is to invert a logic LOW or HIGH input signal into a logic HIGH or LOW output signal, respectively, an inverter is generally operated in its saturated region, and will only incidentally pass through its linear region as its output switches between logic LOW (GND) and logic HIGH (VDD).
- bias circuits 220 (A) and 220 (B) forces inverters 210 (A) and 210 (B), respectively, to operate in their linear regions, so that inverters 210 (A) and 210 (B) can be used to provide signal amplification.
- the DC bias voltages supplied by bias circuits 220 (A) and 220 (B) force the nominal inverter output voltages (i.e., the voltages at the outputs of the inverters when no AC signal is present) for inverters 210 (A) and 210 (B) to levels between upper supply voltage VDD and lower supply voltage VSS.
- inverters 210 (A) and 210 (B) will therefore swing around this nominal inverter output voltage, thereby ensuring that inverters 210 (A) and 210 (B) provide AC output signals that are proportional to their AC input signals (so long as the AC input signal amplitude does not push inverters 210 (A) and 210 (B) into saturation).
- the nominal inverter output voltage equal to half of the difference between upper supply voltage VDD and lower supply voltage VSS (e.g., if supply voltage VSS is ground, then the nominal inverter output voltage would be VDD/2)
- the total output swing of differential amplifier 200 can be maximized (i.e., output swing equal to 2*VDD). Note that because inverters 210 (A) and 210 (B) do not include any resistive elements, this increased gain does not result in output signal distortion (unlike the results described with respect to conventional differential amplifier 100 shown in FIG. 1 ).
- bias circuit 220 (A) includes resistors R_IN(A) and R_OUT(A), optional capacitors C 221 (A) and C 222 (A), and an operational amplifier (op-amp) 240 (A).
- Resistor R_IN(A) is connected between the input of inverter 210 (A) and the output of op-amp 240 (A), while resistor R_OUT(A) is connected between the output of inverter 210 (A) and the non-inverting input of op-amp 240 (A).
- Capacitor C 221 (A) is connected between the output of op-amp 240 (A) and ground, while capacitor C 222 (A) is connected between the non-inverting input of op-amp 240 (A) and ground.
- bias circuit 220 (B) includes resistors R_IN(B) and R_OUT(B), optional capacitors C 221 (B) and C 222 (B), and an operational amplifier (op-amp) 240 (B).
- Resistor R_IN(B) is connected between the input of inverter 210 (B) and the output of op-amp 240 (B), while resistor R_OUT(B) is connected between the output of inverter 210 (B) and the non-inverting input of op-amp 240 (B).
- Capacitor C 221 (B) is connected between the output of op-amp 240 (B) and ground, while capacitor C 222 (B) is connected between the non-inverting input of op-amp 240 (B) and ground.
- Reference voltage source 230 provides a reference voltage V_MID to the inverting inputs of op-amps 240 (A) and 240 (B). Meanwhile, the non-inverting inputs of op-amps 240 (A) and 240 (B) receive the outputs of inverters 210 (A) and 210 (B), respectively (via resistors R_OUT(A) and R_OUT(B), respectively). If the voltage at the output of inverter 210 (A) is less than reference voltage V_MID, op-amp 240 (A) decreases its output voltage (and hence the voltage provided at the input of inverter 210 (A) via resistor R_IN(A)), thereby raising the output of inverter 210 (A).
- op-amp 240 (A) increases its output voltage to decrease the output of inverter 210 (A).
- Op-amp 240 (B) regulates the output of inverter 210 (B) in a similar manner.
- op-amps 240 (A) and 240 (B) create DC bias voltages at the inputs of inverters 210 (A) and 210 (B), respectively, such that each inverter has a DC offset voltage at its output that is equal to reference voltage V_MID.
- This DC biasing of the inverter inputs forces inverters 210 (A) and 210 (B) to operate in the linear mode, so that gain can be applied to signals provided to inverters 210 (A) and 210 (B) without distortion (clipping).
- reference voltage V_MID can be set to any value between supply voltage VDD and ground (the upper and lower supply voltages)
- resistors R_IN(A) and R_OUT(A) isolate op-amp 240 (A) from any AC signals that are provided to or generated by inverter 210 (A) by suppressing the bulk of those signals before they reach op-amp 240 (A).
- optional capacitors C 221 (A) and C 222 (A) can provide a direct path to ground for any AC that does get by resistors R_IN(A) and R_OUT(A), respectively, or is generated by op-amp 240 (A).
- resistors R_IN(B) and R_OUT(B) and capacitors C 221 (A) and C 222 (B) provide AC isolation for op-amp 240 (B).
- bias circuits 220 (A) and 220 (B) do not include constant bias currents (e.g., currents I_BIAS- 1 and I_BIAS- 2 shown in FIG. 1 ) flowing through large resistive elements (e.g., resistors RD(A) and/or RD(B) shown in FIG. 1 ), the power consumption of amplifier circuit 200 shown in FIG. 2A can be significantly less than the power consumption of conventional amplifier 100 .
- inverters 210 (A) and 210 (B) can both provide a significant amount of gain (while operating in their linear regions).
- Ro 2 represents the parallel resistance of Ro 1 and Ro 2 , and resolves to the equation: Ro l
- Gain G is therefore proportional to transconductance g m and output resistance Ro.
- the gain provided by an inverter-based differential amplifier such as shown in FIG. 2A is inversely proportional to drain current, and is therefore not subject to the output distortion associated with common-source based amplifier 100 shown in FIG. 1 .
- the transconductance is proportional to the aspect ratio (width/length) of the gate. Therefore, by adjusting the gate dimensions of transistors M 1 (A) and M 2 (A), the gain provided by branch 200 (A) of amplifier circuit 200 can be adjusted. For similar reasons, by adjusting the gate dimensions of transistors M 1 (B) and M 2 (B), the gain provided by branch 200 (B) can be adjusted.
- supply voltage VDD can be 1.8V
- reference voltage V_MID can be set to 0.9V
- transistors M 1 (A) and M 1 (B) can have aspect ratios of 27/0.3
- transistors M 2 (A) and M 2 (B) can have aspect ratios of 21.6/0.3
- resistors R_IN(A), R_OUT(A), R_IN(B), and R_OUT(B) can have resistances of 1.5 k ⁇ each
- capacitors C_IN(A), C_OUT(A), C_IN(B), and C_OUT(B) can have capacitances of 150 fF each.
- Branches 200 (A) and 200 (B) would then provide between 10-15 dB of RF gain each.
- branches 200 (A) and 200 (B) shown in FIG. 2A are described as single stages for exemplary purposes, each of branches 200 (A) and 200 (B) can comprise a stage in a series of cascaded amplifier stages, or a predriver for additional amplifier circuitry, as indicated by optional (dotted line) amplifier stage circuitry 290 (A) and 290 (B).
- FIG. 3 shows a detailed view of branch 200 (A) that depicts a schematic diagram for op-amp 240 (A), according to an embodiment of the invention.
- Op-amp 240 (A) includes PMOS transistors M 3 and M 5 , NMOS transistors M 4 , M 6 , M 7 , and M 8 , a current source 241 , a capacitor C_CP, and a resistor R_CP.
- Transistors M 3 and M 4 are connected in series between supply voltage VDD and transistor M 8 , and transistors M 5 and M 6 are connected in series between supply voltage VDD and transistor M 8 .
- Transistor M 8 is coupled between transistor M 4 and ground, and current source 241 and transistor M 7 are connected in series between supply voltage VDD and ground.
- capacitor C_CP and resistor R_CP are connected in series between the gate of transistor M 4 and the drain of transistor M 6 .
- the gate of transistor M 4 forms the non-inverting input of op-amp 240 (A), and is accordingly coupled to the input of inverter 210 (A) via resistor R_OUT(A).
- the gate of transistor M 6 forms the inverting input of op-amp 240 (A), and is therefore coupled to reference voltage circuit 230 (A).
- the junction between transistors M 5 and M 6 forms the output of op-amp 240 (A), and is therefore coupled to the input of inverter 210 (A) via resistor R_IN(A).
- capacitor C_CP and resistor R_CP are coupled between the non-inverting input and the output of op-amp 240 (A).
- Capacitor C_CP and resistor R_CP form a compensation circuit that improves the stability of op-amp 240 (A) by preventing unwanted oscillations. Note that various other op-amp compensation circuits will be readily apparent.
- transistor M 7 The gate and drain of transistor M 7 are shorted, and the gates of transistors M 7 and M 8 are connected to form a current mirror. Therefore, a current I_BIAS from current source 241 that is sunk by transistor M 7 is also mirrored by transistor M 8 . Therefore, a total current I_BIAS flows through the two branches formed by transistors M 3 and M 4 (first branch) and by transistors M 5 and M 6 (second branch).
- transistor M 3 the gate and drain of transistor M 3 are shorted, and the gates of transistors M 3 and M 5 are connected to form another current mirror that provides a load for the differential pair formed by transistors M 4 and M 6 .
- transistors M 3 and M 5 split the flow of current I_BIAS equally through transistors M 4 and M 6 .
- transistor M 5 adjusts its drain voltage (i.e., the output of op-amp 240 (A)) in response.
- transistor M 4 For example, if the voltage provided at the gate of transistor M 4 (i.e., the voltage at the output of inverter 210 (A)) is greater than the voltage provided at the gate of transistor M 6 (i.e., reference voltage V_MID), then transistor M 4 is turned on more strongly than transistor M 6 , and the current flow through transistor M 4 increases. Since the total current flow through transistors M 4 and M 6 is fixed at current I_BIAS by transistor M 8 , this increase in current flow through transistor M 4 means that the current flow through transistor M 6 must decrease.
- the drain voltage of transistor M 6 is increased. This has the effect of reducing the gate-drain voltage of transistor M 6 , which in turn reduces the current flow through transistor M 6 . Meanwhile, this increased drain voltage of transistor M 6 is applied to the input of inverter 210 (A) (via resistor R_IN(A)), thereby driving the voltage at the output of inverter 210 (A) down towards reference voltage V_MID.
- transistor M 4 is turned on less strongly than transistor M 6 , and the current flow through transistor M 4 decreases. Therefore, the current flow through transistor M 6 must increase, and the drain voltage of transistor M 6 is decreased to increase the gate-drain voltage of transistor M 6 .
- This decreased drain voltage of transistor M 6 is applied to the input of inverter 210 (A), thereby driving the voltage at the output of inverter 210 (A) up towards reference voltage V_MID.
- circuitry shown for op-amp 240 (A) in FIG. 3 is exemplary only. Alternatives may be found in the conventional art.
- capacitors C_IN(A) and C_IN(B) could be removed from differential amplifier 200 in FIG. 2A , thereby enabling amplification of DC input voltages at input terminals 201 (A) and 201 (B).
- the invention is limited only by the following claims and their equivalents.
Abstract
Description
- 1. Field of the Invention
- The invention relates to the field of high frequency communications, an in particular to a complementary metal-oxide-semiconductor high frequency amplifier.
- 2. Related Art
- A differential amplifier is a fundamental electronic circuit that generates an output signal based on the difference between two input signals (a differential input signal). The output signal is therefore representative of the magnitude of the difference between the two input signals. To reduce costs (which is particularly important for consumer goods such as cellular telephones), differential amplifiers are often implemented using a metal-oxide-semiconductor (MOS) or complementary MOS (CMOS) process instead of the more expensive bipolar process.
-
FIG. 1 shows a conventional RF MOSdifferential amplifier 100. MOSdifferential amplifier 100 includes input terminals 101-1 and 101-2, capacitors C1_IN, C2_IN, and C_GND, NMOS resistors R1_SET, R2_SET, R1_DN, R2_DN, and R_BIAS,transistors - Capacitor C1_IN is coupled between input terminal 101-1 and the gate of
transistor 111, while capacitor C2_IN is coupled between input terminal 101-2 and the gate oftransistor 112. Capacitors C1_IN and C2_IN therefore provide DC filtering of input RF signals V_IN+ and V_IN−, respectively, which are applied to input terminals 101-1 and 101-2, respectively. - Meanwhile, resistors R1_SET and R2_SET (which typically are the same resistance) couple the drains of
transistors transistors transistors - Thus,
transistors transistors transistors - The differential input signal V_DIFF(IN) (equal to V_IN+ minus V_IN−) provided to
differential amplifier 100 during balanced operations is equal to zero, and a bias current I_BIAS provided by current source CS1 is equally divided betweentransistors 111 and 112 (if resistors R1_SET and R2_SET have equal resistances). However, as is known in the art, when differential input signal V_DIFF(IN) is not equal to zero, a differential current I_DIFF flows acrosstransistors
I — DIFF=V — DIFF(IN)/(1/g m111+1/g m112) (1)
where gm111 and gm112 are the transconductances oftransistors - The magnitude of output signals V_OUT+ and V_OUT− are then determined by the magnitude of differential current I_DIFF and resistors R1_SET and R2_SET, respectively. For example, output signal V_OUT+ is given by the following:
V — OUT + =VDD−R 1 — SET(½I — BIAS+I — DIFF) (2)
Similarly, output signal V_OUT− is given by the following:
V — OUT − =VDD−R 2 — SET(½I — BIAS−I — DIFF) (3)
If resistors R1_SET and R2_SET are both equal to the same resistance R_SET, equations 2 and 3 can be combined to determine the magnitude of an output differential signal V_DIFF(OUT) (equal to V_OUT+ minus V_OUT−) as follows:
V — DIFF(OUT)=−2R — SET*I — DIFF (4)
Finally, if the transconductances oftransistors equation 1 can be substituted into equation 4, so that the magnitude of output differential signal V_DIFF(OUT) resolves to: - Thus, as indicated by equation 5, the gain provided by
differential amplifier 100 can be increased by either increasing resistance R_SET (i.e., the resistances of resistors R1_SET and R2_SET), or by increasing transconductance gm (i.e., by increasing transconductances gm111 and gm112). - Unfortunately, because of the common-source implementations used in
differential amplifier 100, increasing resistance R_SET and/or increasing transconductance gm can result in undesirable output signal degradation. For example, increasing the resistance of resistors R1_SET and R2_SET can lead to excessive voltage drops between supply voltage VDD and output terminals 102-1 and 102-2, respectively, that distort the output signal swing. Similarly, increasing transconductances gm111 and gm112 (and possibly increasing bias current I_BIAS) will result in larger current magnitudes through resistors R1_SET and R2_SET, respectively, which once again can lead to excessive voltage drops. - Another problematic issue relates to the fact that increasing the size of resistors R1_SET and R2_SET and/or increasing current I_BIAS can significantly increase the power consumption of
differential amplifier 100. This power inefficiency is generally undesirable, and can be particularly problematic in devices that run off of a self-contained power supply (a battery). For example, usingamplifier 100 in a cellular telephone to reduce the overall cost of the phone may result in an unacceptable decrease in talk time for that phone. - Accordingly, it is desirable to provide a power-efficient, high frequency CMOS differential amplifier.
- According to an embodiment of the invention, a high-frequency differential amplifier includes two CMOS inverters and biasing circuitry. The CMOS inverters apply a desired gain to a differential input signal based on the transconductance and output impedance values of the transistors making up the inverters. Meanwhile, the biasing circuitry applies linear biasing to the CMOS inverters without consuming excessive power.
- The biasing circuitry provides a DC feedback loop that forces a DC bias voltage to appear at the outputs of the inverters. By selecting the DC bias voltage to be between the logic HIGH and LOW output levels of the inverters, the inverters can be forced to operate in their linear region. AC signals at the inputs of the inverters will then be amplified by the inverters without distortion (clipping), so long as the amplitudes of the AC signals are not large enough to drive either inverter out of its linear mode of operation.
- According to an embodiment of the invention, the biasing circuitry includes a reference voltage source and a separate bias circuit for each inverter, with each bias circuit including an operational amplifier (op-amp). The op-amp in each bias circuit is connected in a feedback loop between the output and input of one of the inverters, while the reference voltage source provides a reference voltage to the non-inverting input of the op-amp. The op-amp therefore adjusts the input voltage of its associated inverter to regulate the output of that inverter to be equal to the reference voltage.
- This DC control provided by each op-amp ensures that the inverters will operate in their linear regions as long as the input signals are not large enough to push the transistors of the inverters into saturation. By setting the reference voltage equal to half of the voltage difference between the upper and lower supply voltages provided to the amplifier, the output range of the amplifier can be maximized.
- These and other aspects of the invention will be more fully understood in view of the following description of the exemplary embodiments and the drawings thereof.
-
FIG. 1 is a schematic diagram of a conventional CMOS RF differential amplifier. -
FIG. 2A is a schematic diagram of a CMOS high-frequency differential amplifier circuit in accordance with an embodiment of the invention. -
FIG. 2B is a sample graph of the response curve of an inverter, depicting the linear and saturated regions of operation of the inverter. -
FIG. 3 is a schematic diagram of a branch of the CMOS high-frequency differential amplifier circuit ofFIG. 2A that includes a detail view of a schematic for an operational amplifier in accordance with an embodiment of the invention. -
FIG. 2A shows a high-frequency amplifier circuit 200 in accordance with an embodiment of the invention.Amplifier circuit 200 is formed from two branches 200(A) and 200(B). Branch 200(A) includes an input terminal 201(A), an output terminal 202(A), a CMOS inverter 210(A), a capacitor C_IN(A), and a bias circuit 220(A). Capacitor C_IN(A) is coupled between input terminal 201(A) and the input of inverter 210(A) and provides DC filtering at the input of inverter 210(A). Bias circuit 220(A) is connected between the output and input of inverter 210(A). - Inverter 210(A) includes a PMOS transistor M1(A) and an NMOS transistor M2(A) that are serially coupled between an upper supply voltage VDD and a lower supply voltage (e.g., ground). The gate terminals of transistors M1(A) and M2(A) are connected to form the input of inverter 210(A), while the drain terminals of transistors M1(A) and M2(A) are connected to form the output of inverter 210(A).
- Branch 200(B) is substantially similar to branch 200(A), and includes an input terminal 201(B), an output terminal 202(B), a CMOS inverter 210(B), a capacitor C_IN(B), and a bias circuit 220(B). Capacitor C_IN(B) is coupled between input terminal 201(B) and the input of inverter 210(B) and provides DC filtering at the input of inverter 210(B). Bias circuit 220(B) is connected between the output and input of inverter 210 (B).
- Inverter 210(B) includes a PMOS transistor M1(B) and an NMOS transistor M2(B) that are serially coupled between upper supply voltage VDD and lower supply voltage VSS. The gate terminals of transistors M1(B) and M2(B) are connected to form the input of inverter 210(B), while the drain terminals of transistors M1(B) and M2(B) are connected to form the output of inverter 210(B).
-
Amplifier circuit 200 is coupled to receive a high-frequency input signal V_IN+ at input terminal 201(A) and a high-frequency input signal V_IN− at input terminal 201(B). High-frequency signals V_IN+ and V_IN− can, for example, comprise RF signals. - Meanwhile, bias circuits 220(A) and 220(B) provide linear biasing feedback loops between the outputs and inputs of inverters 210(A) and 210(B), respectively. In other words, bias circuit 220(A) provides a DC bias voltage to the input of inverter 210(A) that forces the nominal output of inverter 210(A) to a level between upper supply voltage VDD and lower supply voltage VSS, which in turn causes inverter 210(A) to operate in its linear region. Similarly, bias circuit 220(B) provides a DC bias voltage to the input of inverter 210(B) that forces the nominal output of inverter 210(B) to a level between upper supply voltage VDD and lower supply voltage VSS, which in turn causes inverter 210(B) to operate in its linear region. Ideally, the outputs of both inverters 210(A) and 210(B) are forced to midway between upper supply voltage VDD and lower supply voltage VSS to allow for maximum output swing.
- Note that this DC biasing of the outputs of inverters 210(A) and 210(B) is common mode for both inverters, and therefore cancels itself out when the outputs of the inverters are taken as a differential output. Therefore, blocking capacitors are not required between the outputs of inverters 210(A) and 210(B) and output terminals 202(A) and 202(B), respectively.
-
FIG. 2B shows an exemplary response curve C for inverters 210(A) and 210(B). Response curve C consists of two main regions—a saturated region that corresponds to all input voltages less than a lower limit voltage V_DN or greater than an upper limit voltage V_UP, and a linear region that corresponds to all input voltages between voltages V_DN and V_UP. Because the normal use of an inverter is to invert a logic LOW or HIGH input signal into a logic HIGH or LOW output signal, respectively, an inverter is generally operated in its saturated region, and will only incidentally pass through its linear region as its output switches between logic LOW (GND) and logic HIGH (VDD). - However, the linear biasing provided by bias circuits 220(A) and 220(B) forces inverters 210(A) and 210(B), respectively, to operate in their linear regions, so that inverters 210(A) and 210(B) can be used to provide signal amplification. Specifically, the DC bias voltages supplied by bias circuits 220(A) and 220(B) force the nominal inverter output voltages (i.e., the voltages at the outputs of the inverters when no AC signal is present) for inverters 210(A) and 210(B) to levels between upper supply voltage VDD and lower supply voltage VSS.
- The outputs of inverters 210(A) and 210(B) will therefore swing around this nominal inverter output voltage, thereby ensuring that inverters 210(A) and 210(B) provide AC output signals that are proportional to their AC input signals (so long as the AC input signal amplitude does not push inverters 210(A) and 210(B) into saturation). By setting the nominal inverter output voltage equal to half of the difference between upper supply voltage VDD and lower supply voltage VSS (e.g., if supply voltage VSS is ground, then the nominal inverter output voltage would be VDD/2), the total output swing of
differential amplifier 200 can be maximized (i.e., output swing equal to 2*VDD). Note that because inverters 210(A) and 210(B) do not include any resistive elements, this increased gain does not result in output signal distortion (unlike the results described with respect to conventionaldifferential amplifier 100 shown inFIG. 1 ). - Returning to
FIG. 2A , according to an embodiment of the invention, bias circuit 220(A) includes resistors R_IN(A) and R_OUT(A), optional capacitors C221(A) and C222(A), and an operational amplifier (op-amp) 240(A). Resistor R_IN(A) is connected between the input of inverter 210(A) and the output of op-amp 240(A), while resistor R_OUT(A) is connected between the output of inverter 210(A) and the non-inverting input of op-amp 240(A). Capacitor C221(A) is connected between the output of op-amp 240(A) and ground, while capacitor C222(A) is connected between the non-inverting input of op-amp 240(A) and ground. - Similarly, bias circuit 220(B) includes resistors R_IN(B) and R_OUT(B), optional capacitors C221(B) and C222(B), and an operational amplifier (op-amp) 240(B). Resistor R_IN(B) is connected between the input of inverter 210(B) and the output of op-amp 240(B), while resistor R_OUT(B) is connected between the output of inverter 210(B) and the non-inverting input of op-amp 240(B). Capacitor C221(B) is connected between the output of op-amp 240(B) and ground, while capacitor C222(B) is connected between the non-inverting input of op-amp 240(B) and ground.
-
Reference voltage source 230 provides a reference voltage V_MID to the inverting inputs of op-amps 240(A) and 240(B). Meanwhile, the non-inverting inputs of op-amps 240(A) and 240(B) receive the outputs of inverters 210(A) and 210(B), respectively (via resistors R_OUT(A) and R_OUT(B), respectively). If the voltage at the output of inverter 210(A) is less than reference voltage V_MID, op-amp 240(A) decreases its output voltage (and hence the voltage provided at the input of inverter 210(A) via resistor R_IN(A)), thereby raising the output of inverter 210(A). Likewise, if the voltage at the output of inverter 210(A) is greater than reference voltage V_MID, op-amp 240(A) increases its output voltage to decrease the output of inverter 210(A). Op-amp 240(B) regulates the output of inverter 210(B) in a similar manner. - In this manner, op-amps 240(A) and 240(B) create DC bias voltages at the inputs of inverters 210(A) and 210(B), respectively, such that each inverter has a DC offset voltage at its output that is equal to reference voltage V_MID. This DC biasing of the inverter inputs forces inverters 210(A) and 210(B) to operate in the linear mode, so that gain can be applied to signals provided to inverters 210(A) and 210(B) without distortion (clipping). Note that, while reference voltage V_MID can be set to any value between supply voltage VDD and ground (the upper and lower supply voltages), the maximum output range of
amplifier circuit 200 will be provided by setting reference voltage V_MID halfway between supply voltage VDD and ground (i.e., V_MID=VDD/2). - Note further, that it is desirable that the linear biasing provided by bias circuits 220(A) and 220(B) not be affected by (or affect) the AC signal being amplified by
amplifier circuit 200. Accordingly, resistors R_IN(A) and R_OUT(A) isolate op-amp 240(A) from any AC signals that are provided to or generated by inverter 210(A) by suppressing the bulk of those signals before they reach op-amp 240(A). Meanwhile, optional capacitors C221(A) and C222(A) can provide a direct path to ground for any AC that does get by resistors R_IN(A) and R_OUT(A), respectively, or is generated by op-amp 240(A). In a similar manner, resistors R_IN(B) and R_OUT(B) and capacitors C221(A) and C222(B) provide AC isolation for op-amp 240(B). - Practitioners will readily appreciate that because bias circuits 220(A) and 220(B) do not include constant bias currents (e.g., currents I_BIAS-1 and I_BIAS-2 shown in
FIG. 1 ) flowing through large resistive elements (e.g., resistors RD(A) and/or RD(B) shown inFIG. 1 ), the power consumption ofamplifier circuit 200 shown inFIG. 2A can be significantly less than the power consumption ofconventional amplifier 100. - Furthermore, because of the linear biasing provided by bias circuits 220(A) and 220(B), inverters 210(A) and 210(B) can both provide a significant amount of gain (while operating in their linear regions). For example, the actual gain G provided by inverter 210(A) is given by the following equation:
G=(g m1 +g m2)*(Ro 1 ||Ro 2) (6)
where gm1 and gm2 are the transconductances of transistors M1(A) and M2 (A), respectively, and Ro1 and Ro2 are the output resistances of transistors M1(A) and M2(A), respectively. - The term “Ro1||Ro2” represents the parallel resistance of Ro1 and Ro2, and resolves to the equation:
Ro l ||Ro 2=(Ro 1 *Ro 2)/(Ro 1 +Ro 2) (7)
Substituting equation (7) into equation (6) therefore yields a gain equation of:
G=(g m1 +g m2)/(Y 1 +Y 2) (8)
where Y1 is equal to 1/Ro1 and Y2 is equal to 1/Ro2. - Note that if transconductances gm1 and gm2 are equal, and if output resistances Ro1 and Ro2 are equal, equation 8 resolves to the following:
G=g m *Ro (9)
where gm=gm1=gm2, and Ro=Ro1=Ro2. Gain G is therefore proportional to transconductance gm and output resistance Ro. - MOS transconductance gm is given by the following:
where kp is the intrinsic transconductance parameter for the MOS transistor, w/l is the aspect ratio of the transistor, and ID is the drain current. Meanwhile, output resistance Ro is given by the following:
where λ is the channel length modulation parameter for the transistor. Therefore, by substituting equations 10 and 11 into equation 9, gain G can be expressed by the following:
Thus, as indicated by equation 12, the gain provided by an inverter-based differential amplifier such as shown inFIG. 2A is inversely proportional to drain current, and is therefore not subject to the output distortion associated with common-source basedamplifier 100 shown inFIG. 1 . - As indicated by equation 10, in a MOS transistor, the transconductance is proportional to the aspect ratio (width/length) of the gate. Therefore, by adjusting the gate dimensions of transistors M1(A) and M2(A), the gain provided by branch 200(A) of
amplifier circuit 200 can be adjusted. For similar reasons, by adjusting the gate dimensions of transistors M1(B) and M2(B), the gain provided by branch 200(B) can be adjusted. - For example, according to an embodiment of the invention, supply voltage VDD can be 1.8V, reference voltage V_MID can be set to 0.9V, transistors M1(A) and M1(B) can have aspect ratios of 27/0.35, transistors M2(A) and M2(B) can have aspect ratios of 21.6/0.35, resistors R_IN(A), R_OUT(A), R_IN(B), and R_OUT(B) can have resistances of 1.5 kΩ each, and capacitors C_IN(A), C_OUT(A), C_IN(B), and C_OUT(B) can have capacitances of 150 fF each. Branches 200(A) and 200(B) would then provide between 10-15 dB of RF gain each.
- Note that while branches 200(A) and 200(B) shown in
FIG. 2A are described as single stages for exemplary purposes, each of branches 200(A) and 200(B) can comprise a stage in a series of cascaded amplifier stages, or a predriver for additional amplifier circuitry, as indicated by optional (dotted line) amplifier stage circuitry 290(A) and 290(B). -
FIG. 3 shows a detailed view of branch 200(A) that depicts a schematic diagram for op-amp 240(A), according to an embodiment of the invention. (A similar op-amp circuit could be used for op-amp 240(B) inFIG. 2A .) Op-amp 240(A) includes PMOS transistors M3 and M5, NMOS transistors M4, M6, M7, and M8, acurrent source 241, a capacitor C_CP, and a resistor R_CP. - Transistors M3 and M4 are connected in series between supply voltage VDD and transistor M8, and transistors M5 and M6 are connected in series between supply voltage VDD and transistor M8. Transistor M8 is coupled between transistor M4 and ground, and
current source 241 and transistor M7 are connected in series between supply voltage VDD and ground. Finally, capacitor C_CP and resistor R_CP are connected in series between the gate of transistor M4 and the drain of transistor M6. - The gate of transistor M4 forms the non-inverting input of op-amp 240(A), and is accordingly coupled to the input of inverter 210(A) via resistor R_OUT(A). Meanwhile, the gate of transistor M6 forms the inverting input of op-amp 240(A), and is therefore coupled to reference voltage circuit 230(A). And the junction between transistors M5 and M6 forms the output of op-amp 240(A), and is therefore coupled to the input of inverter 210(A) via resistor R_IN(A).
- Thus, capacitor C_CP and resistor R_CP are coupled between the non-inverting input and the output of op-amp 240(A). Capacitor C_CP and resistor R_CP form a compensation circuit that improves the stability of op-amp 240(A) by preventing unwanted oscillations. Note that various other op-amp compensation circuits will be readily apparent.
- The gate and drain of transistor M7 are shorted, and the gates of transistors M7 and M8 are connected to form a current mirror. Therefore, a current I_BIAS from
current source 241 that is sunk by transistor M7 is also mirrored by transistor M8. Therefore, a total current I_BIAS flows through the two branches formed by transistors M3 and M4 (first branch) and by transistors M5 and M6 (second branch). - Meanwhile, the gate and drain of transistor M3 are shorted, and the gates of transistors M3 and M5 are connected to form another current mirror that provides a load for the differential pair formed by transistors M4 and M6. When the gate voltages provided to transistors M4 and M6 (i.e., the inputs to op-amp 240(A)) are the same, transistors M3 and M5 split the flow of current I_BIAS equally through transistors M4 and M6. However, when the gate voltages of transistors M4 and M6 are different, transistor M5 adjusts its drain voltage (i.e., the output of op-amp 240(A)) in response.
- For example, if the voltage provided at the gate of transistor M4 (i.e., the voltage at the output of inverter 210(A)) is greater than the voltage provided at the gate of transistor M6 (i.e., reference voltage V_MID), then transistor M4 is turned on more strongly than transistor M6, and the current flow through transistor M4 increases. Since the total current flow through transistors M4 and M6 is fixed at current I_BIAS by transistor M8, this increase in current flow through transistor M4 means that the current flow through transistor M6 must decrease.
- To provide this current reduction, the drain voltage of transistor M6 is increased. This has the effect of reducing the gate-drain voltage of transistor M6, which in turn reduces the current flow through transistor M6. Meanwhile, this increased drain voltage of transistor M6 is applied to the input of inverter 210(A) (via resistor R_IN(A)), thereby driving the voltage at the output of inverter 210(A) down towards reference voltage V_MID.
- Similarly, if the voltage provided at the gate of transistor M4 is less than the voltage provided at the gate of transistor M6, then transistor M4 is turned on less strongly than transistor M6, and the current flow through transistor M4 decreases. Therefore, the current flow through transistor M6 must increase, and the drain voltage of transistor M6 is decreased to increase the gate-drain voltage of transistor M6. This decreased drain voltage of transistor M6 is applied to the input of inverter 210(A), thereby driving the voltage at the output of inverter 210(A) up towards reference voltage V_MID.
- Of course, the circuitry shown for op-amp 240(A) in
FIG. 3 is exemplary only. Alternatives may be found in the conventional art. - The various embodiments of the structures and methods of this invention that are described above are illustrative only of the principles of this invention and are not intended to limit the scope of the invention to the particular embodiments described. For example, capacitors C_IN(A) and C_IN(B) could be removed from
differential amplifier 200 inFIG. 2A , thereby enabling amplification of DC input voltages at input terminals 201(A) and 201(B). Thus, the invention is limited only by the following claims and their equivalents.
Claims (25)
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US10/802,894 US6937071B1 (en) | 2004-03-16 | 2004-03-16 | High frequency differential power amplifier |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US10/802,894 US6937071B1 (en) | 2004-03-16 | 2004-03-16 | High frequency differential power amplifier |
Publications (2)
Publication Number | Publication Date |
---|---|
US6937071B1 US6937071B1 (en) | 2005-08-30 |
US20050206412A1 true US20050206412A1 (en) | 2005-09-22 |
Family
ID=34862005
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US10/802,894 Expired - Lifetime US6937071B1 (en) | 2004-03-16 | 2004-03-16 | High frequency differential power amplifier |
Country Status (1)
Country | Link |
---|---|
US (1) | US6937071B1 (en) |
Cited By (10)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20110150245A1 (en) * | 2009-12-23 | 2011-06-23 | Stmicroelectronics Design And Application S.R.O. | Capacitive load driving amplifier |
WO2013043950A1 (en) * | 2011-09-23 | 2013-03-28 | Tensorcom, Inc. | A DIFFERENTIAL SOURCE FOLLOWER HAVING 6dB GAIN WITH APPLICATIONS TO WiGig BASEBAND FILTERS |
US8487695B2 (en) | 2011-09-23 | 2013-07-16 | Tensorcom, Inc. | Differential source follower having 6dB gain with applications to WiGig baseband filters |
US8680899B2 (en) | 2011-09-23 | 2014-03-25 | Tensorcom, Inc. | High performance divider using feed forward, clock amplification and series peaking inductors |
US9893692B2 (en) | 2015-11-17 | 2018-02-13 | Tensorcom, Inc. | High linearly WiGig baseband amplifier with channel select filter |
US10218310B2 (en) | 2016-09-09 | 2019-02-26 | Skyworks Solutions, Inc. | Power amplifier systems with differential ground |
WO2020097419A1 (en) * | 2018-11-09 | 2020-05-14 | Butterfly Network, Inc. | Trans-impedance amplifier (tia) for ultrasound devices |
US11128267B2 (en) | 2015-12-02 | 2021-09-21 | Bfly Operations, Inc. | Trans-impedance amplifier for ultrasound device and related apparatus and methods |
US11169248B2 (en) | 2015-12-02 | 2021-11-09 | Bfly Operations, Inc. | Multi-level pulser and related apparatus and methods |
US11215703B2 (en) | 2015-12-02 | 2022-01-04 | Bfly Operations, Inc. | Time gain compensation circuit and related apparatus and methods |
Families Citing this family (4)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7245179B2 (en) * | 2004-10-04 | 2007-07-17 | Industrial Technology Research Institute | Auto gain controller |
WO2011069231A1 (en) * | 2009-12-11 | 2011-06-16 | Ess Technology, Inc. | No load amplifier |
US8872592B2 (en) * | 2012-06-19 | 2014-10-28 | Taiwan Semiconductor Manufacturing Company, Ltd. | High-speed transimpedance amplifier |
EP2779456B1 (en) * | 2013-03-15 | 2018-08-29 | Dialog Semiconductor B.V. | Method for reducing overdrive need in mos switching and logic circuit |
Citations (51)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3947779A (en) * | 1974-12-11 | 1976-03-30 | The Birtcher Corporation | Input bias and signal conditioning circuit with overload indication for differential amplifiers |
US3982233A (en) * | 1974-02-19 | 1976-09-21 | Ampex Corporation | Core memory with improved sense-inhibit recovery time |
US3982826A (en) * | 1973-12-17 | 1976-09-28 | Nihon Beru-Haueru Kabushiki Kaisha (Bell & Howell Japan, Ltd.) | Cine camera |
US4015117A (en) * | 1975-08-28 | 1977-03-29 | Opcon, Inc. | Unbiased modulated photo sensing systems |
US4024462A (en) * | 1975-05-27 | 1977-05-17 | International Business Machines Corporation | Darlington configuration high frequency differential amplifier with zero offset current |
US4206316A (en) * | 1976-05-24 | 1980-06-03 | Hughes Aircraft Company | Transmitter-receiver system utilizing pulse position modulation and pulse compression |
US4274039A (en) * | 1980-02-25 | 1981-06-16 | Fluid Data Systems | Servo-amplifier circuit |
US4331157A (en) * | 1980-07-09 | 1982-05-25 | Stimtech, Inc. | Mutually noninterfering transcutaneous nerve stimulation and patient monitoring |
US4482866A (en) * | 1982-02-26 | 1984-11-13 | Barcus-Berry, Inc. | Reference load amplifier correction system |
US4517523A (en) * | 1982-10-22 | 1985-05-14 | Tektronix, Inc. | Oscilloscope trigger circuit combining wideband trigger signal amplification and trigger coupling mode selection |
US4528515A (en) * | 1983-02-07 | 1985-07-09 | Tektronix, Inc. | High frequency differential amplifier with adjustable damping factor |
US4638258A (en) * | 1982-02-26 | 1987-01-20 | Barcus-Berry Electronics, Inc. | Reference load amplifier correction system |
US4698842A (en) * | 1985-07-11 | 1987-10-06 | Electronic Engineering And Manufacturing, Inc. | Audio processing system for restoring bass frequencies |
US4769616A (en) * | 1985-12-31 | 1988-09-06 | U.S. Philips Corporation | High-frequency differential amplifier stage and amplifier comprising such a differential amplifier stage |
US4825173A (en) * | 1988-01-18 | 1989-04-25 | Motorola, Inc. | High gain, programmable differential amplifier circuitry |
US4847831A (en) * | 1987-03-30 | 1989-07-11 | Honeywell Inc. | Bidirectional repeater for manchester encoded data signals |
US4942399A (en) * | 1989-03-15 | 1990-07-17 | International Business Machines Corporation | Adaptive flash analog/digital converter for differential input signal |
US5025456A (en) * | 1989-02-02 | 1991-06-18 | At&T Bell Laboratories | Burst mode digital data receiver |
US5148119A (en) * | 1991-03-22 | 1992-09-15 | Linear Technology Corporation | Precise reference voltage generator for feedforward compensated amplifiers |
US5220164A (en) * | 1992-02-05 | 1993-06-15 | General Atomics | Integrated imaging and ranging lidar receiver with ranging information pickoff circuit |
US5283514A (en) * | 1992-09-08 | 1994-02-01 | Hybricon Corporation | Fast response current regulator for DC power supply |
US5349644A (en) * | 1992-06-30 | 1994-09-20 | Electronic Innovators, Inc. | Distributed intelligence engineering casualty and damage control management system using an AC power line carrier-current lan |
US5357209A (en) * | 1991-05-15 | 1994-10-18 | Telefonaktiebolaget L M Ericsson | Limiting amplifier |
US5426542A (en) * | 1994-01-21 | 1995-06-20 | Seagate Technology, Inc. | Electronically coupled high-impedance magnetoresistive preamplifier |
US5485433A (en) * | 1991-12-19 | 1996-01-16 | Canon Kabushiki Kaisha | Information recording method and apparatus for determining whether recording has been correctly performed |
US5550513A (en) * | 1995-06-22 | 1996-08-27 | Northern Telecom Limited | High frequency, differential limiting distributed amplifier |
US5568099A (en) * | 1995-09-27 | 1996-10-22 | Cirrus Logic, Inc. | High frequency differential VCO with common biased clipper |
US5608328A (en) * | 1994-11-18 | 1997-03-04 | Radar Engineers | Method and apparatus for pin-pointing faults in electric power lines |
US5648743A (en) * | 1993-08-10 | 1997-07-15 | Fujitsu Limited | Amplifying circuit for an integrated circuit with low-noise characteristic |
US5708391A (en) * | 1996-05-02 | 1998-01-13 | Altmann; Michael | High frequency differential filter with CMOS control |
US5764110A (en) * | 1996-07-15 | 1998-06-09 | Mitsubishi Denki Kabushiki Kaisha | Voltage controlled ring oscillator stabilized against supply voltage fluctuations |
US5945883A (en) * | 1996-07-15 | 1999-08-31 | Mitsubishi Denki Kabushiki Kaisha | Voltage controlled ring oscillator stabilized against supply voltage fluctuations |
US6061702A (en) * | 1996-05-15 | 2000-05-09 | Intel Corporation | Random number generator |
US6160578A (en) * | 1998-06-18 | 2000-12-12 | Redlake Imaging Corporation | High speed, increased bandwidth camera |
US6163219A (en) * | 1998-09-17 | 2000-12-19 | Fujitsu Limited | Amplification circuit and integrated circuit having such and controlling method of the amplification circuit |
US6169462B1 (en) * | 1999-07-14 | 2001-01-02 | Thomson Licensing S.A. | Oscillator with controlled current source for start stop control |
US6175885B1 (en) * | 1996-11-19 | 2001-01-16 | Sgs-Microelectronics S.A. | System for series to parallel conversion of a low-amplitude and high frequency signal |
US6177832B1 (en) * | 1998-07-06 | 2001-01-23 | Motorola, Inc. | High frequency differential to single-ended converter |
US6296116B1 (en) * | 1999-08-26 | 2001-10-02 | Karsten Manufacturing Corporation | Golf bag with cam actuated support stand and detachable body |
US6392486B1 (en) * | 2001-08-14 | 2002-05-21 | Xilinx, Inc. | Low-noise common-gate amplifier for wireless communications |
US6411151B1 (en) * | 1999-12-13 | 2002-06-25 | Inter Corporation | Low jitter external clocking |
US6545502B1 (en) * | 2001-11-09 | 2003-04-08 | Institute Of Microelectronics | High frequency MOS fixed and variable gain amplifiers |
US6563377B2 (en) * | 2001-10-09 | 2003-05-13 | Evenstar, Inc. | Class D switching audio amplifier |
US6618305B2 (en) * | 2001-05-02 | 2003-09-09 | Infineon Technologies Ag | Test circuit for testing a circuit |
US6624697B2 (en) * | 2001-01-12 | 2003-09-23 | Jennic Limited | High frequency differential amplifier |
US6643324B1 (en) * | 2000-05-08 | 2003-11-04 | Lsi Logic Corporation | Pad cell circuit-integrated, differential-signal equalization receiver for integrated circuit and method of boosting and equalizing high frequency differential signals |
US6654900B1 (en) * | 2000-04-19 | 2003-11-25 | Sigmatel, Inc. | Method and apparatus for producing multiple clock signals having controlled duty cycles by controlling clock multiplier delay elements |
US6664609B2 (en) * | 2001-03-26 | 2003-12-16 | Matsushita Electric Industrial Co., Ltd. | High frequency differential amplification circuit with reduced parasitic capacitance |
US6693457B2 (en) * | 2002-07-18 | 2004-02-17 | Northrop Grumman Corporation | Ultra high speed flip-flop |
US6737924B1 (en) * | 2002-12-11 | 2004-05-18 | Intel Corporation | Differential, double feedback CMOS transimpedance amplifier with noise tolerance |
US6798251B1 (en) * | 2002-08-13 | 2004-09-28 | Analog Devices, Inc. | Differential clock receiver with adjustable output crossing point |
Family Cites Families (1)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6292116B1 (en) | 1999-05-17 | 2001-09-18 | Altera Corporation | Techniques and circuitry for accurately sampling high frequency data signals input to an integrated circuit |
-
2004
- 2004-03-16 US US10/802,894 patent/US6937071B1/en not_active Expired - Lifetime
Patent Citations (54)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3982826A (en) * | 1973-12-17 | 1976-09-28 | Nihon Beru-Haueru Kabushiki Kaisha (Bell & Howell Japan, Ltd.) | Cine camera |
US3982233A (en) * | 1974-02-19 | 1976-09-21 | Ampex Corporation | Core memory with improved sense-inhibit recovery time |
US3947779A (en) * | 1974-12-11 | 1976-03-30 | The Birtcher Corporation | Input bias and signal conditioning circuit with overload indication for differential amplifiers |
US4024462A (en) * | 1975-05-27 | 1977-05-17 | International Business Machines Corporation | Darlington configuration high frequency differential amplifier with zero offset current |
US4015117A (en) * | 1975-08-28 | 1977-03-29 | Opcon, Inc. | Unbiased modulated photo sensing systems |
US4065724A (en) * | 1975-08-28 | 1977-12-27 | Opcon, Inc. | Balanced low impedance differential input line preamplifier |
US4206316A (en) * | 1976-05-24 | 1980-06-03 | Hughes Aircraft Company | Transmitter-receiver system utilizing pulse position modulation and pulse compression |
US4274039A (en) * | 1980-02-25 | 1981-06-16 | Fluid Data Systems | Servo-amplifier circuit |
US4331157A (en) * | 1980-07-09 | 1982-05-25 | Stimtech, Inc. | Mutually noninterfering transcutaneous nerve stimulation and patient monitoring |
US4482866A (en) * | 1982-02-26 | 1984-11-13 | Barcus-Berry, Inc. | Reference load amplifier correction system |
US4638258A (en) * | 1982-02-26 | 1987-01-20 | Barcus-Berry Electronics, Inc. | Reference load amplifier correction system |
US4517523A (en) * | 1982-10-22 | 1985-05-14 | Tektronix, Inc. | Oscilloscope trigger circuit combining wideband trigger signal amplification and trigger coupling mode selection |
US4528515A (en) * | 1983-02-07 | 1985-07-09 | Tektronix, Inc. | High frequency differential amplifier with adjustable damping factor |
US4698842A (en) * | 1985-07-11 | 1987-10-06 | Electronic Engineering And Manufacturing, Inc. | Audio processing system for restoring bass frequencies |
US4769616A (en) * | 1985-12-31 | 1988-09-06 | U.S. Philips Corporation | High-frequency differential amplifier stage and amplifier comprising such a differential amplifier stage |
US4847831A (en) * | 1987-03-30 | 1989-07-11 | Honeywell Inc. | Bidirectional repeater for manchester encoded data signals |
US4825173A (en) * | 1988-01-18 | 1989-04-25 | Motorola, Inc. | High gain, programmable differential amplifier circuitry |
US5025456A (en) * | 1989-02-02 | 1991-06-18 | At&T Bell Laboratories | Burst mode digital data receiver |
US4942399A (en) * | 1989-03-15 | 1990-07-17 | International Business Machines Corporation | Adaptive flash analog/digital converter for differential input signal |
US5148119A (en) * | 1991-03-22 | 1992-09-15 | Linear Technology Corporation | Precise reference voltage generator for feedforward compensated amplifiers |
US5357209A (en) * | 1991-05-15 | 1994-10-18 | Telefonaktiebolaget L M Ericsson | Limiting amplifier |
US5485433A (en) * | 1991-12-19 | 1996-01-16 | Canon Kabushiki Kaisha | Information recording method and apparatus for determining whether recording has been correctly performed |
US5220164A (en) * | 1992-02-05 | 1993-06-15 | General Atomics | Integrated imaging and ranging lidar receiver with ranging information pickoff circuit |
US5349644A (en) * | 1992-06-30 | 1994-09-20 | Electronic Innovators, Inc. | Distributed intelligence engineering casualty and damage control management system using an AC power line carrier-current lan |
US5283514A (en) * | 1992-09-08 | 1994-02-01 | Hybricon Corporation | Fast response current regulator for DC power supply |
US5648743A (en) * | 1993-08-10 | 1997-07-15 | Fujitsu Limited | Amplifying circuit for an integrated circuit with low-noise characteristic |
US5734298A (en) * | 1993-08-10 | 1998-03-31 | Fujitsu Limited | FET amplifying circuit which can improve low-consumptive current |
US5426542A (en) * | 1994-01-21 | 1995-06-20 | Seagate Technology, Inc. | Electronically coupled high-impedance magnetoresistive preamplifier |
US5608328A (en) * | 1994-11-18 | 1997-03-04 | Radar Engineers | Method and apparatus for pin-pointing faults in electric power lines |
US5550513A (en) * | 1995-06-22 | 1996-08-27 | Northern Telecom Limited | High frequency, differential limiting distributed amplifier |
US5638030A (en) * | 1995-09-27 | 1997-06-10 | Cirrus Logic, Inc. | High frequency differential VCO with common biased clipper |
US5568099A (en) * | 1995-09-27 | 1996-10-22 | Cirrus Logic, Inc. | High frequency differential VCO with common biased clipper |
US5708391A (en) * | 1996-05-02 | 1998-01-13 | Altmann; Michael | High frequency differential filter with CMOS control |
US6061702A (en) * | 1996-05-15 | 2000-05-09 | Intel Corporation | Random number generator |
US5764110A (en) * | 1996-07-15 | 1998-06-09 | Mitsubishi Denki Kabushiki Kaisha | Voltage controlled ring oscillator stabilized against supply voltage fluctuations |
US5945883A (en) * | 1996-07-15 | 1999-08-31 | Mitsubishi Denki Kabushiki Kaisha | Voltage controlled ring oscillator stabilized against supply voltage fluctuations |
US6175885B1 (en) * | 1996-11-19 | 2001-01-16 | Sgs-Microelectronics S.A. | System for series to parallel conversion of a low-amplitude and high frequency signal |
US6160578A (en) * | 1998-06-18 | 2000-12-12 | Redlake Imaging Corporation | High speed, increased bandwidth camera |
US6177832B1 (en) * | 1998-07-06 | 2001-01-23 | Motorola, Inc. | High frequency differential to single-ended converter |
US6163219A (en) * | 1998-09-17 | 2000-12-19 | Fujitsu Limited | Amplification circuit and integrated circuit having such and controlling method of the amplification circuit |
US6169462B1 (en) * | 1999-07-14 | 2001-01-02 | Thomson Licensing S.A. | Oscillator with controlled current source for start stop control |
US6296116B1 (en) * | 1999-08-26 | 2001-10-02 | Karsten Manufacturing Corporation | Golf bag with cam actuated support stand and detachable body |
US6411151B1 (en) * | 1999-12-13 | 2002-06-25 | Inter Corporation | Low jitter external clocking |
US6654900B1 (en) * | 2000-04-19 | 2003-11-25 | Sigmatel, Inc. | Method and apparatus for producing multiple clock signals having controlled duty cycles by controlling clock multiplier delay elements |
US6643324B1 (en) * | 2000-05-08 | 2003-11-04 | Lsi Logic Corporation | Pad cell circuit-integrated, differential-signal equalization receiver for integrated circuit and method of boosting and equalizing high frequency differential signals |
US6624697B2 (en) * | 2001-01-12 | 2003-09-23 | Jennic Limited | High frequency differential amplifier |
US6664609B2 (en) * | 2001-03-26 | 2003-12-16 | Matsushita Electric Industrial Co., Ltd. | High frequency differential amplification circuit with reduced parasitic capacitance |
US6618305B2 (en) * | 2001-05-02 | 2003-09-09 | Infineon Technologies Ag | Test circuit for testing a circuit |
US6392486B1 (en) * | 2001-08-14 | 2002-05-21 | Xilinx, Inc. | Low-noise common-gate amplifier for wireless communications |
US6563377B2 (en) * | 2001-10-09 | 2003-05-13 | Evenstar, Inc. | Class D switching audio amplifier |
US6545502B1 (en) * | 2001-11-09 | 2003-04-08 | Institute Of Microelectronics | High frequency MOS fixed and variable gain amplifiers |
US6693457B2 (en) * | 2002-07-18 | 2004-02-17 | Northrop Grumman Corporation | Ultra high speed flip-flop |
US6798251B1 (en) * | 2002-08-13 | 2004-09-28 | Analog Devices, Inc. | Differential clock receiver with adjustable output crossing point |
US6737924B1 (en) * | 2002-12-11 | 2004-05-18 | Intel Corporation | Differential, double feedback CMOS transimpedance amplifier with noise tolerance |
Cited By (19)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US20110150245A1 (en) * | 2009-12-23 | 2011-06-23 | Stmicroelectronics Design And Application S.R.O. | Capacitive load driving amplifier |
US8897467B2 (en) * | 2009-12-23 | 2014-11-25 | Stmicroelectronics Design And Application S.R.O. | Capacitive load driving amplifier |
WO2013043950A1 (en) * | 2011-09-23 | 2013-03-28 | Tensorcom, Inc. | A DIFFERENTIAL SOURCE FOLLOWER HAVING 6dB GAIN WITH APPLICATIONS TO WiGig BASEBAND FILTERS |
US8487695B2 (en) | 2011-09-23 | 2013-07-16 | Tensorcom, Inc. | Differential source follower having 6dB gain with applications to WiGig baseband filters |
US8674755B2 (en) | 2011-09-23 | 2014-03-18 | Tensorcom, Inc. | Differential source follower having 6dB gain with applications to WiGig baseband filters |
US8680899B2 (en) | 2011-09-23 | 2014-03-25 | Tensorcom, Inc. | High performance divider using feed forward, clock amplification and series peaking inductors |
US8803596B2 (en) | 2011-09-23 | 2014-08-12 | Tensorcom, Inc. | Differential source follower having 6dB gain with applications to WiGig baseband filters |
US10734957B2 (en) | 2015-11-17 | 2020-08-04 | Tensorcom, Inc. | High linearly WiGig baseband amplifier with channel select filter |
US10277182B2 (en) | 2015-11-17 | 2019-04-30 | Tensorcom, Inc. | High linearly WiGig baseband amplifier with channel select filter |
US9893692B2 (en) | 2015-11-17 | 2018-02-13 | Tensorcom, Inc. | High linearly WiGig baseband amplifier with channel select filter |
US11128267B2 (en) | 2015-12-02 | 2021-09-21 | Bfly Operations, Inc. | Trans-impedance amplifier for ultrasound device and related apparatus and methods |
US11169248B2 (en) | 2015-12-02 | 2021-11-09 | Bfly Operations, Inc. | Multi-level pulser and related apparatus and methods |
US11215703B2 (en) | 2015-12-02 | 2022-01-04 | Bfly Operations, Inc. | Time gain compensation circuit and related apparatus and methods |
US11573309B2 (en) | 2015-12-02 | 2023-02-07 | Bfly Operations, Inc. | Time gain compensation circuit and related apparatus and methods |
US11863133B2 (en) | 2015-12-02 | 2024-01-02 | Bfly Operations, Inc. | Trans-impedance amplifier for ultrasound device and related apparatus and methods |
US10218310B2 (en) | 2016-09-09 | 2019-02-26 | Skyworks Solutions, Inc. | Power amplifier systems with differential ground |
US10658977B2 (en) | 2016-09-09 | 2020-05-19 | Skyworks Solutions, Inc. | Power amplifiers isolated by differential ground |
WO2020097419A1 (en) * | 2018-11-09 | 2020-05-14 | Butterfly Network, Inc. | Trans-impedance amplifier (tia) for ultrasound devices |
US11662447B2 (en) | 2018-11-09 | 2023-05-30 | Bfly Operations, Inc. | Trans-impedance amplifier (TIA) for ultrasound devices |
Also Published As
Publication number | Publication date |
---|---|
US6937071B1 (en) | 2005-08-30 |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US8063702B2 (en) | High frequency receiver preamplifier with CMOS rail-to-rail capability | |
US7750738B2 (en) | Process, voltage and temperature control for high-speed, low-power fixed and variable gain amplifiers based on MOSFET resistors | |
EP3437187B1 (en) | System and method for controlling common mode voltage via replica circuit and feedback control | |
US8410854B2 (en) | Semiconductor integrated circuit device | |
US6002276A (en) | Stable output bias current circuitry and method for low-impedance CMOS output stage | |
US6937071B1 (en) | High frequency differential power amplifier | |
US20020158686A1 (en) | Linear voltage subtractor/adder circuit and MOS differential amplifier circuit therefor | |
US5343164A (en) | Operational amplifier circuit with slew rate enhancement | |
US5381112A (en) | Fully differential line driver circuit having common-mode feedback | |
US20100066449A1 (en) | Three-stage frequency-compensated operational amplifier for driving large capacitive loads | |
US20070013438A1 (en) | Chopper-stabilized operational amplifier and method | |
US7391201B2 (en) | Regulated analog switch | |
US7378908B2 (en) | Variable gain differential amplifier, and variable degeneration impedance control device and method for use in the same | |
US7068090B2 (en) | Amplifier circuit | |
KR100468358B1 (en) | Variable Gain Amplifier Having Improved Gain Slope Characteristic | |
EP2154783A1 (en) | Amplifying circuit | |
US6985038B2 (en) | Operational amplifier generating desired feedback reference voltage allowing improved output characteristic | |
US6940353B2 (en) | High frequency power amplifier | |
US6946907B2 (en) | Common mode feedback amplifier | |
US7002405B2 (en) | Linear low noise transconductance cell | |
US6717470B1 (en) | Voltage amplifier with output stages having high capacitive load tolerance | |
US6188284B1 (en) | Distributed gain line driver amplifier including improved linearity | |
US7098702B2 (en) | Transconductor circuit for compensating the distortion of output current | |
US7193468B2 (en) | Active load circuit for low-voltage CMOS voltage gain amplifier with wide bandwidth and high gain characteristic | |
KR100668455B1 (en) | Variable gain amplifier |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
AS | Assignment |
Owner name: MICREL, INCORPORATED, CALIFORNIA Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:MORAVEJI, FARHOOD;REEL/FRAME:015118/0574 Effective date: 20040316 |
|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |
|
FPAY | Fee payment |
Year of fee payment: 4 |
|
FPAY | Fee payment |
Year of fee payment: 8 |
|
FPAY | Fee payment |
Year of fee payment: 12 |
|
AS | Assignment |
Owner name: MICROCHIP TECHNOLOGY INCORPORATED, ARIZONA Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:MICREL LLC;REEL/FRAME:051291/0211 Effective date: 20191213 Owner name: MICREL LLC, CALIFORNIA Free format text: MERGER;ASSIGNOR:MICREL, INCORPORATED;REEL/FRAME:051295/0334 Effective date: 20150803 |
|
AS | Assignment |
Owner name: MICREL LLC, CALIFORNIA Free format text: CORRECTIVE ASSIGNMENT TO CORRECT THE CORRECT PATENT NUMBER 6876244 PREVIOUSLY RECORDED AT REEL: 051295 FRAME: 0334. ASSIGNOR(S) HEREBY CONFIRMS THE ASSIGNMENT;ASSIGNOR:MICREL, INCORPORATED;REEL/FRAME:051345/0367 Effective date: 20150803 |
|
AS | Assignment |
Owner name: MICROCHIP TECHNOLOGY INCORPORATED, ARIZONA Free format text: RELEASE BY SECURED PARTY;ASSIGNOR:JPMORGAN CHASE BANK, N.A., AS ADMINISTRATIVE AGENT;REEL/FRAME:051398/0809 Effective date: 20191220 Owner name: ATMEL CORPORATION, ARIZONA Free format text: RELEASE BY SECURED PARTY;ASSIGNOR:JPMORGAN CHASE BANK, N.A., AS ADMINISTRATIVE AGENT;REEL/FRAME:051398/0809 Effective date: 20191220 Owner name: MICROCHIP TECHNOLOGY INCORPORATED, ARIZONA Free format text: RELEASE BY SECURED PARTY;ASSIGNOR:WELLS FARGO BANK, NATIONAL ASSOCIATION, AS NOTES COLLATERAL AGENT;REEL/FRAME:051398/0827 Effective date: 20191220 Owner name: ATMEL CORPORATION, ARIZONA Free format text: RELEASE BY SECURED PARTY;ASSIGNOR:WELLS FARGO BANK, NATIONAL ASSOCIATION, AS NOTES COLLATERAL AGENT;REEL/FRAME:051398/0827 Effective date: 20191220 |
|
AS | Assignment |
Owner name: SONRAI MEMORY LIMITED, IRELAND Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNORS:MICROCHIP TECHNOLOGY INC.;ATMEL CORPORATION;MICROSEMI CORPORATION;REEL/FRAME:051799/0956 Effective date: 20200203 |
|
AS | Assignment |
Owner name: NERA INNOVATIONS LIMITED, IRELAND Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:SONRAI MEMORY LIMITED;REEL/FRAME:066778/0178 Effective date: 20240305 |