US20070021076A1 - Direct-conversion transmitting circuit and integrated transmitting/receiving circuit - Google Patents
Direct-conversion transmitting circuit and integrated transmitting/receiving circuit Download PDFInfo
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- US20070021076A1 US20070021076A1 US11/529,326 US52932606A US2007021076A1 US 20070021076 A1 US20070021076 A1 US 20070021076A1 US 52932606 A US52932606 A US 52932606A US 2007021076 A1 US2007021076 A1 US 2007021076A1
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/14—Balanced arrangements
- H03D7/1425—Balanced arrangements with transistors
- H03D7/1433—Balanced arrangements with transistors using bipolar transistors
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03C—MODULATION
- H03C3/00—Angle modulation
- H03C3/38—Angle modulation by converting amplitude modulation to angle modulation
- H03C3/40—Angle modulation by converting amplitude modulation to angle modulation using two signal paths the outputs of which have a predetermined phase difference and at least one output being amplitude-modulated
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/14—Balanced arrangements
- H03D7/1425—Balanced arrangements with transistors
- H03D7/145—Balanced arrangements with transistors using a combination of bipolar transistors and field-effect transistors
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/14—Balanced arrangements
- H03D7/1425—Balanced arrangements with transistors
- H03D7/1458—Double balanced arrangements, i.e. where both input signals are differential
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
- H03D2200/0001—Circuit elements of demodulators
- H03D2200/0025—Gain control circuits
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
- H03D2200/0001—Circuit elements of demodulators
- H03D2200/0033—Current mirrors
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
- H03D2200/0041—Functional aspects of demodulators
- H03D2200/0043—Bias and operating point
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- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D2200/00—Indexing scheme relating to details of demodulation or transference of modulation from one carrier to another covered by H03D
- H03D2200/0041—Functional aspects of demodulators
- H03D2200/0047—Offset of DC voltage or frequency
Abstract
A transmitter is provided which includes a transmitting circuit that does not require a high-performance low noise VCO restricting cost reduction thereof and that can reduce the number of parts without requiring an RF filter. A direct conversion that does not require a transmission VCO is applied to the transmitting circuit. In order to achieve noise reduction in a receiving band, low-pass filters are provided at IQ input sections of a modulator that converts IQ signals into RF signals. In comparison with a conventional transmitter using offset PLL, an external VCO required in addition to an RF integrated circuit, a power amplifier, and a front end circuit is reduced. Even in current transistor performance, by using a filter having rapid waveform characteristics such as a SAW more inexpensive than the VCO, or the like, it is possible to provide a GSM/GSM 1800/GSM1900 triple band transmitter.
Description
- This is a continuation application of U.S. application Ser. No. 10/073,029, filed Feb. 12, 2002, the entire disclosure of which is hereby incorporated by reference.
- The present invention relates to mobile communication equipment, and particularly to a direct-conversion transmitting circuit suitable for large scale integration and to an integrated transmitting/receiving circuit using the same.
- According to rapid spread of the mobile communication equipment, requests for miniaturization and lower cost thereof have increased. Because of this, it is expected to apply a voltage control type oscillator (VCO), or an integrated circuit whose filter number is reduced and whose integration is enhanced. What is given as one conventional example of transmitting equipment is “RF Circuits Technique of Dual-Band Transceiver IC for GSM and DCS1800 applications” published in pages 278 to 281 of manuscripts for IEEE 25th European Solid-State Circuits Conference on 1999 by K. Takikawa et al.
- As an important item on a transmitting circuit design, reduction of noise leakage into receiving frequency band has been given.
FIG. 2 shows a relationship between transmission power defined by specifications for European cellular phones (GSM) and noise generated in receiving band. As indicated by an allowedoutput spur level 202 in a GSM receiving band, it is required that noise in receiving aband 204 at slightly 20 MHz distant from an upper limit of atransmission band 203 is suppressed up to −79 dBm/100 kHz (−129 dBm/Hz) or less relative to a maximum output power of 33 dBm in aGSM output signal 201. That is, a difference between a transmission signal and a noise level is required to be −112 dBc or more. If a band pass filter or the like is applied to an output portion of a power amplifier, the above-mentioned specification can be achieved. However, decrease in efficiency thereof is generated due to influence on losses of the filter. Thus, an offset PLL is applied as a constitution using no filter. -
FIG. 18 is a circuit constitution diagram showing a transmitter to which the offset PLL is applied. The transmitter is composed of an IF signal generatingsection 1815 and aPLL section 1814. First, an operation of the IF section will be described here. I and Q signals each having a band of 200 kHz are input. These input signals are mixed in intermediate frequency (IF)local signals mixers local signals VCO 1806 by a 90°phase shifter 1807. By adding outputs of themixers phase comparator 1802 located at a downstream side of the circuit, the IF signal is amplified at anamplifier 1810. After harmonics generated by themixers amplifier 1810 are removed by a low-pass filter 1811, the IF signal is input to thephase comparator 1802 of the aPLL section 1814. - The
PLL section 1814 is characterized by including amixer 1801, and converts a frequency of an output signal of theVCO 1800 operated by an RF frequency, into an IF frequency fIF (270 MHz), by means of amixer 1801, and outputs amounts of error between the IF signal and the output signal of theVCO 1800 by means of thephase comparator 1802. A frequency of the output error signal is lowered up to a baseband signal band that is the same as the I and Q input signals. High frequency noise of the error signal is suppressed by the low-pass filter 1803. A cutoff frequency of a PLL closed loop of a filter, the PLL closed loop which is denoted byreference numeral 1816, is about 1.6 MHz in a signal band of 200 kHz, and a noise of 20 MHz is greatly suppressed. Because of this, the noise generated in band which is a 20 MHz distant from an output signal of the VCO 1800 is greatly suppressed. Therefore, even if an output of theVCO 1800 is directly connected to a power amplifier PA, it is possible to suppress noises of receiving band up to −79 dBm/100 kHz (−129 dBm/Hz) or less without newly connecting a filter to an RF signal. - In a transmitter using the offset PLL, although a
portion 1817 enclosed in solid line shown inFIG. 18 is integrated, theVCO 1800 is an external part because low noise characteristics are required. However, if the offset PLL is used, an external filter for high frequency is not required. Therefore, it is possible to be widely applied as a transmitter with high efficiency. - As described previously, a conventional example applying an offset PLL has been used as a transmitter because requiring no external filter. However, in the transmitter applying the offset PLL, there has been the limit of cost reduction because an external VCO with low noise is required.
- An object of the present invention is to provide a direct-conversion transmitting circuit that does not require a low noise VCO having high performance and restricting the cost reduction in order to achieve reduction of the number of parts thereof, and that does not require an expensive and external high frequency filter such as a surface acoustic wave (SAW) filter or the like.
- An object of the present invention is also to provide a transmitter/receiver using a direct-conversion transmitting circuit.
- In order to solve problems described above, a transmitting circuit according to the present invention has an element of a circuit that uses a direct-conversion requiring no transmission VCO, and that provides a low-pass filter with each of I and Q (hereinafter, referred to as IQ) input portions of a modulator for converting IQ signals to a RF signal in order to achieve noise reduction in receiving band. An integrated transmitting/receiving circuit according to the present invention uses this direct-conversion transmitting circuit in a transmitting circuit section thereof. Concrete descriptions will be made in the following embodiments.
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FIG. 1 is a configurational view showing an embodiment that is a direct-conversion transmitting circuit according to the present invention. -
FIG. 2 is a view showing a relationship between transmission power defined by a GSM specification and noises in receiving band. -
FIG. 3 is a view showing a relationship between transmission power defined by a GSM1800 specification and noises in receiving band. -
FIG. 4 is a view showing noise generation factors of a direct-conversion transmitter. -
FIG. 5 is a trial manufacture circuit diagram used for checking effects of the present invention. -
FIG. 6 is a characteristic-wiring diagram between an input signal level and a noise level showing measurement results of the circuit shown inFIG. 5 . -
FIG. 7 is a characteristic-wiring diagram between a cutoff frequency and a phase precision of a filter showing the measurement result of the circuit shown inFIG. 5 . -
FIG. 8 is a level diagram showing a transmitter of a first embodiment that is the present invention used as an example of a GSM1800. -
FIG. 9A is a view showing amplitude response of a first and second filters. -
FIG. 9B is a view showing group delay response of a first and second filters. -
FIG. 10 is a configurational view of a mixer circuit using a second filter showing a second embodiment of the present invention. -
FIG. 11 is a view showing a third embodiment of the present invention, and is a circuit configurational view provided with a function of correcting a DC offset of a mixer constituting an modulator. -
FIG. 12 is a view showing a fourth embodiment of the present invention, and is a configurational view reducing the use number of AD converters for DC offset correction. -
FIG. 13 is a view showing a fifth embodiment of the present invention, and is a circuit diagram of an important part of a configuration using a second order filter as input of a mixer section which constitutes a modulation section. -
FIG. 14 is a view showing a sixth embodiment of the present invention, and is a circuit diagram of an important part of a configuration using a MOSFET as a second order filter provided for an input of a mixer section which constitutes a modulation section. -
FIG. 15 is a view showing a seventh embodiment of the present invention, and is a circuit diagram of an important part showing a configuration of a transmitting/receiving integrally integrated circuit that corresponds to a GSM, GSM1800, and GSM 1900. -
FIG. 16 is a view showing an eight embodiment of the present invention, and is a configurational view showing the case of having doubled an operating frequency of an oscillator in the circuit ofFIG. 15 . -
FIG. 17 is a view showing a ninth embodiment of the present invention, and is a circuit diagram of an important part showing a configuration of a transmitting/receiving integrally integrated circuit that corresponds to a GSM850, GSM, GSM1800, and GSM1900. -
FIG. 18 is a circuit diagram of an important part showing a configuration of a conventional transmitter which uses an offset PLL circuit. -
FIG. 19 is a level diagram of a transmitter according to a first embodiment of the present invention that uses a GSM as an example. -
FIG. 20 is views showing an input waveform, a two-frequency division waveform and a four-frequency division waveform of two frequency dividers that are used for generating a local signal in the constitution ofFIG. 16 - Preferred embodiments of a direct-conversion transmitting circuit and an integrated transmitting/receiving circuit using the same according to the present invention will be described in detail below, with reference to the accompanying drawings.
- A first embodiment of the present invention will be described with reference to FIGS. 1 to 8 and
FIG. 19 . In this embodiment, similarly to a conventional example, European cellular phone GSM (900 MHz band) and GSM1800 (1800 MHz band) are targeted as applications. Although there has already been given in the description of the conventional technique, a noise level from a transmitter within receiving band that the GSM specification should satisfy has been shown inFIG. 2 .FIG. 3 shows conditions that the GSM1800 should satisfy. A transmission band (TX Band) 303 is within the range of 1710 to 1785 MHz, and a receiving band (RX Band) 304 is within a range of 1805 to 1880 MHz. Therefore, similarly to the case of the GSM, there is an interval of 20 MHz. It is required to suppress a noise of −71 dBm/100 kHz (−121 dBm/Hz) or less in order to satisfy an allowedspur level 302 in receiving band relative to a maximum transmission power of 30 dBm in anoutput signal 301 of the GSM1800. Thus, a level difference between themaximum transmission power 301 of the transmission signal and the noise in receiving band is −101 dBc as shown inFIG. 3 . As compared with the GSM, this specification is reduced by 11 dB. - Here, problems, which arise in a direct-conversion transmitting circuit and should be solved, will be clarified with reference to
FIG. 4 . - A direct-up-conversion transmitter has the same circuit constitution as the conventional IF
section 1815 shown inFIG. 18 , but a modulation method for directly generating a RF modulation signal is applied therein. Relative to the direct conversion, a direct-up-conversion means the case of executing a frequency conversion that straight increases a frequency from a baseband to a transmission frequency band, and a direct-down-conversion means the case of executing a frequency conversion that straight decreases a frequency from a receiving frequency band to a baseband. - In
FIG. 4 , AD converters (ADC) 402 and 403 generating respective IQ signals are also shown in addition to amodulation 106 constituted bymixers rectangle 410 and displays frequency characteristics of a signal/noise level of an output section in theAD converter 402, an output section of theAD converter 402 includes asignal body 411, aturn noise 412, and athermal noise 413 which a circuit thereof has. An output section of theAD converter 403 also has the same characteristics as that of theAD converter 402. - In order to suppress the turn noise, low-
pass filters pass filter 404 is shown in a rectangle denoted byreference number 414, and suppresses a signal equal to or more than a cutoff frequency fCUTOFF of the low-pass filter 404, and includes a signal 415 and anoise 416 which is equal to or less than the cutoff frequency. The low-pass filter 405 also has the same output as the low-pass filter 404. Output signals of the low-pass filters I input 108 and aQ input 109 of the transmitting circuit, respectively. An optimal output signal amplitude of the AD converter is about 2 Vpp in normal differential. On the other hand, an optimal input level of the mixer depends on a circuit constitution and, for example, is 0.8 Vpp, and so is different from the output signal level of the AD converter. Besides, since optimal bias levels thereof are also different from each other,attenuators - Since generating noises, an output of the
attenuator 103 includes asignal 419,noise 420 equal to or less than the cutout frequency of the low-pass filter, and further anoise 421 generated by the attenuator as shown in arectangle 418. Theattenuator 104 also has the same output as theattenuator 103. - Each noise of the
attenuators mixers mixers phase shifter 100 in accordance with an output of anoscillator 105. As typical phase shifters, there are two types of one using a CR filter and one using a frequency divider.. However, in the case of one using the frequency divider, an oscillation frequency of theoscillator 105 is twice as high as the carrier frequency. - An output of the
modulator 106 composed of themixers rectangle 422. This modulation signal includes a modulator output signal 424, and athermal noise 425 caused by the AD converter for a modulator output, and anoise 426 caused by the attenuator for a modulator output. In particular, thenoise 421 of the attenuator extends within a wide band, and the modulated noise also exists as thenoise 426 caused by the attenuator in a wide band. - A modulation signal is further amplified by a
driver amplifier 107, and is output via anoutput terminal 132. This signal includes a wide band noise therein. Thus, in order to reduce a noise within a receiving band which is 20 MHz distant from a transmission band, a high frequency (RF)filter 430 which has a rapid waveform characteristic and which applies a SAW (surface acoustic wave) device, dielectric resonator, or the like is required. - In order to eliminate the high frequency filter, it is required to reduce a wide band noise each generated by the
attenuators FIG. 1 , low-pass filters mixers FIG. 1 , the outputs of theAD converters IQ input terminals FIG. 4 . That is, the output of theAD converter 402 contains asignal body 411, aturn noise 412, and athermal noise 413 that the circuit has, as shown in the rectangular 410 ofFIG. 4 . As indicated in therectangle 414 ofFIG. 4 , signals in theIQ input terminals noise 416 equal to or less than the cutoff frequency fCUTOFF of the external low-pass filters - In the
attenuators FIG. 4 , a wide band noise is generated, and the noise having a frequency equal to or more than a cutoff frequency fCUTOFF2 is damped by a low-pass filter having the cutoff frequency fCUTOFF2 connected immediately before themixers rectangle 118 ofFIG. 1 , and thereby the wide band noise caused by the attenuator becomes anoise 121 equal to or less than the cutoff frequency fCUTOFF2. Thus, an output of themodulator 106 composed of themixers rectangle 123, becomes a signal modulated in both sides by regarding the carrier frequency “fc” as a center, but the noise level becomes small which is a frequency equal to or more than fCUTOUT2 distant from the center carrier frequency “fc” in the modulated RF signal. Therefore, it is possible to reduce the noise level in receiving band without adding, to theRF output 132, anRF filter 430 shown inFIG. 4 . -
FIG. 8 shows an example of a level diagram of a transmitter that satisfies the specification of a GSM1800. The transmitter is composed of themodulator 106, thedriver amplifier 107, apower amplifier 801, and a front end circuit (FEM) 802 constituted by switches or the like. InFIG. 8 ,reference number 804 denotes an input end of the transmitter, andreference number 803 denotes an output end of the transmitter. The respective gains of thedriver amplifier 107,power amplifier 801, andfront end circuit 802, and performance of a noise index NF are estimated from a circuit existing currently so as to have characteristics capable of being realized. In the case where an output signal level of themodulator 106 is −8 dBm, it is understood that the noise level in receiving band at theinput end 804 of the transmitter satisfies the specification of the GSM1800 when being −160 dBm/Hz (−152 dBc) or less. - Specification of the case of the GSM is shown in
FIG. 19 , similarly toFIG. 8 . As the frequency is lower, it is expected that each noise index NF of thedriver circuit 107 and thepower amplifier 801 is reduced. However, since the specification is strict, it is necessary that the noise level in receiving band is made −171 dBm/Hz (−166 dBc) or less when themodulator 106 has an output of −5 dBm. - In order to make performance checks on a direct-up-conversion transmitting circuit adopted in the present embodiment, the circuit shown in
FIG. 5 has been manufactured for trial by a Bi-CMOS process in which a transistor cutoff frequency “fT” is 20 GHz at a rule of 0.35 μm, and evaluation thereof has been executed. The circuit manufactured for trial is a portion of amodulator 510 composed of: a group of IQ signal mixer circuits having themixers buffer amplifiers input terminal 505 of a 90°phase shifter 100. The 90°phase shifter 100, theattenuators pass filters filters resistor 511 and acapacitor 512 as shown in the circuit diagram in a dotted line denoted byreference number 500. Amatching circuit 503 composed of individual parts is provided at an output end of themodulator 510, and is constituted to fetch a transmission signal from anoutput terminal 504 after a 50Ω alignment and a differential-single conversion are carried out. - In order to investigate effects of a low-pass filter in a direct-up-conversion transmitting circuit proposed in the present embodiment, evaluation oriented to a GSM1800 has been carried out about the case of no use of a filter and the case where the filter has cutoff frequencies of 4.9 MHz and 440 kHz.
FIG. 6 shows evaluation results obtained when a local signal (LO) has an input level of 0 dBm, each IQ input signal has a DC level of 1.2 V, and an output has a frequency of 1.75 GHz. An abscissa denotes a voltage level (dBV) of each IQ input signal, and a vertical axis denotes a ratio between a transmission signal and a noise in receiving band. In this figure, a characteristic line indicated by a black filled circle denotes noise characteristics in receiving band, in the case of no use of filter. A characteristic line indicated by a square denotes the case where a filter with a cutoff frequency of 440 kHz is used. A characteristic line indicated by asterisk “*” denotes noise characteristics in receiving band, in the case where a filter with a cutoff frequency of 4.9 MHz is used. The level of −17 dBV is an-allowable maximum input level to satisfy strain specification included in a GSM, a GSM1800 and the like. - When the IQ input level is −17 dBV, an output thereof is −7 dBm. In comparison with respective characteristics under this case, the case of no use of filter has a level of −142 dBc/Hz (−149 dBm/Hz) while the case of use of a filter having a cutoff frequency of 440 kHz has a level of −156 dBc/Hz (−163 dBm/Hz). Therefore, it is understood that the case of use of a filter having a cutoff frequency of 440 kHz is improved by 14 dB. The similar results have been obtained even in the case of GSM transmission frequency band. In this trial testing, although performances relative to the GSM specification are insufficient, it is considered that improvement of device characteristics by use of a SiGe (silicon/germanium) bipolar transistor, and the like, can be achieved, and thereby effects of the present invention are expected.
- Although it is possible to reduce the noise level in receiving band by lowering the cutoff frequency of the filter, degradation of phase precision of a modulation signal is considered due to an affect of frequency characteristics of a group delay.
FIG. 7 shows a relationship between the cutoff frequency of the filter and modulation signal of phase precision. The GSM specification has a phase precision of 5° or less, but if a target value of the phase precision is set at 3° on the basis of a margin thereof, sudden degradation is observed at the cutoff frequency of about 300 kHz or less. Therefore, a phase precision of 3° regarded as the target value cannot be satisfied. Assuming that dispersion of resistance values of the integrated circuit is ±20% and dispersion of capacitance values thereof is ±30%, it is desired that a design value of the cutoff frequency has lower limit of about 500 kHz. - As described above, according to the constitution of the first embodiment that is the present invention in which a low-pass filter is connected immediately before the IQ inputs of each mixer circuit constituting a direct-up-conversion transmitting circuit, it is understood that the spur level in receiving band for an output can be eminently improved.
- A second embodiment of the present invention will be described with reference to
FIG. 9 andFIG. 10 . In an example of the first embodiment described above, there is shown characteristics of the case where the first order filter is connected immediately before the mixer. Although it is desirable that the cutoff frequency is lowered to reduce noise in receiving band, such noise reduction is limited because the phase precision is affected by a group delay deviation which the filter has. Thus, in the present embodiment, an attempt is made to ensure damping quantity in receiving band and suppression of the group delay deviation in signal band by using a high order filter. -
FIG. 9A shows respective amplitude characteristics of a first order low-pass filter 90 a, a secondorder Butterworth filter 90 b, and a thirdorder Butterworth filter 90 c. The characteristics of the first order low-pass filter in the cutoff frequency is 440 kHz, wherein the first order low-pass filter is set under the same conditions as the testing results which is indicated by asterisks “*” shown inFIG. 6 and which is carried out for checks on the first embodiment. The damping quantity at 20 MHz of the first order low-pass filter is about 33 dB. Each damping quantity of the second order and third order Butterworth filters 90 b and 90 c is also set to have a value of 33 dB at 20 MHz. -
FIG. 9B shows group delay characteristics of respective filters. The GSM signal band is about 100 kHz. In the characteristics of thefirst order filter 90 a, a group delay deviation is 20 nsec within a band from 0 KHz to 100 KHz. An error of 1° in a signal of 100 kHz corresponds to 28 nsec, and so, in the case of use of thefirst order filter 90 a, a deviation of about 0.7° occurs in a band of 100 kHz. Since spectrums of the GMSK modulation signal used in the GSM system are not uniform in a band of 100 kHz, this deviation is not equal to an absolute value of phase precision of the signal. However, the case of no filter as indicated by the testing results shown inFIG. 7 is well coincident with degenerative amount of phase errors of the case of using the first order filter having a cutoff frequency of 440 kHz. In the characteristics of the second order and third order Butterworth filters 90 b and 90 c, both group delay deviations thereof are below 0.1 nsec, and so this does not affect the phase deviation thereof. Therefore, in the case of changing filter order from a first order to a second order or more, it is found that eminently improved effects are attained. -
FIG. 10 shows a concrete example of a circuit constitution. Themixers phase shifter 1001 composed of a frequency divider. After the local signal has been amplified at abuffer amplifier 501, two groups ofdifferential pairs 1006 which do switching operation in theGilbert mixer circuit 101 are driven. An I signalinput 108 is damped at theattenuator 103, and an input noise of 20 MHz or more is suppressed at a second order Sallen-Key type active low-pass filter 130 composed of afeedback transistor 1003, aresistor 1004, and acapacitor 1005. - The Sallen-Key filter can be composed of a Butterworth type filter or a Chabyshev type filter by selecting an element value. The filter output drives a mixer
input stage transistor 1002, and is converted into a high frequency signal by means of two groups ofdifferential pairs 1006, and is fetched from respective connection ends betweenload resistances 1007 and thedifferential pairs 1006 of a mixer. Here, although the second order filter is shown, the third order filter can easily be used instead of the second order filter. However, as far as the GSM system is concerned, as is evident fromFIG. 9 , necessary and sufficient characteristics can be obtained by the second order Butterworth type filter. By the present embodiment, the direct-up-conversion transmitter capable of suppressing band free noises and reducing phase errors can be achieved. - A third embodiment of the present invention will be described with reference to
FIG. 11 . The same components asFIG. 1 are denoted by the same reference number asFIG. 1 . The present embodiment relates to reduction of a carrier leak generated by an effect of a DC offset of a mixer circuit input in respective circuit constitutions of the first and second embodiments described above. In a direct-conversion transmitting circuit that is the present invention, since many circuits such as a filter, a attenuator and the like are connected to a mixer input, increase in a DC offset generated at a mixer input terminal is considered. As a countermeasure thereof, a circuit constitution of the present embodiment is proposed. - First, a mixer carrier leak will be described here. This mixer functions as a multiplier. As shown in formula (1), a modulation wave fc(t) is generated by multiplying a baseband input signal f(t) and a local signal cos(2πfc).
fc(t)=f(t)×cos(2πfc) (1) - When a DC offset a is added to the mixer input, as shown in formula (2), a single term of a carrier signal is generated and this causes degradation of modulation precision.
fc(t)=f(t)×cos(2πfc)+αcos(2πfc) (2) - In order to correct the DC offset, in the present embodiment respective channels of the I and Q are provided with a
bias correction circuit 1103 consisting of an AD converter ADC for detecting an offset, a DA converter DAC for generating a correction bias, and a control section CNT which carries out control for minimizing an offset and simultaneously stores correction conditions. Correction is carried out within time from supply of power to beginning of transmission. The control section CNT is composed of a control register and a logic circuit and the like. By the present embodiment, a direct-up-conversion transmitter reducing an effect of a DC offset can be achieved. - A fourth embodiment according to the present invention will be described with reference to
FIG. 12 . The same components asFIG. 11 are denoted by the same reference number asFIG. 1 . In the third embodiment described above the DC offsetcorrection circuit 1103 has been provided separately from the I and Q while in the present invention AD converters ADC whose each circuit scale become large are shared with the I and Q and thereby the entire circuit scales are reduced. - The AD converter ADC is selectively connected to the I and Q signal lines by means of a switch SW. The DA converter DAC is provided exclusively for each of the I and Q signal lines. The control section CNT is also provided exclusively for each of the I and Q signal lines, and thereby each of the DC offsets is independently controlled. Since correction cannot be made for the I and Q simultaneously, correction time of the present embodiment is required about twice further than that of the third embodiment. By the present embodiment, a direct-up-conversion transmitter reducing an effect of the DC offset can be achieved with a small circuit scale.
- A fifth embodiment according to the present invention will be described with reference to
FIG. 13 . The same components asFIG. 10 are denoted by the same reference number asFIG. 10 . In the second embodiment shown inFIG. 10 , the active low-pass filter 130 and themixer 101 have been constituted as independent circuits, respectively. In contrast, in the present embodiment, respective functions of both theinput transistor 1002 of the mixer and the emitterfollower circuit transistor 1003 of an active low-pass filter 130, as shown inFIG. 10 , are integrated with atransistor 1302 shown inFIG. 13 . In this manner, it is possible to achieve a circuit which is not saturated even if a large baseband signal is applied thereto. - The I input signal 108 is converted to current from voltage at a differential input circuit composed of a
PNP type transistor 1303. Thetransistors differential pairs 1006 for mixer, and a mixer output is supplied from a connection end connected to eachload resistor 1007. A low-pass filter 1300 is composed of resistors R1 and R2 connected in series to bases of thetransistors transistor 1302, respectively. In addition, emitters of thetransistors - A sixth embodiment according to the present invention will be described with reference to
FIG. 14 . The same components asFIG. 10 are denoted by the same reference number asFIG. 10 . In the fifth embodiment shown inFIG. 13 , since the resistors R1 and R2 each constituting the filter are connected in series to bases of thetransistors - In contrast, in the present embodiment,
MOSFETs attenuator 103 is changed from thePNP transistor 1303 to aP type MOSFET 1402. This is because input impedances thereof are increased and driving thereof can be achieved by using small amount of power. - By the present embodiment, since a large resistor can be applied to a filter, capacitive value thereof can be reduced. As a result, a low noise direct-conversion transmitter having small element area can be achieved.
- A seventh embodiment according to the present invention will be described with reference to
FIG. 15 . The present embodiment is a transceiver IC adopting direct-conversion and using transmission/reception that is applied to a triple band of GSM/GSM1800/GSM1900. A receiving circuit of thistransceiver IC 150 receives frequency bands from 925 to 960 MHz, 1805 to 1880 MHz, and 1930 to 1990 MHz of the respective GSM/GSM 1800/GSM1900. A large interference wave other than each frequency band is erased by means of anexternal RF filter 1506. Therefore, the signal is amplified bylow noise amplifiers conversion mixers mixers frequency divider 1512. The signal having I and Q components are subjected to processing for eliminating interference waves and for gain regulation in baseband programmable gain variable amplifiers/channel low-pass filter rows (PGA & LPS) 1513 and 1514, and thereafter is output as I and Q signals in a downstream side of the circuit. - A transmitting circuit applies any of the embodiments introduced previously. The IQ transmission signals are adjusted to a desired signal level at the
attenuator 103, and a wide band noise generated by theattenuator 103 is suppressed at the low-pass filter 130. The signals whose noises are suppressed at thefilters 130 are converted into modulation signals having RF frequency, by themodulator 106 composed of a group ofmixers mixers frequency divider 100. - The output signal of the
modulator 106 is amplified by a GSMdriver amplification circuit 1500 or adriver circuit 1501 compatible with the GSM1800 and GSM1900. Aband pass filter 1502 such as a SAW filter having rapid waveform characteristics, or the like is connected to an output of the GSM driver circuit, and thereby residual noise in receiving band which is 20 MHz distant is eliminated. Here, although the SAW filter having rapid waveform characteristics is connected to the GSM output in accordance with the testing results shown in the first embodiment, this filter can replace an inexpensive LC filter according to higher performance of the circuit. The output signal of the filter is amplified by a power amplifier module (PA module) 801. - A
simple LC filter 1503 is connected to an output of thedriver amplification circuit 1501 compatible with the GSM1800 and GSM1900, and the signal thereof is amplified by thepower amplifier module 801 after high harmonics are eliminated. Here, thepower amplifier module 801 packages the GSM modulator and the modulator compatible with the GSM1800 and GSM1900. The amplified signal is transmitted from an antenna via a low-pass filter (LPF) 1504 that eliminates high harmonics generated by an output of the amplifier in thepower amplifier module 801, and via a transmission/reception changeover switch (S/W) 1505. - A voltage control oscillator (RF VCO) 1515 receives and constantly oscillates a feedback loop by means of a synthesizer (RF PLL Synth) 1516, and generates a transmission/reception signal as follows.
- GSM reception: An
oscillator 1515 oscillates within a range from 3700 to 3840 MHz. The output of this oscillator is frequency-divided into two sections by means of afrequency divider 1517, and further is frequency-divided into two sections by thefrequency divider 1512. Thereby, a local signal for a GSM reception, which drives themixers - GSM1800 reception: The
oscillator 1515 oscillates within a range from 3610 to 3760 MHz. The output of this oscillator is directly connected to thefrequency divider 1512 without passing through thefrequency divider 1517, and is frequency-divided into two sections by aswitch 1518. Thereby, the local signal for the GSM1800 reception, which drives themixers - GSM1900 reception: The
oscillator 1515 oscillates within a range from 3860 to 3980 MHz. The output of this oscillator is directly connected to thefrequency divider 1512 without passing through thefrequency divider 1517, and is frequency-divided into two section by theswitch 1518. Thereby, the local signal for the GSM1900 reception, which drives themixers - GSM transmission: The
oscillator 1515 oscillates within a range from 3520 to 3660 MHz. The output of this oscillator is frequency-divided into two sections by means of afrequency divider 1519, and further is frequency-divided by means of thefrequency divider 100. Thereby, a local signal for the GSM transmission, which drives themodulator 106, is obtained. - GSM1800 transmission: The
oscillator 1515 oscillates within a range from 3420 to 3570 MHz. The output of this oscillator is directly connected to thefrequency divider 100 without passing through thefrequency divider 1519, and is frequency-divided into two sections by means of aswitch 1520 without passing through afrequency divider 1519. Thereby, a local signal for the GSM1800 transmission, which drives themodulator 106, is obtained. - GSM1900 transmission: The
oscillator 1515 oscillates within a range from 3700 to 3820 MHz. The output of this oscillator is directly connected to thefrequency divider 100 without passing through thefrequency divider 1519, and is frequency-divided into two sections by theswitch 1520. Thereby, a local signal for the GSM1900 transmission, which drives themodulator 106, is obtained. - In order to make such operations, the
oscillator 1515 operates within a range from 3420 to 3980 MHz. By the present embodiment, a direct-conversion circuit can be achieved for both of transmission and reception by using one voltage control oscillator. - An eighth embodiment according to the present invention will be described with reference to
FIG. 16 andFIG. 220 . InFIG. 16 , the same components asFIG. 15 are denoted by the same reference number asFIG. 15 . In the sixth embodiment described previously, the case of the GSM frequency-divides the output of theoscillator 1515 into four sections, and the cases of the GSM1800 and the GSM1900 each are frequency-divided into four sections and generate the local signal.FIG. 20 shows an input waveform, a two-frequency division waveform, and a four-frequency waveform of a two-frequency divider (i.e., an output waveform of the oscillator 1515). - When the output of the
oscillator 1515 is inputted into the two-frequency divider, two waveforms ofoutputs edges outputs edges 2005 of the input of the frequency divider (that is, the output of the oscillator), and the other is synchronized with a fallingedge 2006 of the input of the frequency divider. - If the output of the
oscillator 1515 has a duty ratio of 50%, a phase difference between these two outputs is 90°. In the case where the duty ratio is shifted from 50%, an error occurs in the phase difference. When the waveform of theoutput 1 is further frequency-divided into two sections, waveforms ofoutputs edges edge 2005 of the oscillator output, and a signal precisely having a phase difference of 90° phase difference can be generated without depending on the duty ratio of the oscillation waveforms. - Therefore, although a signal having a precise phase difference can be generated relative to the GSM in the sixth embodiment described above, there occurs each error depending on the duty ratio of the oscillation waveforms in the GSM1800 and the GSM1900. The
transceiver IC 160 employing a direct-conversion for both of transmission and reception, which is applied to the triple band of the GSM/GSM1800/GSM1900 in the present embodiment shown inFIG. 16 , is constituted in which the oscillation frequency of thevoltage control oscillator 1515 is set within such a range from 6840 to 7960 MHz as to double the sixth embodiment, and afrequency divider 1600 is newly connected to the oscillator output. In this manner, by using eight-frequency division for the GSM and using four-frequency divisions for the GSM1800 and GSM1900, a local signal precisely having a phase difference of 90° can be generated without depending on the duty ratio of the oscillator waveforms. - A ninth embodiment according to the present invention will be described with reference to
FIG. 17 andFIG. 19 . In FIG. 17, the same components asFIG. 15 are denoted by the same reference number asFIG. 15 . The sixth and seventh embodiments described above has been the case of the triple band compatible ICs for the GSM, GSM1800 and GSM1900. In contrast, the present embodiment is the case a 4-band compatible transceiver IC to which a GMS850 is newly added. Alow noise amplifier 1700 for receiving a range from 869 to 894 MHz has been added to a receiving circuit of thistransceiver IC 170. In the transmitting circuit, a GSMexclusive driver circuit 1500 is actuated for signals of both of the GSM and GSM850. External RF filters for the GSM and GSM850 each are composed of an LC filter, and eliminates high harmonics. In order to use the GSM and GSM850 circuits in combination, during the GSM and GSM850 transmissions, the noise level in receiving band for the driver circuit output is required to satisfy a value of −160 dBm/Hz or less at an output of 3 dBm (seeFIG. 19 ). In the case where the noise level does not meet this condition, there may be provided a construction in which thedriver circuits 1500 compatible currently therewith are provided for each of the GSM and GSM850 to increases two systems, and an exclusive SAW filter is used for each of the systems. - By the present embodiment described above, a 4-band compatible transceiver IC can be achieved by using the small number of external elements.
- As described above, several preferred embodiments of the present invention has been described. However, the present invention is not limited to these embodiments, and of course various design modifications thereof can be made without departing from the spirit of the present invention.
- According to the present invention, in comparison with a conventional transmitter applying offset PLL, even if a required external VCO in addition to an RF integrated circuit, a power amplifier and a front end circuit is reduced and current transistor performance is maintained, then a GSM/GSM1800/GSM 1900 triple band transmitter/receiver can be achieved by using one filter having rapid waveform characteristics, such as a SAW or the like more inexpensive than the VCO. Further, by improving transistor characteristics, a triple band or quadrant band transmitter/receiver can be formed without using expensive external parts.
Claims (3)
1. A direct-conversion transmitting circuit comprising:
a modulator to modulate I and Q signals into a transmitting frequency signal, the I and Q signals being inputted from a base band circuit to said modulator;
a first driver amplification circuit coupled to an output node of said modulator to amplify a first transmitting frequency signal being modulated into a first frequency band through said modulator;
a second driver amplification circuit coupled to an output node of said modulator to amplify a second transmitting frequency signal being modulated into a second frequency band through said modulator, the second frequency band being higher than the first frequency band;
first and second low-pass filters being coupled at output nodes thereof to input nodes of said modulator;
first and second gain/bias adjusters being coupled at output nodes thereof to input nodes of said first and second low-pass filters, respectively; and
wherein said modulator comprises first and second mixers, and a first phase shifter,
wherein high frequency output terminals of said first and second mixers are connected to each other,
wherein an output terminal of said first low-pass filter is connected to an input terminal of said first mixer, and an input terminal of said first low-pass filter is connected to an output terminal of said first gain/bias adjuster to suppress a noise generated by said first gain/bias adjuster,
wherein an output terminal of said second low-pass filter is connected to an input terminal of said second mixer, and an input terminal of said second low-pass filter is connected to an output terminal of said second gain/bias adjuster to suppress a noise generated by said second gain/bias adjuster,
wherein a first output terminal of said first phase shifter is connected to a local signal input terminal of said first mixer, and a second output terminal of said first phase shifter is connected to a local signal input terminal of said second mixer,
wherein an input signal generated from an output signal of a first AD converter is applied to an input terminal of said first gain/bias adjuster to reduce difference in gain and bias levels between an input signal of said first mixer and an output signal of said first AD converter, and
wherein an input signal generated from an output signal of a second AD converter is applied to an input terminal of said second gain/bias adjuster to reduce difference in gain and bias levels between an input signal of said second mixer and an output signal of said second AD converter.
2. The direct-conversion transmitting circuit according to claim 1 , wherein said first phase shifter is comprised of a frequency divider circuit.
3. The direct-conversion transmitting circuit according to claim 1 , wherein each circuit of said first and second low-pass filters is comprised of a filter whose order is at least a second order.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US11/529,326 US20070021076A1 (en) | 2001-11-15 | 2006-09-29 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
Applications Claiming Priority (4)
Application Number | Priority Date | Filing Date | Title |
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GB0127422.4 | 2001-11-15 | ||
GB0127422A GB2382242B (en) | 2001-11-15 | 2001-11-15 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
US10/073,029 US7116950B2 (en) | 2001-11-15 | 2002-02-12 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
US11/529,326 US20070021076A1 (en) | 2001-11-15 | 2006-09-29 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
Related Parent Applications (1)
Application Number | Title | Priority Date | Filing Date |
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US10/073,029 Continuation US7116950B2 (en) | 2001-11-15 | 2002-02-12 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
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Publication Number | Publication Date |
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US20070021076A1 true US20070021076A1 (en) | 2007-01-25 |
Family
ID=9925825
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US10/073,029 Expired - Fee Related US7116950B2 (en) | 2001-11-15 | 2002-02-12 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
US11/529,326 Abandoned US20070021076A1 (en) | 2001-11-15 | 2006-09-29 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
Family Applications Before (1)
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US10/073,029 Expired - Fee Related US7116950B2 (en) | 2001-11-15 | 2002-02-12 | Direct-conversion transmitting circuit and integrated transmitting/receiving circuit |
Country Status (5)
Country | Link |
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US (2) | US7116950B2 (en) |
EP (1) | EP1315284A1 (en) |
JP (1) | JP4005391B2 (en) |
GB (1) | GB2382242B (en) |
TW (1) | TW531981B (en) |
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Also Published As
Publication number | Publication date |
---|---|
JP4005391B2 (en) | 2007-11-07 |
US7116950B2 (en) | 2006-10-03 |
JP2003152563A (en) | 2003-05-23 |
GB0127422D0 (en) | 2002-01-09 |
GB2382242A (en) | 2003-05-21 |
TW531981B (en) | 2003-05-11 |
GB2382242B (en) | 2005-08-03 |
US20030092416A1 (en) | 2003-05-15 |
EP1315284A1 (en) | 2003-05-28 |
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