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Numéro de publicationUS2341067 A
Type de publicationOctroi
Date de publication8 févr. 1944
Date de dépôt1 mai 1942
Date de priorité14 juin 1941
Numéro de publicationUS 2341067 A, US 2341067A, US-A-2341067, US2341067 A, US2341067A
InventeursWise Raymond O
Cessionnaire d'origineBell Telephone Labor Inc
Exporter la citationBiBTeX, EndNote, RefMan
Liens externes: USPTO, Cession USPTO, Espacenet
Variable frequency bridge stabilized oscillator
US 2341067 A
Résumé  disponible en
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Description  (Le texte OCR peut contenir des erreurs.)

Feb. 8, 1944. R. o. wlsE 2,341,067

VARIABLE FREQUENCY BRIDGE STABILIZED OSC-ILLATOR Original Filed June 14. 1941 2 Sheets-Sheet 1 ATUPNE Feb. 8, 1.944.

2 Sheets-Sheet 2 Ffa. 5



Patente-d Feb. 8, 1944 cette? UNITED sfrmi=z-sy etant ortica VARIABLE FREQUENGY I- fci STABILIZED osCILLAToa aas/'mana' o; Wise, simi ning; N; reserveer ai Bell Telephone N ew'York', N. Y., a corporationiof' New Yorl Original application June 1,4,",1941 SeriaLfNol. 398,042,Vncw Patent No. 2,319,965,dated May.k 2,5, 194:3. Divided and this "applicationwil/[ay'1,'r

v 1942,' Serial No. 441318 7 claims; (ci. estL-3er This invention relates tov oscillation generating circuits, thatA is, oscillators, and more particu'- larly'to oscillators of the bridge stabilized type. This application is a division of application Serial No. 398,042, led June 14, 1941, Patent No. 2,319,965 granted May 25, 1 943.V

It is the object, generally, .of the invention to improve the frequency stability of vacuum tube oscillators, especially,` as resulting fromv changes in the various supply voltages Aand* non-linearity in stability-signicance circuit eleinents, eithery comprised in the circuits immediately associated with the vacuum tube 4or elsewhere. Asubsidiary object is, analogously yto the above, to improve,the` stability of such-oscillators with respect to the amplitude of the generated oscillations. Another object is the improvementof the wavenigrm oi the oscillations and therefore the suppression of harmonics of the fundamentalV ,frequency,

1n recent years it has been reccgmzed thatv'tis largely the harmonics producedby 4the, above non-linearity Whichatfect the irequency stability This affect occurs Ysince Vanyclianges in the-sup.-

ply voltages, which because ofthe!nonlinearity,` will tend to produce relative changes in the am:

plitudes of the harmonics, -will cause achange in the reactance 'at the fundamental frequency., as Weil and'hence a change in that'freq'uency, l Various methods of minimizing 4these affects4 have been proposed `butwnone have been `completely :"vf

satisfactory. For instance, Llewellyn hasshown in hisfpaper Constant frequency oscillatorsi in the Proceedings yofthe Institute 0f RadiowEngineers for December, 1931, that nby including ap'- propriate impedances in the grid andplate ,conf nectionsof the tube,l the frequency ofv'oscillations may bernade independent oi changes inthe grid and plate resistances of the tube. This method,

however,` is not adapted to ,theelimination' oiinstability resulting from harmonic reaction. An

approach was made by Arguirnbau his paper An ,oscillatorl Yllaa/ing a linear' operating :characteristic, Proceedings of the Institute of Radio Engineers for January, 1933, by suggesting thel operationV of, the 'oscillator vacuum `tube'as a linear ampliiier,` as by 'rectifying a lportionof the output to furnish agrid bias to the yacuulntube yand' thereforeY achieving an', automatic yolume control which maintainsl an' equality -betWeenthe gain 'of the amplilierfand' th loss of the assowith ai`1.t'ornaticcont,1 t s of ,the Institute coniernA lated hat the /acu'iirrivtube'jsliuldinou aci. il? any ther cpaaiy than, that@ Ilolifier. Applicant"Vbelievesfgtljlatg-the Meacham plirlffiplaslbestce ,able @fr beineuilizetl ,io` et: terliniasimumlfrequenr andampiaude .stability as effected by the. rinus`- aboyeconsiderat ons.

tion hereinafter to de# Applicants ihren scared; ,contemplate@napraten @free above Y- principle y alternative ftclthat specifically disclosed by'Meeehaia. #specific abject of4 the invention, therefor'egis to improve the operation ci, bridge stabiliZed`A cscillators with respect to, ireo'juency andarnplitude stability, having regard'cto the 1.5, "various above yconsiderations'especially the eect irl) of harrn'onicA reaction.

Since applicants mention hastev do mo'recparti'cularly with comparatively 4lovvf frequencies' and Wit-ha ineansA for readily varyingitbe vfirzfequenc'y,v

anV object subsidiaryto the above' object is to adaiitthev bridge stabilized loscillatorspci the prior art, having particularly, inrnind the Meacham circuits,` to relatively low frequency; an'd yariable frequency operation, Another subsidiary object is to; achieve the rstementionel Qbjectsby bridge netwbr'ks 'and' theili .immediately ,asseiaisdieircuits Wliifzh @emprise triples'. 'of impedance means Y which are themselves inherently stable in"-func tion,` and which; rnroreor/"er,l so cooperate `jin the 'achievement of the desired resultA scas to necesb destinataire@ rlgrleers 'fQr may be outside of the network although still in l the feedback path and, specifically, in shunt to the network as a whole. This means may be a thermal device, that is a thermistor. The ampli-v tude control employed permits the analysis of the circuits by linear circuit methods.k .A like parallel-T network has been described by Augustadt in his United States Patent 2,106,785, February 1, 1938, for use as a iilter and by Scott in his United States Patent 2,173,427, September 19, 1939, and in his paper beginning in page 226 of the Proceedings of the Institute of Radio Engineers for February, 1938, for use as a selective analyzer and as an element of an oscillator circuit. In the network as used in the present instance, the elements are so altered as to make any feedback path additional to said network un' necessary and, further. as above indicated a current responsive amplitude control feature is provided. This control adjusts the feedback to the proper value so that the tube functions as a linear amplifier. this therefore resulting in better frequency stability.

In the detailed description of the invention following the brief description of the invention identifying the figures of the drawings which exemplify the invention. the general operating conditions for the oscillator will be treated. A theory for the frequency stability for conditions of large amplifier gain and under actual operating conditions will be developed. A comparison of the theoretical results with those achieved with experimental oscillators will be made and a description will be given of an oscillator found practicable in use which was adapted and modified from the experimental model by use of which the above results were secured.

In the accompanying drawings, Fig. 1 is a schematic showing of an oscillator embodying my invention, the form of the showing being adapted less for a teaching of the details of a complete,

Ipractical, embodiment of the circuit, than to provide a simple means for teaching the basic principles of the invention and to make possible the simple analytical treatment which follows;

Figs. 2 and 3 illustrate graphically the performance characteristics of the oscillator schematically shown in Fig. 1; and

Figs. 4 and 5 illustrate two alternative forms of the oscillator in practicable embodiments.

In Fig. 1 which epitomizes a complete oscillator circuit, the round trip energy fiow path which principally characterizes any self-contained oscillator circuit comprises the feedback and frequency determining network l and the amplifier 2. The network I is identical in continuity with that disclosed in the above-mentioned United States patent to Augustadt 2,106,785, February 1,1938, although for practical convenience and to take account of the somewhat diiferent function and emphasis, it is given a somewhat different analytical treatment herein. It is a parallel-T network comprising two individual T networks one comprising series resistances R and the shunt capacitance C1 and the other comprising series capacitances C and shunt resistance R1, which shunt resistance is current responsive so as to enable it to function as an amplitude control resistance as will be hereinafter explained. The two networks are connected as shown in common to the input circuit of the amplifier represented by equivalent resistance R2 and likewise in common to the output circuit of the amplifier represented by the equivalent resistance Rp. The first identified T network is a filter of the low-pass type which transmits direct currents and low frequency alternating currents with a relatively small loss and attenuates high frequency currents. The last-mentioned T net-r work is a high-pass iilter which therefore provides a' high attenuation to low frequency currents. The two,l in combination, constitute a band elimination iilter which may be proportioned to substantially suppress the transmission of alternating currents in any selected frequency range and to eiiect the complete suppression of a selected 'frequency in that range, as pointed out specifically in the above Augustadt patent. l

The following analysis will treat these two T networkspas a whole, that is, as constituting parts of a single network I. Fig. 1 shows, with identications, the four significant current iiow paths,

certain of which will be made use of specically.

in the analysis.' See the arrows indicating currents i1, i2, i3 and i4. It is evident that currents i1 and i4 are respectively the current entering and leaving the network and that currents iz and is" arev conceived as being'coniined to meshes of the network, each mesh being made up of a part ofV cach of the component T networks. Specifically, the current paths may be traced as follows: cur.- rent i1 follows the path from the resistance Rp through the left-hand resistance R, the capacitance C1, back to Rp; current i2 follows a path through left-hand resistance R, capacitance C1, lamp resistance R1, left-hand capacitance C back to resistance R; currentz'x follows the path from right-hand resistance R through right-hand capacitance C, the lamp resistance R1, and capacitance C1 back to resistance R; andcurrent i4 follows a path from right-hand resistance R. through, the feedback path including input resistance R2, and through capacitance C1 back to said right-hand resistance R.

In the above description of the Fig. lcircuit, the

resistance and capacitance elements perhaps could better be denominated as resistors and capacitors. The selection of the literal labels to identify the elements has purposely been made such that they may be used also to represent the impedance values of the elements as far as may be. For example, the impedances (resistances) of the resistors R have because of the resultant simplification both as to structure and as to analysis.

It is useful to consider the conditions which values R. I'he imped-y ances, that is the reactances, of the capacitors Cv lead'. tofa null: vfor "transmission throughlthe network. This is for the reason thatwith'a Treasonable amplifier gain, :the conditions for generation of oscillations 'will beapproximately satisfied only for such null'condition, for it is only a frequency 'near the null that 'the phase 'and arnplitude requirements Afor oscillation may 'be realized. This `will be more :clearly evident from the discussion of phase and amplitude characteristics of the network givenbelow. An analysis on this basis may provide approximate criteria for the electrical dimensionsiof ,the .network for a desired operating condition. .fi-lso, the expressions which result are useful `in :understanding the operation 4of the circuit. v

The conditions for such null transmission through the network of Fig. 1,'that is for the relation vis 'found to be The plus or minus signs indicate vthat 'the same This equation may be easily proved by Vconventional analysis but itis occasion for proof, ,with the considerable detail involved, is not 'presented by thisfspecification. 'A like statement maybe made for any other item of analysis to be foundvin this specication where Y a complete prooi is lacking.

By combining the Equations the values of reactances X formulae, the conditions for the circuit constants is 4LRC RC1 f Since an exact null is not possible, if oscillations are to occur., that is, since no oscillations could 1 after expressing and X1 bythe usual the null in terms of occur if the effective 'bridge constituted bythe parallel-T network were exactly balanced, vlet Equation 3 be changed to allow some degree of scope in variation from' the conditions taught'by said Equation 3 so as to read RC1 n It will be shown below that avalue of K less than 1 must be used. Also fox-'the null condition,- the input impedance of 'the network, as expressed in 'v complex notation is,

l -l- 2C Figs. 2 and 3 represent respectively the attenuation (insertion loss) and phase characteristic `(insertion phase shift) 'of the" parallel-T network.

The "significance ofthe ordinates is obvious. The

abscissae in both .instances are plotted iin terrns of frequency'andfspecica'lly inV terms of l.the ratio believed that `suiicient fit) V..=1/1t"m=.twee`n ag'iven frequency fandithe :frequencyffor .the .null :condition 'fn as `expressed iin Equation s2. Y" The various .curves 1in Fig. 2 corre- .spon'd tod'different -selections of 'the 'constant K rasfexpressed :in Equation .3a. Although the value of K may be obtained'by correspondinglyvarying thel electrical dimensions of any of the several component elements, ias is apparent rfrom Equation 'Sagitwis l'here assumed 'that its variation is achievedby variationsinthe value of the shunt 'resistance R1. This is partly because, as will be vapparent', flater, casit is apparent from Equation A2, Vchanges in other elements would, .undesirably, also change the frequency and because element r R1 is the element consc1ously included in the network for this purpose, as the amplitude control resistor. It is intended to and does automatically change the balance of the bridge constituted by the parallel-T network to accord with the desired vconditions 'of-operation. However, the curves of Fig.l2, andfth'e same is 'true of Fig. 3, 'equally well represent the situation if the factor K were otherwise changed without attendant change of frequency. Of course the change of frequency represented by theabscssae contemplates a change in the frequency of waves incident on a given network and -not a change in the frequency dimensions of the network itself as expressed in Equation 2 since at this point it is the network per se which is 'being subjected'to analysis and the conditions for itsvuse in a self-oscillatory circuit have not vbeen explored. YIt is noted from Fig. 2 that 'the 'significant frequency is that where it equals the null frequency, that where As indicated by Fig. 3 this is vthe frequency at which there is a phaseshift of 180 degrees.

After the introduction to Figs. 2 and 3 provided by the above, it should be noted that for a critical value of the .shunt resistance, the network offers'anrinnite'los's at the frequency for which the lphase shift is 189 degrees. For all values of u vthis resistance less than the critical value that 1s for. a value of K less than 1, the loss at this frequency point,that is vfor a phase shift of 180 degrees, is finite and its value depends'upon how :nearly'the critical condition is approached. It is Vevident from Fig. 3 that, at the critical (significant) frequency, the phase shift passes Ythrough 180 degrees for a 'shunt resistance either equal to Aor less than the critical value but that for larger resistances, the phase shift passes r through 'zero instead. This is the reason for the v statement made above that a value of K which is less -than 1 must be used.

In order to make the operation of the circuit automatic, the shunt resistance arm, that is the element R1, must include an element who-se resistance -increases with vtemperature where this element is positioned vas in Fig. 1. The cold resistance must be sufficiently lowfso that the network loss is le-ss thanthe amplifier gain at the lfledeg-ree phase point, to respond to the basic condition that there must be sufficient amplification to make up the loss in' the remaining part of the round trip path so that the oscillations `may be capable of being perpetuated. |The amplifier phase shift is also assumed to be nearly 180 degrees. vFrom Fig. 2, it may be noted that the network 'loss around the critical frequency point increases very greatly with increasing shunt resistance. This means that the automatic control function 'of v'the resistance R1 is most effective 'at 'this point and 'therefore most effective at a -that the value of R1 is if the frequency is varied by changing the fre Iput amplitude and frequency near balance of the network. Thus, when the .circuit is completed as'by energizing the amplifier circuit, oscillations will tend to build up, simultaneously increasing the shunt resistance by the heat produced in R1 until the network loss is just equal to the amplifier gain.

One of the most desirable characteristics of the circuit, as will be apparent from Figs. 2 and 3, is that because of the very critical conditions as to loss and phase shift around the critical frequency point, all harmonics are fed back degeneratively and therefore in such a manner as to reduce the net value over that present with no degenerative feedback. Since the generated harmonics are at a very low level as a result of this degeneration together with the use of the very sensitive amplitude responsive control means, there results a very nearly sinusoidal output.

From Equation 2 it may be seen that the frequency of oscillation may be varied by changing R, C1, or C. The amplitude of oscillation will be determined by the value of R1 required to make the loss through the bridge (network) just equalto the gain of the amplifier. should be constant over the frequency range. The value R1 therefore depends on the circuit constants as given by Equation 3a. Thus if the frequency is varied by changing R, C1 or C individually Ri the frequency since these quantities occur in Equation 3a. This is true both because the value Ri to produce the given loss in the vnetwork would have to be changed and because the input impedance of the network as given by Equation 4 will change with resulting variations in the amplifier gain which must be balanced by the network loss. However, if the frequency is varied by varyingthe capacitances C1 and C with a fixed ratio between them and with a constant value of R, the control resistances R1 will'be independent of frequency. This is true with respect to both of the abovev reasons vfor deducng a function of frequency quency significant element individually, as appears from Equations 3 and 4. It may also be demonstrated that thev attenuation and phase characteristics of the circuit depend only on the ratio of these capacitances. As a result the outstability of the oscillator will be independent ofthe frequency provided that the amplifier characteristics are independent of the frequency, For some applications it may not be advantageous by capacitance variation. In such a case the frequency may best be varied by a change of resistance R, but in order to do this most eifectively, a wide range control resistance R1 is required and one which is very sensitive to current changes. Such resistances will be discussed in a later section. f y

An analysis, not here described, has been made which is valid for the limiting condition of very large amplifier gain. In vthe analysis it wasassumed that the oscillator amplifier is a constant current generator and vall parasitic capacitances are neglected. While the analysis covered both the asymptotic and extended theories, the expressions found from the extended analysis are cumbersome although indicating the limiting correctness of the asymptotic expressions. These asymptotic expressions will be given here without derivation and deductions will be made from them to illustrate their application for design This gain would necessarily be a function of tov vary frequency voscillationif R is fixed and I.-legrees in either direction and by (have any vphase shift whatever.

which if differentiated With respect to a variation of supply voltage gives dE fo-T R92 d In Equations A5 and 6 fois the frequency at the null f is the operating departure from fo H is the amplifier phase shift g is the transoonductance of the amplifier E is any supply voltage.

It is to be understood that an asymptotic analysis is intended to show the conditions that are approached as the amplifier gain is allowed to increase without limit.v Equation 5 expresses the stability of the oscillator and says that the stability increases as R and g are increased and decreases as the phase shift of the amplifier, as might be caused by plate supply,choke coils and the like, departs from degrees'.

Equation 6 indicates that thev frequency stability should be independent of the frequency of should be infinite if the departure of the phase shift `of the amplifier from 180 degrees is zero. This independence of frequency stability and frequency on the assumption that R is fixed also assumes, in the formulation of the Equation 5 wherein the quantity R is used'in place of a more complex quantity, that the frequency is changed in the manner above described, namely, by varying capacitances C1 and C while maintaining a constant ratio between It. also indicates that the greater the transconductance of the tube, the more stable the oscillator should be with respect to frequency. It should be noted that the phase shift of a vacuum tube-per se, is necessarily 180 degrees but that the phase shift of an amplifier, which includes a tube as an element, may easily differ from 180 special design external to the tube per se could In deference to convention in the specification hereafter and in the claims, the term zero phase shift or the like will be taken to means a departure from this 180-degree phase shift that is inherent in the tube.

The extended analysis, asl distinguished from the asymptotic analysis here given, bears out the above statements. Also curves plotted from the equations resulting from the said extended analysis show that, as above, the optimum phase shift approaches zero (havingl in mind the definition of zero phase shift a little above) as the transconductance becomes very large and the frequency stability tends to become independent of phase shift. Although it has been stated above and as appears from Equation 6, the frequency of its circuits stability should be infinite if the phase shift of the perimental oscillatorl was built according to the schematic` circuit of. Fig;. l. This circuit will be explained in. detail later.. The control element was composed of twelve .Western Electric C-2 switchboard lamps in. series, since these lamps were. found. to have the best characteristics of any available control lamps of this type. The frequency wasv changed. byv changing the three capacitances simultaneously with` the resistance R fixed.. Thisgvariation` resulted quite closely in the maintenance of a constant ratio between C and C1. as was. desired according. tothe above analysis'.

The` amplitude. of the oscillations was not absolutely constant. with the variation of frequency from. 50 to 20,000 cycles although it. varied only about; 2 decibels. This Variation Was probably due to phase.` shift. inA the amplifier or perhapsy to some. progressive unbalancejnl the tuning capacitances. The' frequency stability was found to become more nearly independent of frequency as the frequency increased. Over a large partv of this range, the. stability was: as low` as about six parts in a million.v The` frequency stability was with ref erence` to. changes in; plate voltage supply. The slight deviation..frompredicted performance may'be. ascribed. to phaseishift. in the amplifier at low frequency dueto thexplatesupply choke coil, cathode. biased network andthe like.

Similar. experiments wereI performed to determine the effect.oftransconductance of the amplifier tube on the; frequency stabilityy as effected by changesin plate supply.. The. transconductance was. varied. by use. of apotentiometer grid leak. Consistentlyswith.theory, the frequency stabilityA increased with the. transconductance and infact became. alinear function. thereof atlarge values of transconductance.

A: similar; en'rperim'ental.v test' was v made to demonstratethe frequency stability, as effected by the; plate supply. voltage as before, as a function of the amplierphaseishift. The amplifier phase shift was. assumed to be zero in the normal operating; state'and phase shift was then produced by shunting..the inputvofthe.'network withV reactance.. As wasito' bev expected andconsistently with the.. above. analysis the stability was-found to be. a maximum; at. very nearly zero phase shift an'd,. specifically., andy likewise consistently with the. theory,.whenx therewas aslightlagging phase.v shift. At thefrequency ofA oscillation, a capacitive reactance: appears; between; the. input terminals of,.as `indicated by Equation Ll. Therefore, .a shuntin'giinductance at this point would tend to.increasetha amplifier. gain andal capacitanceswouldtendl to reduce, the.v gain. In any event', it would beA dicult. to. causev a 1 phase. shift` without an rattendant change in gain. Howeven. the-testwas informative. andtheuse of a shuntcapacitanceat that point to produce the optimum. phase shift wasfoundto offer aA convenient and simple'method'of obtainingv high frequency stability i evenwith' a' comparatively low gain amplifier. Such a-shunt capacitancewouldi` tend to' be mostusefulffor a--xedfrequencyoscillator since.. it Would-benecessary to `vary Ithe capacitance inversely with frequency'for a-variable. oscillator. However, for al variable frequency-- oscillator., the desinable` characteristics' ofv an improving frequency stability in the frequency.. increases. may:

oe-obtained by1.shunting.;the 'network with a ca-VA pacitance which willv produce-1 the optimuml condition at the maximum frequency. y

Fig, 4. illustrates..` a-`4 physical;y embodiment of Ian osoiuatorottheinventcmasconstructed-toleran eral laboratory purposes and as was used for the above-described tests. The elem-ents of the oscillator which are duplicates of the oscillator illustrated schematically in Fig. l have like identifications. Since the feedback network is illustrated in almost complete detail in Fig. 1,y the principal additional showing by this Fig. 4 is with.

reference to the amplifier element. The three capacitances in the network should be equal and, as shown, are caused to vary together simultaneously to change the frequency. This satisfies the. condition earlier stated that the ratio of capacitances. C and C1 shouldA be kept constant when the frequency is changed by corresponding change in said.- capacitances that the control resistanceRi is independent of frequency. Since said resistance Ri is independent of frequency, it may be adjusted to remain at the steepest point of its current-resistance characteristic, this resulting in the most sensitive amplitude control. In the oscillator as used thisl resistance Ri. comprises severalWestern Electric C-2 lamps in series. These lamps have tungsten filaments. and a positive temperature coefcient of resistance.. Such. resistance constitutes a. thermistor since. the effect is thaty of a .change in resistance as. effected by. a change infthermal condition.v which is inturn induced by the. change of the current. traversing.. it. The oscillator' is.`

adapted to be varied infrequency from about 40 cycles to 50..-kilocycles.. The` theoretical characteristics ofsuch an oscillator havebeendndicated. in: the earlier Ypart of' the specificationy and the operational characteristics of ann oscillatorl corre.-

spondingl strictly tof-this .particularl design have been indicatedl mere recently. The. harmonic contentofl the generated"wavev wasfound tobe principallyf the secondwharmonic and-Was about decibels below the fundamental over the entire i0 frequency range sothat the purity of the output wave was mostJ satisfactory. This result islargely attributable, as previously mentioned, to the linear operation of the amplifier and thefeedback properties of the network.

The' amplifier' element of the above" oscillator is'electively a single'tube amplifier, that' is,y the' amplification occurs' in a' single stage' rather than` in" a` plurality of` stages. For practical reasons', having todo with practical convenience' and' the' obtaining 'of'suiiicient' power', two tubes 3` and' 4 are used, these tubes being in effectively parallel' relationship.' The variousresistances" in4 circuit' with the' electrodes of the respective tubes have the functions usually' attributable' to resistances in amplifier tube circuits' generally, as well as insuring a proper balance and matching between' the two tubes here adapted to function in par allel. The p'oints 5' and 5 may, for practical purposes', be treated as the inputf terminals of' the network andthe output' terminals of the' amplier sincev ground point 5 is effectivelyat cathode potential. The'output terminals of the network, and. correspondingly the input. terminals of the ampli-fier, .are at points 5 and Ga. The tubes are given the requisite degree of negative feedback tu accord with the above-indicated design conditions of jthe tubes by networks T and' 8' inl the respective cathode leads'. The l'o'adcircuitv 9 is connected. betweenY point 6 of the network and ground 5 and. therefore effectively across the.

input circuiti of theV network.` Thevarious-impedance elements. illustrated, exclusive. ofthe resistancesrin the. tube electrode leads-v which have 75. beers-:separatem.mentionedf aboveA are weventional in character and function and do not merit specific description.

- The 'use of the tungsten lament lamps for the amplitude control resistance of Fig. 4 places some restrictions upon the oscillator circuit which lead to the use, in some instances, of the alternative circuit arrangement of Fig. 5 presently to be described and claimed specifically herein. The best lamps commercially obtainable require about 2 milliamperes of current flowing through them in order to operate on a steep portion of their resistance-current characteristic. That statement is with reference particularly to the Western Electric C-2 switchboard lamps described as having been used in the circuit of Fig. 4, which type of control resistance means was found to be most elective in practice.

For a permissible plate swing of the oscillator, this limits both the shunt and series resistance arms of the network if the capacitances of the network are to be equal. An inspection of the Equation 5 shows that this tends to restrict the stability of the oscillator, which physically means that an upper limit is imposed by the possible amplifier gain. If the shunt and series capacif tances arenot made equal, more amplifier gain may be attained but this requires more capacitance to cover a given frequency range, if frequency changes are produced by simultaneous variation of the capacitances.

A solution to the problem of obtaining a highly sensitive control is by fixing R1 at the largest value possible for the rangel desired and connecting a resistance having a very large negative temperature coefficient between the grid and the plate of the amplifier as a separate feedback path. This principle will be found to be utilized in the circuit of Fig. 5 to be described later. These negativeV resistance coefficient thermistors are obtainable in a wide range of impedance and current ratings and have a very high rate of change of resistance with current. Their simplicity and compactness, also, makes them highly useful foi circuit applications. The types here had in mind are described in a paper by G. L. Pearson in the Physical Review, volume 57, June 1, 1941, beginning in page 1065. Examples of these thermistors are what are sometimes known as bead thermistors. Frequently, the"bead comprises silver sulphide. As used in the present considered circuit as above described the thermistor would have a very high negative temperature coefficient and would be of the high speed'type;` It would incidentally tend\to have a high resistance. In View of the Pearson paper, it is not believed to be necessary to treat the specific characteristics of such therinistors in this specification.

An accurate calculation of the required thermistor' resistance is difficult, However, an approm'mation for design purposes may be calculated by assuming that the impedance looking back into the network from the grid is the value corresponding to a null. Then, as a, particular example, given an amplifier gain of decibels, a network loss of 38 decibels, an input impedance at the null of 10,000 (1-7`\/2) and a grid leak resistance' of 100,000 ohms, if oscillations are to begin the thermistor impedance must be vgreater than 4.53 megohms. If the network loss were 36 decibels, the required initial thermistor resistancewould drop to 1.32 megohms. As the xed network loss is made to approach the gain of the amplifier in order to realize the maximum frequency stability, the thermistor resistance at the operating point must increase very rapidly. 75.

This tends to Vresult in poorer amplitude stability for it will be apparent vthat a small .per-'f turbation in amplifier gain caused by the 'voltage variation or the like requires a. large change of the thermistor resistance.. A compromise between amplitude and .frequency stabilitymust therefore be made. In the oscillators described, a K of about .85 was used, resulting in an operating thermistor resistance of a few hundred A good amplitude controlis aided by using the constants of the circuit and' thousand ohms.

of the thermistor to obtain operation in the region where a large change in resistance is veifected by a small change in voltage. This may be further improved by the additionofa xed resistance. If a suiciently high resistancethermistor is not obtainable, or is otherwise unsuitable, some of the required loss through the thermistor bridge may be taken by a linear resistance attenuator following the thermistor.

rihe use of these negative temperature coeicient thermistors makespossible an alternative method of control for this oscillator. An.

example is illustrated by Fig, 5 in the circuit of which the frequency may be changed by varying, the series resistance arms simultaneously. As

range. For practical purposes, this system of frequency variation may be superposed on the system applied in the circuit of yFigni wherein the three capacitances are similarly'ganged to vary the frequency. Practically it is.of convenience to vary the frequency over a 10 to 1 range with the variable resistances and to vary the range by variation of the capacitances together in decade steps. In this way, in an experimental oscillator, a range of from 12 to 50,000 cycles wasY readily obtained in four steps. 'I'his amplitude control made it possible to reduce further the harmonic content of the oscillator circuit to. a maximum value of 60 decibels below the fundamental at the high frequency end of each range and about decibels at the low frequency end. These values could be further reduced if needed by choosing a thermistor to control at a lower amplitude.

above, the only significant difference over theA circuit of Fig. 4 relates to the network and thermistor as above. Accordingly, no labelling isapplied to the amplifier element. Since the shunt resistance is no longer Van amplitude;A control means, but only a variable resistance, itis labeled R similarly asA the series resistances y,with which it is ganged. The amplitude control resistance R1 is shown, Vconsistently with the above description, connected in circuit between the gridv and plate of the parallel .connected tubes. K -j Other modications of the invention than above described, will occur to a person skilled in the art, kand all such are considered to fall within the spirit and scope of the invention', yas defined in the appended claims. '1

What is claimed is: y

1. An oscillator comprising-an amplifier', input and output circuits therefor,

The incorporated feedback ampli-v er, however, had a harmonic content 60 decibels a network connected between said circuits, said network comprising a pair of symmetrical T-networks connected in separate paths between said circuits, one of said T-networks consisting of two series resistances and a shunt capacitance, and the other of said T-networks consisting of two series capacitances and a shunt resistance, said capacitances and resistances being proportioned to provide maximum attenuation and phase reversal in the network at substantially the preassigned desired irequency, a current-responsive thermistor having a negative temperature coeiiicient of resistance connected across the series impedance elements of said T-networks and constituting with said T- networks the sole external coupling lbetween said input and output circuits, and means for continuously varying the frequency of the generated oscillations, comprising means coupling the resistances of said networks and adapted to cause them to be simultaneously varied and similar means coupling the capacitances of said networks and adapted to cause them to be likewise simultaneously varied.

2. An oscillator comprising an amplier, input and output circuits therefor, a network connected between said circuits, said network comprising a pair of symmetrical T-networks each comprising three impedance elements connected in separate paths between said circuits, the impedance elements of one of said networks consisting of two series resistances and a shunt capacitance, and the impedance elements of the other of said networks consisting of two series capacitances and a shunt resistance, said capacitances and resistances being proportioned to provide maximum attenuation and phase reversal in the network at substantially the preassigned desired frequency, said shunt resistance also having the largest value possible for operation over the desired frequency range, a current responsive thermistor having a very large negative temperature coecient connected across the series impedence elements of said T-networks and constituting with said T-networks the sole external coupling between said circuits, and means acting unitarily on all the impedance elements consisting of at least one .of the two kinds of impedance of said network for continuously adjusting the frequency of the 'generated oscillations.

3. The combination recited in claim 2 in which the two series elements of each T-network are equal, and in which the three elements consisting of at least one of the two kinds of impedance are variable and initially adjusted and ganged so that the series impedances thereof continue equal in value at al1 times and have a constant ratio to the shunt impedance, whereby the frequency may -be adjusted independently of an amplitude adjustment by adjustment ofthe other shunt impedance, said other shunt impedance being capable of such adjustment for a desired amplitude.

4. The combination recited in claim 2 in which the series impedances of each T-network are equal and the three impedance elements consisting oi' at least one of the two kinds of impedance are variable and initially adjusted and ganged so that when varied together the impedances of the series elements maintain equality and a constant relation to the impedance of the corresponding shunt element, whereby the frequency may be changed in such manner that the control of amplitude by the other shunt impedance is independent of frequency, said other shunt impedance being made adjustable for this purpose, the values of all of the impedances of the parallel T-network as a whole being initially adjusted so that the ratio of four times the product of the shunt resistance and one of said series capacitances to the product of one of said series resistances and said shunt capacitance is approximate to but less than 1.

5. The combination recited in claim 2 in which the impedances of the series elements in each said T-network are equal and the three impedance elements of each group comprising like impedance elements are variable and initially adjusted and ganged so that when varied together `the impedances of the series elements maintains equality and a constant relation to the impedance of the shunt element, thereby providing two independent means for frequency variation while making possible the adjustment oi amplitude by the initial adjustment of the shunt resistance, the values of all of the resistances and capacitances being initially adjusted also so that the ratio of four times the product of said resistance and one of said series capacitances to the product of one of said series resistances and said shunt capacitance is approximate to but less than 1.

6. An oscillator comprising an amplifier, input and output circuits therefor, a network connected between said circuits, said network comprising a pair o parallel symmetrical T-networks each comprising three impedance elements, one of said networks consisting of two series resistances and.

a shunt capacitance, and the other or said networks consisting oftwo series capacitances and a shunt resistance, said capacitances and resistances being proportioned to provide maximum attenuation and phase reversal in the network at substantially a preassigned desired frequency, and a current-responsive thermistor having a large negative temperature coeiiicient of resistance connected across the series impedance elements vof said T-networks and constituting with said T- networks the sole external coupling Ibetween said circuits, said amplifier having substantially zero phase shift and a very large transconductance, and in which the impedances oi the two series elements of each T-network are equal and the three impedance elements consisting of at least one of the two kinds of impedance are Varia-ble and initially adjusted and ganged so that the impedances of the series element continue equal in value at all times and have a constant ratio to the impedance of the shunt element whereby the frequency may be adjusted independently oi an amplitude adjustment by variation of the impedance of the other shunt element, the impedance of said other shunt element being so adjustable for amplitude change.

'7. The combination recited in claim 6 in which the impedances Vof the three elements oi each group comprising like impedance elements are variable and initially adjusted and ganged so that the impedances oi the series elements continue equal in value at all times and have a constant ratio to the impedance of the shunt element of the group, whereby the frequency may be adjusted by two independent means independently of the amplitude elect of an adjustment of the' impedance of the shunt element of either T-network, the impedance of the shunt element 'of at least one said T-network being initially adjusted for the desired amplitude value.


Référencé par
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Classification aux États-Unis331/142, 333/171, 331/183, 333/172
Classification internationaleH03B5/22, H03B5/00
Classification coopérativeH03B5/22
Classification européenneH03B5/22