US5867012A - Switching bandgap reference circuit with compounded ΔV.sub.βΕ - Google Patents
Switching bandgap reference circuit with compounded ΔV.sub.βΕ Download PDFInfo
- Publication number
- US5867012A US5867012A US08/907,839 US90783997A US5867012A US 5867012 A US5867012 A US 5867012A US 90783997 A US90783997 A US 90783997A US 5867012 A US5867012 A US 5867012A
- Authority
- US
- United States
- Prior art keywords
- junction
- amplifier
- bandgap reference
- reference circuit
- current source
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired - Lifetime
Links
Images
Classifications
-
- G—PHYSICS
- G05—CONTROLLING; REGULATING
- G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
- G05F3/00—Non-retroactive systems for regulating electric variables by using an uncontrolled element, or an uncontrolled combination of elements, such element or such combination having self-regulating properties
- G05F3/02—Regulating voltage or current
- G05F3/08—Regulating voltage or current wherein the variable is dc
- G05F3/10—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics
- G05F3/16—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices
- G05F3/20—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices using diode- transistor combinations
- G05F3/30—Regulators using the difference between the base-emitter voltages of two bipolar transistors operating at different current densities
Definitions
- This invention relates to a switching bandgap reference circuit with compounded ⁇ V BE .
- Bandgap reference circuits are used to provide a stable reference voltage independent of temperature variation.
- One continuous time bandgap reference circuit known as a Brokaw cell utilizes two matched bipolar transistors operating at different current densities to develop V BE which decreases with temperature and ⁇ V BE which increases with temperature and scales the ⁇ V BE with a resistor network to just offset the V BE decrease to produce the stabilized reference voltage.
- Another continuous time bandgap reference circuit fabricated in CMOS technology makes use of the parasitic substrate bipolar transistors to combine V BE and the offseting ⁇ V BE to obtain the temperature stabilized reference output voltage.
- the transistors must be matched and the output is a high impedance instead of the operational amplifier of the Brokaw cell. In this construction and in the Brokaw cell the continuous nature of the output makes it difficult to correct the inherent amplifier offset.
- a single PN junction is used to develop both the V BE and offsetting ⁇ V BE but it is a switching band gap reference circuit which alternates between two modes: the auto-zero mode in which the output voltage is not temperature stabilized and the valid reference mode in which it is.
- the PN junction is typically a parasitic substrate bipolar transistor in a CMOS circuit.
- the ⁇ V BE is relatively quite small which does not help with signal to noise ratio considerations.
- One way to increase ⁇ V BE is to increase the current density ratio but this only aids up to a point due to the logarithmic nature of ⁇ V BE with respect to current density. Further, when the current density is increased the ability to match the PN junction becomes more difficult.
- the invention results from the realization that an improved switching bandgap reference circuit with increased ⁇ V BE can be effected without increasing the current density ratio by developing ⁇ V BE voltage across a plurality of PN junctions and summing those ⁇ V BE voltages to produce a larger total ⁇ V BE voltage at the input to the amplifier to be scaled to the temperature stabilized output voltage by the feedback network of the amplifier.
- This invention features a switching bandgap reference circuit with compounded ⁇ V BE .
- There is an amplifier having an output, an inverting input and a non-inverting input, a first PN junction connected to the non-inverting input, and a second PN junction connected to the inverting input through an input capacitor.
- a switching device applies in the auto zero mode the low current source to a first terminal of the first PN junction and the high current source to a first terminal of the second PN junction for establishing the V BE , of the first junction at both the inputs of the amplifier and for applying in the valid reference mode the high current source to the first terminal of the first PN junction and the low current source to the first terminal of the second PN junction for establishing the positive ⁇ V BE1 of the first PN junction to both the inputs of the amplifier and applying the negative ⁇ V BE2 of the second PN junction to the input capacitor to produce a voltage of ⁇ V BE1 plus (- ⁇ V BE2 ) across the input capacitor.
- a feedback capacitor is connected between the output and inverting input of the amplifier to define the gain on the combined ⁇ V BE voltages to produce a temperature stabilized voltage at the output of the amplifier.
- a reset switching device discharges the feedback capacitor and enables the amplifier to equalize its inputs in the auto zero mode.
- the amplifier may be an operational amplifier
- the PN junction may be included in the diode or a transistor
- the transistor may be a bipolar transistor.
- the PN junction may be included in a parasitic substrate bipolar transistor in a CMOS circuit.
- the first PN junction may have second terminal connected to a common node to establish the amplifier output voltage relative to that node.
- the collectors of the transistors may be connected to ground.
- FIG. 1 is a schematic diagram of a prior art Brokaw cell bandgap reference circuit
- FIG. 2 is a schematic diagram of a prior art continuous time CMOS bandgap reference circuit
- FIG. 3 is a schematic diagram of a prior art switching bandgap reference circuit
- FIG. 3A illustrates waveforms that occur in the circuit of FIG. 3
- FIG. 4 is a schematic diagram of a switching bandgap reference circuit with compounded ⁇ V BE according to this invention.
- FIG. 4A illustrates waveforms that occur in the circuit of FIG. 4.
- FIG. 5 is a schematic diagram of the circuit of FIG. 4 with amplifier offset cancellation.
- a stable reference voltage is required in many electronic systems and integrated circuits, especially in analog to digital converters where an input voltage is to be measured with respect to a stable voltage and in digital to analog converters where the output voltage is a code-dependent fraction of this reference voltage. This voltage must be generated on-chip and should be independent of temperature.
- bandgap reference One well-known technique for generating such a reference voltage is the so-called bandgap reference.
- This type of circuit utilizes the fact that the base-emitter voltage (V BE ) of a bipolar transistor decreases with increasing temperature in a well-behaved manner. This behavior is known as CTAT or Complementary-to-Absolute Temperature. If a voltage which is PTAT, or Proportional-to-Absolute-Temperature, is generated and added with the correct scaling to the V BE of the bipolar transistor, then the resulting voltage may be made independent of temperature.
- V BE base-emitter voltage
- CTAT Complementary-to-Absolute Temperature
- FIG. 1 shows one prior art example known as the Brokaw cell 10 to illustrate this technique.
- transistors 12 and 14 are operated at equal currents I 1 , I 2 but are of unequal emitter area giving rise to a difference in Base-Emitter voltage which may be written as:
- This ⁇ V BE appears across resistance 20 and determines the operating current of transistor 12 and thereby transistor 14. The sum of these currents flows in resistance 22, developing a voltage which is added to the V BE of transistor 14 and appears at the amplifier 24 output.
- the output voltage may be chosen such as to be independent of temperature as previously discussed. For more detail see A. Paul Brokaw, "A Simple Three Terminal Bandgap Reference", IEEE J. Solid-State Circuits, Vol. SC9, pp. 288-393, Dec. 1994.
- FIG. 2 shows a continuous time circuit 30 using these substrate PNP transistors 32, 34 with the same scaling techniques of FIG. 1 used to produce a ⁇ V BE across resistance 36 and so establish the operating current of the circuit.
- the amplifier 38 sets up the gate-source bias of transistors 40 and 42 so as to provide this operating current.
- a further copy of this current is made by transistor 44.
- This develops a voltage drop across resistance 46 which is added to the V BE of transistor 48 so as to produce a stable bandgap reference voltage.
- This circuit still has the problem of matching between transistors 32 and 34 and also suffers from poor offset voltage in the CMOS input pair of the amplifier 38. Circuits of this type have been used to make bandgap references on CMOS processes but they lack precision.
- FIG. 3 One further prior art example is shown in FIG. 3, described in U.S. Pat. No. 5,563,504, issued to Barrie Gilbert and Shao-Feng Shu, "Switching Bandgap Voltage Reference".
- This utilizes the fact that in circuits such as A/D converters, the reference voltage is not required to be stable at all times and so may be switched from an auto-zero mode to a valid reference mode and back again as required. This enables the use of only one bipolar transistor operated at different switched currents and so eliminates the problem of mismatch.
- the new switched current bandgap reference circuit 80 with compounded ⁇ V BE is shown in FIG. 4.
- the purpose of this circuit is to generate a 2 ⁇ V BE signal from one switching action of the current sources 62, 64.
- Two PNP transistors 66, 68 are now used with currents I and N(I) and the current is switched from one to the other so that as one V BE goes from a low to a high value, the other V BE goes from a high to a low value maintaining a constant overall supply current.
- the resulting two ⁇ V BE signals are summed on capacitor 70 and amplified by amplifier 72 as before to produce the required bandgap voltage at the output of amplifier 72.
- two PNP transistors 66, 68 are now used there is no matching requirement between them. Only the ⁇ V BE of each transistor at the switched currents is summed with the V BE of transistor 66.
- V BE1 (1) and V BE1 (NI) are the V BE of transistor 66 at I and N(I) and V BE2 (1) and V BE2 (N1) are the V BE of transistor 68 at I and NI.
- V BE1 (1) and V BE1 (NI) are the V BE of transistor 66 at I and N(I) and V BE2 (1) and V BE2 (N1) are the V BE of transistor 68 at I and NI.
- the circuit is in auto-zero with the feedback switch 74 closed, shorting feedback capacitor 75, and with switches 76 and 78 set to the left.
- the output is at V BE1 (1), FIG. 4A, as the amplifier 72 is configured as a voltage follower.
- the right hand side of capacitor 70 is also at V BE1 (1).
- the left-hand side of capacitor 70 is at V BE2 (N1).
- Switch 74 is now opened, preserving these initial conditions.
- Switches 76 and 78 are now switched to the right, causing the voltage at the positive terminal of the amplifier 72 to increase by an amount V BE1 (N1) -V BEI (1) !. If the left-hand terminal of capacitor 70 were at a fixed voltage then the output would respond to this signal only and the circuit would default to that of FIG. 3. However, as the positive terminal and thereby the right-hand side of capacitor 70 increases by this ⁇ V BE1 , the current in transistor 68 reduces from NI to I causing a drop in voltage of V BE2 (N1) -V BE2 (1) ! on the left-hand side of capacitor 70. Thus both ⁇ V BE terms are summed and amplified to give the following output: ##EQU2##
- both ⁇ V BE signals are equal and the circuit 60 simply doubles this ⁇ V BE input signal.
- FIG. 5 shows a well-known technique for offset cancellation sometimes known as correlated-double sampling or simply auto-zero.
- auto-zero the feedback switch 74 is closed but the feedback capacitor 75 is no longer connected to the amplifier 72 output as before in FIG. 4. Instead, capacitor 75 is switched to the positive input of the amplifier 72 by switch 80 and so is charged to the amplifier offset voltage.
- switch 74 opens first, storing this offset on capacitor 75. Then capacitor 75 is switched back to the amplifier 72 output by switch 82, cancelling the offset error and finally switches 76 and 78 are switched to create the required ⁇ V BE signals.
- the ratioed current sources are made with a 5 ⁇ 5 array of unit PMOS devices which are laid out with the same care as used in making 8 and 10 bit current-source Digital to Analog converters so the matching performance may be predicted and is well within the requirements of the circuit.
- the array is surrounded by a full ring of dummy devices with the center PMOS device chosen as the single 2 ⁇ A current source.
- the surrounding 24 devices are wired together to give 48 ⁇ A.
Abstract
A switching bandgap reference circuit with compounded ΔVBE includes an amplifier having an output, an inverting input and a non-inverting input; a first PN junction connected to the non-inverting input; a second PN junction connected to the inverting input through an input capacitor; a low current source and a high current source; a switching device for applying in the auto zero mode the low current source to a first terminal of the first PN junction and the high current source to a first terminal of the second PN junction for establishing the VBE, of the first junction at both the inputs of the amplifier and for applying in the valid reference mode the high current source to the first terminal of the first PN junction and the low current source to the first terminal of the second PN junction for establishing the positive ΔVBE, of the first PN junction to both the inputs of the amplifier and applying the negative ΔVBE2 of the second PN junction to the input capacitor to produce a voltage of ΔVBE1 plus (-ΔVBE2) across the input capacitor; a feedback capacitor connected between the output and inverting input of the amplifier to define the gain on the combined ΔVBE voltages to produce a temperature stabilized voltage at the output of the amplifier and a reset switching device for discharging the feedback capacitor and enabling the amplifier to equalize its inputs in the auto zero mode.
Description
This invention relates to a switching bandgap reference circuit with compounded ΔVBE.
Bandgap reference circuits are used to provide a stable reference voltage independent of temperature variation. One continuous time bandgap reference circuit known as a Brokaw cell utilizes two matched bipolar transistors operating at different current densities to develop VBE which decreases with temperature and ΔVBE which increases with temperature and scales the ΔVBE with a resistor network to just offset the VBE decrease to produce the stabilized reference voltage. Another continuous time bandgap reference circuit fabricated in CMOS technology makes use of the parasitic substrate bipolar transistors to combine VBE and the offseting ΔVBE to obtain the temperature stabilized reference output voltage. In the construction the transistors must be matched and the output is a high impedance instead of the operational amplifier of the Brokaw cell. In this construction and in the Brokaw cell the continuous nature of the output makes it difficult to correct the inherent amplifier offset.
In another approach, as disclosed in U.S. Pat. No. 5,563,504, Gilbert et al., a single PN junction is used to develop both the VBE and offsetting ΔVBE but it is a switching band gap reference circuit which alternates between two modes: the auto-zero mode in which the output voltage is not temperature stabilized and the valid reference mode in which it is. The PN junction is typically a parasitic substrate bipolar transistor in a CMOS circuit. In these circuits the ΔVBE is relatively quite small which does not help with signal to noise ratio considerations. One way to increase ΔVBE is to increase the current density ratio but this only aids up to a point due to the logarithmic nature of ΔVBE with respect to current density. Further, when the current density is increased the ability to match the PN junction becomes more difficult.
It is therefore an object of this invention to provide an improved switching bandgap reference circuit.
It is a further object of this invention to provide such a switching bandgap reference circuit which increases the ΔVBE voltage.
It is a further object of this invention to provide such a switching bandgap reference circuit which has an improved signal to noise ratio.
It is a further object of this invention to provide such a switching bandgap reference circuit which increases ΔVBE independent of the current density ratio of the PN junction.
It is a further object of this invention to provide such a switching bandgap reference circuit in which the PN junctions need not be matched.
It is a further object of this invention to provide such a switching bandgap reference circuit in which the amplifier offset error and low frequency noise are easily corrected.
The invention results from the realization that an improved switching bandgap reference circuit with increased ΔVBE can be effected without increasing the current density ratio by developing ΔVBE voltage across a plurality of PN junctions and summing those ΔVBE voltages to produce a larger total ΔVBE voltage at the input to the amplifier to be scaled to the temperature stabilized output voltage by the feedback network of the amplifier.
This invention features a switching bandgap reference circuit with compounded ΔVBE. There is an amplifier having an output, an inverting input and a non-inverting input, a first PN junction connected to the non-inverting input, and a second PN junction connected to the inverting input through an input capacitor. There is a low current source and a high current source. A switching device applies in the auto zero mode the low current source to a first terminal of the first PN junction and the high current source to a first terminal of the second PN junction for establishing the VBE, of the first junction at both the inputs of the amplifier and for applying in the valid reference mode the high current source to the first terminal of the first PN junction and the low current source to the first terminal of the second PN junction for establishing the positive ΔVBE1 of the first PN junction to both the inputs of the amplifier and applying the negative ΔVBE2 of the second PN junction to the input capacitor to produce a voltage of ΔVBE1 plus (-ΔVBE2) across the input capacitor. A feedback capacitor is connected between the output and inverting input of the amplifier to define the gain on the combined ΔVBE voltages to produce a temperature stabilized voltage at the output of the amplifier. A reset switching device discharges the feedback capacitor and enables the amplifier to equalize its inputs in the auto zero mode.
In a preferred embodiment the amplifier may be an operational amplifier, the PN junction may be included in the diode or a transistor, and the transistor may be a bipolar transistor. The PN junction may be included in a parasitic substrate bipolar transistor in a CMOS circuit. The first PN junction may have second terminal connected to a common node to establish the amplifier output voltage relative to that node. The collectors of the transistors may be connected to ground.
Other objects, features and advantages will occur to those skilled in the art from the following description of a preferred embodiment and the accompanying drawings, in which:
FIG. 1 is a schematic diagram of a prior art Brokaw cell bandgap reference circuit;
FIG. 2 is a schematic diagram of a prior art continuous time CMOS bandgap reference circuit;
FIG. 3 is a schematic diagram of a prior art switching bandgap reference circuit;
FIG. 3A illustrates waveforms that occur in the circuit of FIG. 3;
FIG. 4 is a schematic diagram of a switching bandgap reference circuit with compounded ΔVBE according to this invention;
FIG. 4A illustrates waveforms that occur in the circuit of FIG. 4; and
FIG. 5 is a schematic diagram of the circuit of FIG. 4 with amplifier offset cancellation.
A stable reference voltage is required in many electronic systems and integrated circuits, especially in analog to digital converters where an input voltage is to be measured with respect to a stable voltage and in digital to analog converters where the output voltage is a code-dependent fraction of this reference voltage. This voltage must be generated on-chip and should be independent of temperature.
One well-known technique for generating such a reference voltage is the so-called bandgap reference. This type of circuit utilizes the fact that the base-emitter voltage (VBE) of a bipolar transistor decreases with increasing temperature in a well-behaved manner. This behavior is known as CTAT or Complementary-to-Absolute Temperature. If a voltage which is PTAT, or Proportional-to-Absolute-Temperature, is generated and added with the correct scaling to the VBE of the bipolar transistor, then the resulting voltage may be made independent of temperature.
FIG. 1 shows one prior art example known as the Brokaw cell 10 to illustrate this technique. In this example, transistors 12 and 14 are operated at equal currents I1, I2 but are of unequal emitter area giving rise to a difference in Base-Emitter voltage which may be written as:
ΔV.sub.BE =(kT/q)·1n(N)
and is clearly PTAT as it contains the term T, absolute temperature. N is the current density ratio or in this case the area ratio since the currents are equal. This equality is established by making resistance 16 equal to resistance 18. A value of N=8 is shown which may be achieved by scaling emitter area or, more accurately, by use of multiple devices. This ΔVBE appears across resistance 20 and determines the operating current of transistor 12 and thereby transistor 14. The sum of these currents flows in resistance 22, developing a voltage which is added to the VBE of transistor 14 and appears at the amplifier 24 output. Thus by proper choice of N, resistance 20 and resistance 22, the output voltage may be chosen such as to be independent of temperature as previously discussed. For more detail see A. Paul Brokaw, "A Simple Three Terminal Bandgap Reference", IEEE J. Solid-State Circuits, Vol. SC9, pp. 288-393, Dec. 1994.
Two problems exist with this circuit, firstly isolated bipolar transistors are required and so the circuit cannot be implemented in a low cost CMOS process and secondly good VBE matching is required between two transistors operating at different current densities.
In a standard CMOS process one bipolar device type is available. This is the parasitic substrate device which has its collector at the common wafer potential. In an N-well CMOS process this potential is ground in a single supply design. FIG. 2 shows a continuous time circuit 30 using these substrate PNP transistors 32, 34 with the same scaling techniques of FIG. 1 used to produce a ΔVBE across resistance 36 and so establish the operating current of the circuit. The amplifier 38 sets up the gate-source bias of transistors 40 and 42 so as to provide this operating current. A further copy of this current is made by transistor 44. This develops a voltage drop across resistance 46 which is added to the VBE of transistor 48 so as to produce a stable bandgap reference voltage. This circuit still has the problem of matching between transistors 32 and 34 and also suffers from poor offset voltage in the CMOS input pair of the amplifier 38. Circuits of this type have been used to make bandgap references on CMOS processes but they lack precision.
One further prior art example is shown in FIG. 3, described in U.S. Pat. No. 5,563,504, issued to Barrie Gilbert and Shao-Feng Shu, "Switching Bandgap Voltage Reference". This utilizes the fact that in circuits such as A/D converters, the reference voltage is not required to be stable at all times and so may be switched from an auto-zero mode to a valid reference mode and back again as required. This enables the use of only one bipolar transistor operated at different switched currents and so eliminates the problem of mismatch. During the auto-zero mode switch 50 is closed and 52 is open. This sets up the initial conditions of the circuit with Vout =VBE1 at I1. In the valid reference mode, switch 50 opens first, storing VBE1 on capacitor 52. Then switch 52 closes and the emitter 54 of transistor 56 increases to VBE1 at I1 +I2. This change in voltage is amplified by a gain determined by the feedback capacitances 52 and 58 and added to the output such that: ##EQU1## This eliminates the need for good matching between the two bipolar transistors 12 and 14 used in FIG. 1 and transistors 32 and 34 used in FIG. 2 but there remains a problem with amplifier offset and noise as the ΔVBE signal is small. For example with the area ratio 8:1 of transistors 12 and 14, FIG. 1, the ΔVBE signal at room temperature is only 53 mV approximately. This signal can be increased by using a larger ratio in the currents I1 and I2 but with an increase in power dissipation. Also, the effect is not dramatic due to the log term. For example with N=80 rather than 8, ΔVBE =112 mV. The development of Vin and Vout in relation to the clock signal CLK1 and CLK2 that operate switches 50 and 52 are shown in FIGS. 3A.
The new switched current bandgap reference circuit 80 with compounded ΔVBE is shown in FIG. 4. The purpose of this circuit is to generate a 2×ΔVBE signal from one switching action of the current sources 62, 64. Two PNP transistors 66, 68 are now used with currents I and N(I) and the current is switched from one to the other so that as one VBE goes from a low to a high value, the other VBE goes from a high to a low value maintaining a constant overall supply current. The resulting two ΔVBE signals are summed on capacitor 70 and amplified by amplifier 72 as before to produce the required bandgap voltage at the output of amplifier 72. Although two PNP transistors 66, 68 are now used there is no matching requirement between them. Only the ΔVBE of each transistor at the switched currents is summed with the VBE of transistor 66.
For description purposes the various VBE 's are designated as follows: VBE1(1) and VBE1(NI), are the VBE of transistor 66 at I and N(I) and VBE2(1) and VBE2(N1) are the VBE of transistor 68 at I and NI. Initially the circuit is in auto-zero with the feedback switch 74 closed, shorting feedback capacitor 75, and with switches 76 and 78 set to the left. Thus the output is at VBE1(1), FIG. 4A, as the amplifier 72 is configured as a voltage follower. The right hand side of capacitor 70 is also at VBE1(1). The left-hand side of capacitor 70 is at VBE2(N1). Switch 74 is now opened, preserving these initial conditions. Switches 76 and 78 are now switched to the right, causing the voltage at the positive terminal of the amplifier 72 to increase by an amount VBE1(N1) -VBEI(1) !. If the left-hand terminal of capacitor 70 were at a fixed voltage then the output would respond to this signal only and the circuit would default to that of FIG. 3. However, as the positive terminal and thereby the right-hand side of capacitor 70 increases by this ΔVBE1, the current in transistor 68 reduces from NI to I causing a drop in voltage of VBE2(N1) -VBE2(1) ! on the left-hand side of capacitor 70. Thus both ΔVBE terms are summed and amplified to give the following output: ##EQU2##
Under nominal conditions, both ΔVBE signals are equal and the circuit 60 simply doubles this ΔVBE input signal. This improvement is best illustrated with an example: With a unit current source I=2μA and N=24 giving a total switching supply current of 50μA, there is a ΔVBE signal of 81.65 mV at T=25 degrees. The input signal to the amplifier is then 163.3 mV. To achieve the same input signal level with the circuit of FIG. 3 a much larger ratio would be required: 2μA and (24)2 ×2μA giving a total switching current of 1152μA.
As the amplifier 72 is made using MOS input devices, a significant offset voltage must be expected and so offset cancellation circuitry should be included. FIG. 5 shows a well-known technique for offset cancellation sometimes known as correlated-double sampling or simply auto-zero. In auto-zero the feedback switch 74 is closed but the feedback capacitor 75 is no longer connected to the amplifier 72 output as before in FIG. 4. Instead, capacitor 75 is switched to the positive input of the amplifier 72 by switch 80 and so is charged to the amplifier offset voltage. When the circuit switches from auto-zero to the valid reference mode, switch 74 opens first, storing this offset on capacitor 75. Then capacitor 75 is switched back to the amplifier 72 output by switch 82, cancelling the offset error and finally switches 76 and 78 are switched to create the required ΔVBE signals.
The ratioed current sources are made with a 5×5 array of unit PMOS devices which are laid out with the same care as used in making 8 and 10 bit current-source Digital to Analog converters so the matching performance may be predicted and is well within the requirements of the circuit. The array is surrounded by a full ring of dummy devices with the center PMOS device chosen as the single 2μA current source. The surrounding 24 devices are wired together to give 48μA.
Although specific features of this invention are shown in some drawings and not others, this is for convenience only as each feature may be combined with any or all of the other features in accordance with the invention.
Other embodiments will occur to those skilled in the art and are within the following claims:
Claims (9)
1. A switching bandgap reference circuit with compounded ΔVBE, comprising:
an amplifier having an output, an inverting input and a non-inverting input;
a first PN junction connected to said non-inverting input;
a second PN junction connected to said inverting input through an input capacitor;
a low current source and a high current source;
a switching device for applying in an auto zero mode the low current source to a first terminal of said first PN junction and the high current source to a first terminal of said second PN junction for establishing a VBE1 of said first PN junction at both said inputs of said amplifier, and for applying in a valid reference mode the high current source to said first terminal of said first PN junction and the low current source to said first terminal of said second PN junction for establishing a positive ΔVBE1 of said first PN junction to both said inputs of said amplifier and applying a negative ΔVBE2 of said second PN junction to said input capacitor to produce a voltage of ΔVBE1 plus (-ΔVBE2) across said input capacitor;
a feedback capacitor connected between the output and inverting input of said amplifier to define the gain on a combined ΔVBE voltage to produce a temperature stabilized voltage at the output of said amplifier; and
a reset switching device for discharging said feedback capacitor and enabling said amplifier to equalize its inputs in the auto-zero mode.
2. The switching bandgap reference circuit of claim 1 in which said amplifier is an operational amplifier.
3. The switching bandgap reference circuit of claim 1 in which at least one of said first or second PN junctions is included in a diode.
4. The switching bandgap reference circuit of claim 1 in which at least one of said first or second PN junctions is included in a transistor.
5. The switching bandgap reference circuit of claim 1 in which at least one of said first or second PN junctions is included in a bipolar transistor.
6. The switching bandgap reference circuit of claim 1 in which at least one of said first or second PN junctions is included in a parasitic substrate bipolar transistor in a CMOS circuit.
7. The switching bandgap reference circuit of claim 6 in which said transistors include a collector connected to ground.
8. The switching bandgap reference circuit of claim 1 in which said first PN junction has a second terminal connected to a common node to establish the amplifier output voltage relative to that node.
9. The switching bandgap reference circuit of claim 8 in which said common node is ground.
Priority Applications (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US08/907,839 US5867012A (en) | 1997-08-14 | 1997-08-14 | Switching bandgap reference circuit with compounded ΔV.sub.βΕ |
Applications Claiming Priority (1)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US08/907,839 US5867012A (en) | 1997-08-14 | 1997-08-14 | Switching bandgap reference circuit with compounded ΔV.sub.βΕ |
Publications (1)
Publication Number | Publication Date |
---|---|
US5867012A true US5867012A (en) | 1999-02-02 |
Family
ID=25424729
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
US08/907,839 Expired - Lifetime US5867012A (en) | 1997-08-14 | 1997-08-14 | Switching bandgap reference circuit with compounded ΔV.sub.βΕ |
Country Status (1)
Country | Link |
---|---|
US (1) | US5867012A (en) |
Cited By (48)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6031365A (en) * | 1998-03-27 | 2000-02-29 | Vantis Corporation | Band gap reference using a low voltage power supply |
US6107789A (en) * | 1998-10-15 | 2000-08-22 | Lucent Technologies Inc. | Current mirrors |
US6194957B1 (en) * | 1998-10-15 | 2001-02-27 | Lucent Technologies Inc. | Current mirror for preventing an extreme voltage and lock-up |
US6215353B1 (en) * | 1999-05-24 | 2001-04-10 | Pairgain Technologies, Inc. | Stable voltage reference circuit |
US6225856B1 (en) * | 1999-07-30 | 2001-05-01 | Agere Systems Cuardian Corp. | Low power bandgap circuit |
US6288525B1 (en) * | 2000-11-08 | 2001-09-11 | Agere Systems Guardian Corp. | Merged NPN and PNP transistor stack for low noise and low supply voltage bandgap |
US6323801B1 (en) | 1999-07-07 | 2001-11-27 | Analog Devices, Inc. | Bandgap reference circuit for charge balance circuits |
US6529066B1 (en) * | 2000-02-28 | 2003-03-04 | National Semiconductor Corporation | Low voltage band gap circuit and method |
US6538502B2 (en) * | 2000-12-27 | 2003-03-25 | Intel Corporation | High bandwidth switched capacitor input receiver |
US6548991B1 (en) * | 2002-01-19 | 2003-04-15 | National Semiconductor Corporation | Adaptive voltage scaling power supply for use in a digital processing component and method of operating the same |
US20040108887A1 (en) * | 2002-12-09 | 2004-06-10 | Marsh Douglas G. | Low noise resistorless band gap reference |
US6819163B1 (en) | 2003-03-27 | 2004-11-16 | Ami Semiconductor, Inc. | Switched capacitor voltage reference circuits using transconductance circuit to generate reference voltage |
US20050001671A1 (en) * | 2003-06-19 | 2005-01-06 | Rohm Co., Ltd. | Constant voltage generator and electronic equipment using the same |
US20050099752A1 (en) * | 2003-11-08 | 2005-05-12 | Andigilog, Inc. | Temperature sensing circuit |
US20050099163A1 (en) * | 2003-11-08 | 2005-05-12 | Andigilog, Inc. | Temperature manager |
US20050168207A1 (en) * | 2004-01-30 | 2005-08-04 | Analog Devices, Inc. | Voltage source circuit with selectable temperature independent and temperature dependent voltage outputs |
US6930537B1 (en) * | 2002-02-01 | 2005-08-16 | National Semiconductor Corporation | Band-gap reference circuit with averaged current mirror offsets and method |
US7009373B1 (en) | 2004-04-13 | 2006-03-07 | Analog Devices, Inc. | Switched capacitor bandgap reference circuit |
US7161341B1 (en) * | 2004-05-25 | 2007-01-09 | National Semiconductor Corporation | System, circuit, and method for auto-zeroing a bandgap amplifier |
US20070013436A1 (en) * | 2005-06-17 | 2007-01-18 | Yi-Chung Chou | Bandgap reference circuit |
US20070040600A1 (en) * | 2005-08-19 | 2007-02-22 | Fujitsu Limited | Band gap circuit |
US20070072568A1 (en) * | 2005-09-29 | 2007-03-29 | Taner Sumesaglam | High speed receiver |
US20070152740A1 (en) * | 2005-12-29 | 2007-07-05 | Georgescu Bogdan I | Low power bandgap reference circuit with increased accuracy and reduced area consumption |
US20080245237A1 (en) * | 2003-12-30 | 2008-10-09 | Haverstock Thomas B | Coffee infusion press for stackable cups |
US20080259989A1 (en) * | 2007-04-23 | 2008-10-23 | Texas Instruments Incorporated | Systems and Methods for Temperature Measurement Using N-Factor Coefficient Correction |
US20080259999A1 (en) * | 2007-04-23 | 2008-10-23 | Texas Instruments Incorporated | Systems and Methods for Resistance Compensation in a Temperature Measurement Circuit |
US20080258951A1 (en) * | 2007-04-23 | 2008-10-23 | Taxas Instruments Incorporated | Hybrid Delta-Sigma/SAR Analog to Digital Converter and Methods for Using Such |
US20080259997A1 (en) * | 2007-04-23 | 2008-10-23 | Texas Instruments Incorporated | Systems and Methods for PWM Clocking in a Temperature Measurement Circuit |
US20090121701A1 (en) * | 2007-11-08 | 2009-05-14 | Hynix Semiconductor Inc. | Bandgap reference generating circuit |
US20100188141A1 (en) * | 2009-01-26 | 2010-07-29 | Fijitsu Microelectronics Limited | Constant-voltage generating circuit and regulator circuit |
US20110062938A1 (en) * | 2009-09-16 | 2011-03-17 | Patrick Stanley Riehl | Bandgap voltage reference with dynamic element matching |
US7921312B1 (en) | 2007-09-14 | 2011-04-05 | National Semiconductor Corporation | System and method for providing adaptive voltage scaling with multiple clock domains inside a single voltage domain |
US20110260708A1 (en) * | 2010-04-21 | 2011-10-27 | Texas Instruments Incorporated | Bandgap reference circuit and method |
US8151125B1 (en) | 2005-05-23 | 2012-04-03 | National Semiconductor Corporation | System and method for providing multi-point calibration of an adaptive voltage scaling system |
US8421434B2 (en) | 2006-06-02 | 2013-04-16 | Dolpan Audio, Llc | Bandgap circuit with temperature correction |
US20130093748A1 (en) * | 2011-10-12 | 2013-04-18 | Minsu Cho | Organic light emitting diode display device |
US8736354B2 (en) * | 2009-12-02 | 2014-05-27 | Texas Instruments Incorporated | Electronic device and method providing a voltage reference |
CN103986440A (en) * | 2013-02-11 | 2014-08-13 | 全视科技有限公司 | Bandgap reference circuit with offset voltage removal |
CN104375551A (en) * | 2014-11-25 | 2015-02-25 | 无锡中星微电子有限公司 | Band gap voltage generation circuit |
US9013231B1 (en) * | 2013-12-06 | 2015-04-21 | Atmel Corporation | Voltage reference with low sensitivity to package shift |
US9222843B2 (en) | 2003-04-10 | 2015-12-29 | Ic Kinetics Inc. | System for on-chip temperature measurement in integrated circuits |
US20160224146A1 (en) * | 2013-09-27 | 2016-08-04 | Sharon Malevsky | Digital switch-capacitor based bandgap reference and thermal sensor |
US20170060163A1 (en) * | 2014-05-19 | 2017-03-02 | Telefonaktiebolaget Lm Ericsson (Publ) | Method And Apparatus To Minimize Switching Noise Disturbance |
TWI633410B (en) * | 2017-05-12 | 2018-08-21 | 立積電子股份有限公司 | Current mirror device and related amplifier circuit |
US10409312B1 (en) | 2018-07-19 | 2019-09-10 | Analog Devices Global Unlimited Company | Low power duty-cycled reference |
US10528070B2 (en) | 2018-05-02 | 2020-01-07 | Analog Devices Global Unlimited Company | Power-cycling voltage reference |
CN111399575A (en) * | 2019-01-03 | 2020-07-10 | 英飞凌科技奥地利有限公司 | Reference voltage generator |
US20230396223A1 (en) * | 2021-06-06 | 2023-12-07 | Trieye Ltd. | Electronic integration circuit having offset and collected charge reduction circuitries and associated methods |
Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5352972A (en) * | 1991-04-12 | 1994-10-04 | Sgs-Thomson Microelectronics, S.R.L. | Sampled band-gap voltage reference circuit |
US5563504A (en) * | 1994-05-09 | 1996-10-08 | Analog Devices, Inc. | Switching bandgap voltage reference |
US5686823A (en) * | 1996-08-07 | 1997-11-11 | National Semiconductor Corporation | Bandgap voltage reference circuit |
-
1997
- 1997-08-14 US US08/907,839 patent/US5867012A/en not_active Expired - Lifetime
Patent Citations (3)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5352972A (en) * | 1991-04-12 | 1994-10-04 | Sgs-Thomson Microelectronics, S.R.L. | Sampled band-gap voltage reference circuit |
US5563504A (en) * | 1994-05-09 | 1996-10-08 | Analog Devices, Inc. | Switching bandgap voltage reference |
US5686823A (en) * | 1996-08-07 | 1997-11-11 | National Semiconductor Corporation | Bandgap voltage reference circuit |
Cited By (88)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US6031365A (en) * | 1998-03-27 | 2000-02-29 | Vantis Corporation | Band gap reference using a low voltage power supply |
US6107789A (en) * | 1998-10-15 | 2000-08-22 | Lucent Technologies Inc. | Current mirrors |
US6194957B1 (en) * | 1998-10-15 | 2001-02-27 | Lucent Technologies Inc. | Current mirror for preventing an extreme voltage and lock-up |
US6215353B1 (en) * | 1999-05-24 | 2001-04-10 | Pairgain Technologies, Inc. | Stable voltage reference circuit |
US6323801B1 (en) | 1999-07-07 | 2001-11-27 | Analog Devices, Inc. | Bandgap reference circuit for charge balance circuits |
US6225856B1 (en) * | 1999-07-30 | 2001-05-01 | Agere Systems Cuardian Corp. | Low power bandgap circuit |
US6529066B1 (en) * | 2000-02-28 | 2003-03-04 | National Semiconductor Corporation | Low voltage band gap circuit and method |
US6288525B1 (en) * | 2000-11-08 | 2001-09-11 | Agere Systems Guardian Corp. | Merged NPN and PNP transistor stack for low noise and low supply voltage bandgap |
US6538502B2 (en) * | 2000-12-27 | 2003-03-25 | Intel Corporation | High bandwidth switched capacitor input receiver |
US6548991B1 (en) * | 2002-01-19 | 2003-04-15 | National Semiconductor Corporation | Adaptive voltage scaling power supply for use in a digital processing component and method of operating the same |
US7106040B1 (en) | 2002-01-19 | 2006-09-12 | National Semiconductor Corporation | Adaptive voltage scaling power supply for use in a digital processing component and method of operating the same |
US6930537B1 (en) * | 2002-02-01 | 2005-08-16 | National Semiconductor Corporation | Band-gap reference circuit with averaged current mirror offsets and method |
US20040108887A1 (en) * | 2002-12-09 | 2004-06-10 | Marsh Douglas G. | Low noise resistorless band gap reference |
US6864741B2 (en) | 2002-12-09 | 2005-03-08 | Douglas G. Marsh | Low noise resistorless band gap reference |
US6819163B1 (en) | 2003-03-27 | 2004-11-16 | Ami Semiconductor, Inc. | Switched capacitor voltage reference circuits using transconductance circuit to generate reference voltage |
US9222843B2 (en) | 2003-04-10 | 2015-12-29 | Ic Kinetics Inc. | System for on-chip temperature measurement in integrated circuits |
US20050001671A1 (en) * | 2003-06-19 | 2005-01-06 | Rohm Co., Ltd. | Constant voltage generator and electronic equipment using the same |
US7023181B2 (en) * | 2003-06-19 | 2006-04-04 | Rohm Co., Ltd. | Constant voltage generator and electronic equipment using the same |
US20060125461A1 (en) * | 2003-06-19 | 2006-06-15 | Rohm Co., Ltd. | Constant voltage generator and electronic equipment using the same |
US7151365B2 (en) | 2003-06-19 | 2006-12-19 | Rohm Co., Ltd. | Constant voltage generator and electronic equipment using the same |
US20050099163A1 (en) * | 2003-11-08 | 2005-05-12 | Andigilog, Inc. | Temperature manager |
US7857510B2 (en) * | 2003-11-08 | 2010-12-28 | Carl F Liepold | Temperature sensing circuit |
US20050099752A1 (en) * | 2003-11-08 | 2005-05-12 | Andigilog, Inc. | Temperature sensing circuit |
US20080245237A1 (en) * | 2003-12-30 | 2008-10-09 | Haverstock Thomas B | Coffee infusion press for stackable cups |
US20050168207A1 (en) * | 2004-01-30 | 2005-08-04 | Analog Devices, Inc. | Voltage source circuit with selectable temperature independent and temperature dependent voltage outputs |
US7112948B2 (en) | 2004-01-30 | 2006-09-26 | Analog Devices, Inc. | Voltage source circuit with selectable temperature independent and temperature dependent voltage outputs |
US7009373B1 (en) | 2004-04-13 | 2006-03-07 | Analog Devices, Inc. | Switched capacitor bandgap reference circuit |
US7161341B1 (en) * | 2004-05-25 | 2007-01-09 | National Semiconductor Corporation | System, circuit, and method for auto-zeroing a bandgap amplifier |
US8151125B1 (en) | 2005-05-23 | 2012-04-03 | National Semiconductor Corporation | System and method for providing multi-point calibration of an adaptive voltage scaling system |
US20070013436A1 (en) * | 2005-06-17 | 2007-01-18 | Yi-Chung Chou | Bandgap reference circuit |
US20070040600A1 (en) * | 2005-08-19 | 2007-02-22 | Fujitsu Limited | Band gap circuit |
US7236047B2 (en) * | 2005-08-19 | 2007-06-26 | Fujitsu Limited | Band gap circuit |
US7756495B2 (en) | 2005-09-29 | 2010-07-13 | Intel Corporation | High speed receiver |
US20070072568A1 (en) * | 2005-09-29 | 2007-03-29 | Taner Sumesaglam | High speed receiver |
WO2007081634A3 (en) * | 2005-12-29 | 2007-11-08 | Cypress Semiconductor Corp | Low power bandgap reference circuit with increased accuracy and reduced area consumption |
US7683701B2 (en) | 2005-12-29 | 2010-03-23 | Cypress Semiconductor Corporation | Low power Bandgap reference circuit with increased accuracy and reduced area consumption |
WO2007081634A2 (en) * | 2005-12-29 | 2007-07-19 | Cypress Semiconductor Corp. | Low power bandgap reference circuit with increased accuracy and reduced area consumption |
US20070152740A1 (en) * | 2005-12-29 | 2007-07-05 | Georgescu Bogdan I | Low power bandgap reference circuit with increased accuracy and reduced area consumption |
US8941370B2 (en) | 2006-06-02 | 2015-01-27 | Doplan Audio, LLC | Bandgap circuit with temperature correction |
US8421434B2 (en) | 2006-06-02 | 2013-04-16 | Dolpan Audio, Llc | Bandgap circuit with temperature correction |
US9671800B2 (en) | 2006-06-02 | 2017-06-06 | Ol Security Limited Liability Company | Bandgap circuit with temperature correction |
US20080259989A1 (en) * | 2007-04-23 | 2008-10-23 | Texas Instruments Incorporated | Systems and Methods for Temperature Measurement Using N-Factor Coefficient Correction |
US7648271B2 (en) | 2007-04-23 | 2010-01-19 | Texas Instruments Incorporated | Systems and methods for temperature measurement using n-factor coefficient correction |
US7637658B2 (en) | 2007-04-23 | 2009-12-29 | Texas Instruments Incorporated | Systems and methods for PWM clocking in a temperature measurement circuit |
US7524109B2 (en) | 2007-04-23 | 2009-04-28 | Texas Instruments Incorporated | Systems and methods for resistance compensation in a temperature measurement circuit |
US7504977B2 (en) | 2007-04-23 | 2009-03-17 | Texas Instruments Incorporated | Hybrid delta-sigma/SAR analog to digital converter and methods for using such |
US20080259997A1 (en) * | 2007-04-23 | 2008-10-23 | Texas Instruments Incorporated | Systems and Methods for PWM Clocking in a Temperature Measurement Circuit |
US20080258951A1 (en) * | 2007-04-23 | 2008-10-23 | Taxas Instruments Incorporated | Hybrid Delta-Sigma/SAR Analog to Digital Converter and Methods for Using Such |
US20080259999A1 (en) * | 2007-04-23 | 2008-10-23 | Texas Instruments Incorporated | Systems and Methods for Resistance Compensation in a Temperature Measurement Circuit |
US7921312B1 (en) | 2007-09-14 | 2011-04-05 | National Semiconductor Corporation | System and method for providing adaptive voltage scaling with multiple clock domains inside a single voltage domain |
US7834611B2 (en) * | 2007-11-08 | 2010-11-16 | Hynix Semiconductor Inc. | Bandgap reference generating circuit |
US20090121701A1 (en) * | 2007-11-08 | 2009-05-14 | Hynix Semiconductor Inc. | Bandgap reference generating circuit |
US7948304B2 (en) * | 2009-01-26 | 2011-05-24 | Fujitsu Semiconductor Limited | Constant-voltage generating circuit and regulator circuit |
US20100188141A1 (en) * | 2009-01-26 | 2010-07-29 | Fijitsu Microelectronics Limited | Constant-voltage generating circuit and regulator circuit |
US20110062938A1 (en) * | 2009-09-16 | 2011-03-17 | Patrick Stanley Riehl | Bandgap voltage reference with dynamic element matching |
WO2011034501A3 (en) * | 2009-09-16 | 2011-09-09 | Mediatek Singapore Pte. Ltd. | Bandgap voltage reference with dynamic element matching |
US8207724B2 (en) | 2009-09-16 | 2012-06-26 | Mediatek Singapore Pte. Ltd. | Bandgap voltage reference with dynamic element matching |
WO2011034501A2 (en) * | 2009-09-16 | 2011-03-24 | Mediatek Singapore Pte. Ltd. | Bandgap voltage reference with dynamic element matching |
TWI405069B (en) * | 2009-09-16 | 2013-08-11 | Mediatek Singapore Pte Ltd | Voltage reference source and method of providing a reference voltage |
US8736354B2 (en) * | 2009-12-02 | 2014-05-27 | Texas Instruments Incorporated | Electronic device and method providing a voltage reference |
US8324881B2 (en) * | 2010-04-21 | 2012-12-04 | Texas Instruments Incorporated | Bandgap reference circuit with sampling and averaging circuitry |
US20110260708A1 (en) * | 2010-04-21 | 2011-10-27 | Texas Instruments Incorporated | Bandgap reference circuit and method |
US9117404B2 (en) * | 2011-10-12 | 2015-08-25 | Lg Display Co., Ltd. | Organic light emitting diode display device |
US20130093748A1 (en) * | 2011-10-12 | 2013-04-18 | Minsu Cho | Organic light emitting diode display device |
CN103986440A (en) * | 2013-02-11 | 2014-08-13 | 全视科技有限公司 | Bandgap reference circuit with offset voltage removal |
CN103986440B (en) * | 2013-02-11 | 2017-07-04 | 豪威科技股份有限公司 | With the bandgap reference circuit that offset voltage is removed |
US10712875B2 (en) * | 2013-09-27 | 2020-07-14 | Intel Corporation | Digital switch-capacitor based bandgap reference and thermal sensor |
US20160224146A1 (en) * | 2013-09-27 | 2016-08-04 | Sharon Malevsky | Digital switch-capacitor based bandgap reference and thermal sensor |
US9013231B1 (en) * | 2013-12-06 | 2015-04-21 | Atmel Corporation | Voltage reference with low sensitivity to package shift |
US9501078B2 (en) | 2013-12-06 | 2016-11-22 | Atmel Corporation | Voltage reference with low sensitivty to package shift |
US20170060163A1 (en) * | 2014-05-19 | 2017-03-02 | Telefonaktiebolaget Lm Ericsson (Publ) | Method And Apparatus To Minimize Switching Noise Disturbance |
US10429875B2 (en) * | 2014-05-19 | 2019-10-01 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus to minimize switching noise disturbance |
US9904309B2 (en) * | 2014-05-19 | 2018-02-27 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus to minimize switching noise disturbance |
US20180181156A1 (en) * | 2014-05-19 | 2018-06-28 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus to minimize switching noise disturbance |
US10678288B2 (en) | 2014-05-19 | 2020-06-09 | Telefonaktiebolaget Lm Ericsson (Publ) | Method and apparatus to minimize switching noise disturbance |
CN104375551B (en) * | 2014-11-25 | 2017-01-04 | 无锡中感微电子股份有限公司 | Band gap voltage generative circuit |
CN104375551A (en) * | 2014-11-25 | 2015-02-25 | 无锡中星微电子有限公司 | Band gap voltage generation circuit |
CN108874019B (en) * | 2017-05-12 | 2020-07-31 | 立积电子股份有限公司 | Current mirror device and related amplifying circuit |
US10353421B2 (en) | 2017-05-12 | 2019-07-16 | Richwave Technology Corp. | Current mirror device and related amplifier circuit |
CN108874019A (en) * | 2017-05-12 | 2018-11-23 | 立积电子股份有限公司 | Current Mirror Device and Related Amplifying Circuit |
TWI633410B (en) * | 2017-05-12 | 2018-08-21 | 立積電子股份有限公司 | Current mirror device and related amplifier circuit |
US10528070B2 (en) | 2018-05-02 | 2020-01-07 | Analog Devices Global Unlimited Company | Power-cycling voltage reference |
US10409312B1 (en) | 2018-07-19 | 2019-09-10 | Analog Devices Global Unlimited Company | Low power duty-cycled reference |
CN111399575A (en) * | 2019-01-03 | 2020-07-10 | 英飞凌科技奥地利有限公司 | Reference voltage generator |
US10852758B2 (en) | 2019-01-03 | 2020-12-01 | Infineon Technologies Austria Ag | Reference voltage generator |
CN111399575B (en) * | 2019-01-03 | 2023-02-24 | 英飞凌科技奥地利有限公司 | Reference voltage generator |
US20230396223A1 (en) * | 2021-06-06 | 2023-12-07 | Trieye Ltd. | Electronic integration circuit having offset and collected charge reduction circuitries and associated methods |
US11923815B2 (en) * | 2021-06-06 | 2024-03-05 | Trieye Ltd. | Electronic integration circuit having offset and collected charge reduction circuitries and associated methods |
Similar Documents
Publication | Publication Date | Title |
---|---|---|
US5867012A (en) | Switching bandgap reference circuit with compounded ΔV.sub.βΕ | |
US6362612B1 (en) | Bandgap voltage reference circuit | |
US6885178B2 (en) | CMOS voltage bandgap reference with improved headroom | |
JP3586073B2 (en) | Reference voltage generation circuit | |
US5982221A (en) | Switched current temperature sensor circuit with compounded ΔVBE | |
US7164260B2 (en) | Bandgap reference circuit with a shared resistive network | |
US7078958B2 (en) | CMOS bandgap reference with low voltage operation | |
US6710641B1 (en) | Bandgap reference circuit for improved start-up | |
US7088085B2 (en) | CMOS bandgap current and voltage generator | |
JP3647468B2 (en) | Dual source for constant current and PTAT current | |
CA1178338A (en) | Switched capacitor temperature independent bandgap reference | |
US7170336B2 (en) | Low voltage bandgap reference (BGR) circuit | |
US5568045A (en) | Reference voltage generator of a band-gap regulator type used in CMOS transistor circuit | |
US4935690A (en) | CMOS compatible bandgap voltage reference | |
US5686823A (en) | Bandgap voltage reference circuit | |
EP1235132A2 (en) | Reference current circuit and reference voltage circuit | |
EP1966669A2 (en) | Low power bandgap reference circuit with increased accuracy and reduced area consumption | |
GB2425419A (en) | An overtemperature detector for integrated circuits, using hysteresis | |
JP2007518173A (en) | Low offset band gap voltage reference | |
US6242897B1 (en) | Current stacked bandgap reference voltage source | |
US7675353B1 (en) | Constant current and voltage generator | |
CN110895423B (en) | System and method for proportional to absolute temperature circuit | |
US5051686A (en) | Bandgap voltage reference | |
US6380723B1 (en) | Method and system for generating a low voltage reference | |
US5760639A (en) | Voltage and current reference circuit with a low temperature coefficient |
Legal Events
Date | Code | Title | Description |
---|---|---|---|
AS | Assignment |
Owner name: ANALOG DEVICES, INC., MASSACHUSETTS Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:TUTHILL, MICHAEL G.;REEL/FRAME:008748/0659 Effective date: 19970807 |
|
STCF | Information on status: patent grant |
Free format text: PATENTED CASE |
|
FPAY | Fee payment |
Year of fee payment: 4 |
|
REMI | Maintenance fee reminder mailed | ||
FPAY | Fee payment |
Year of fee payment: 8 |
|
SULP | Surcharge for late payment |
Year of fee payment: 7 |
|
FPAY | Fee payment |
Year of fee payment: 12 |