WO1982002806A1 - Switched capacitor bandgap reference - Google Patents

Switched capacitor bandgap reference Download PDF

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Publication number
WO1982002806A1
WO1982002806A1 PCT/US1982/000093 US8200093W WO8202806A1 WO 1982002806 A1 WO1982002806 A1 WO 1982002806A1 US 8200093 W US8200093 W US 8200093W WO 8202806 A1 WO8202806 A1 WO 8202806A1
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WO
WIPO (PCT)
Prior art keywords
voltage
coupled
emitter
circuit
transistor
Prior art date
Application number
PCT/US1982/000093
Other languages
French (fr)
Inventor
Inc Motorola
Richard Walter Ulmer
Roger A Whatley
Original Assignee
Inc Motorola
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Inc Motorola filed Critical Inc Motorola
Priority to DE8282900750T priority Critical patent/DE3273265D1/en
Publication of WO1982002806A1 publication Critical patent/WO1982002806A1/en
Priority to SG759/88A priority patent/SG75988G/en
Priority to HK17289A priority patent/HK17289A/en

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Classifications

    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F3/00—Non-retroactive systems for regulating electric variables by using an uncontrolled element, or an uncontrolled combination of elements, such element or such combination having self-regulating properties
    • G05F3/02—Regulating voltage or current
    • G05F3/08—Regulating voltage or current wherein the variable is dc
    • G05F3/10—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics
    • G05F3/16—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices
    • G05F3/20—Regulating voltage or current wherein the variable is dc using uncontrolled devices with non-linear characteristics being semiconductor devices using diode- transistor combinations
    • G05F3/30—Regulators using the difference between the base-emitter voltages of two bipolar transistors operating at different current densities

Definitions

  • This invention relates generally to bandgap reference circuits and more particularly to CMOS bandgap reference circuits.
  • the best reference for a good reproducible, stable voltage below three volts has been the bandgap reference circuit.
  • the base to emitter voltage V be of a bipolar transistor exhibits a negative temperature coefficient with respect to temperature.
  • Such temperature stable references have been created by generating a V be and summing a ⁇ V be of such value that the sum substantially equals the bandgap voltage of 1.205 volts.
  • a standard CMOS process can be used to fabricate open emitter NPN bipolar transistors for use in a bandgap reference circuit such as that taught in U.S. Patent Application No. 034513.
  • amplifying means such as an operational amplifier
  • two transistors of varying current density were used as emitter followers having resistors in their emitter circuits from which a differential voltage was obtained.
  • An output voltage having a positive, negative or zero coefficient was thereby produced.
  • CMOS circuit affected the initial tolerance variation and temperature variation of the bandgap voltage.
  • the dominant initial tolerance error was caused by the offset voltage associated with the operational amplifier being multiplied by the ratio of two resistors in the emitter circuit of the transistor with lowest current density. Further disadvantages of the prior art are problems with P-resistor matching and a 2:1 variation in the P-resistivity over temperature. Previous CMOS bandgap circuits also required a startup circuit.
  • a first and a second substrate bipolar transistor wherein the emitter area of the first transistor is much larger than the emitter area of the second transistor. Since the second transistor is operated at a higher current density than the first transistor, the V be of the second transistor is greater than the V be of the first transistor.
  • switched capacitors coupled to the emitters of the transistors, the base to emitter voltages of the devices are sampled. When the difference between the two sampled voltages are added in the correct proportion, the result is a voltage with a substantially zero temperature coefficient.
  • Figure 1 is a schematic diagram illustrating one preferred embodiment of the invention.
  • Figure 2 is a graphic timing diagram for the schematic embodiment shown in Figure 1.
  • Figure 3 is a schematic diagram illustrating another embodiment of the amplifier used in the present invention.
  • Figure 4 is a graphic timing diagram for the schematic embodiment shown in Figure 3.
  • the bandgap reference circuit 10 is comprised generally of first and second bipolar transistors 12 and 14, respectively, a clock circuit 16, a first switched capacitance circuit 18, a second switched capacitance circuit 20, and an amplifier circuit 22.
  • Each of the first and second bipolar transistors 12 and 14 has the collector thereof connected to a positive supply V dd , the base thereof connected to a common reference voltage, say analog ground V ag , and the emitter thereof connected to a negative supply V ss via respective current sources 24 and 26.
  • the current sources 24 and 26 are constructed to sink a predetermined ratio of currents, and transistor 12 is fabricated with a larger emitter area than the transistor 14. Since the transistors 12 and 14 are biased at different current densities they will thus develop different base-to-emitter voltages, V be . Because the transistors 12 and 14 are connected as emitter followers, the preferred embodiment may be fabricated using the substrate NPN in a standard CMOS process.
  • a capacitor 28 has an input connected via switches 30 and 32 to the common reference voltage V ag and the emitter of transistor 14, respectively.
  • a capacitor 34 has an input connected via switches 36 and 38 to the emitter of transistors 12 and 14, respectively.
  • Capacitors 28 and 34 have the outputs thereof connected to a node 40.
  • switches 30, 32, 36 and 38 are CMOS transmission gates which are clocked in a conventional manner by the clock circuit 16.
  • Switches 30 and 36 are constructed to be conductive when a clock signal A applied to the control inputs thereof is at a high state, and non-conductive when the clock signal A is at a low state.
  • switches 32 and 38 are preferably constructed to be conductive when a clock signal B applied to the control inputs thereof is at a high state and non-conductive when the clock signal B is at a low state.
  • switches 30 and 32 will cooperate to charge capacitor 28 alternately to the base voltage of transistor 14 and the emitter voltage of transistor 14, thus providing a charge related to V be of transistor 14.
  • switches 36 and 38 cooperated to charge capacitor 34 alternately to the emitter voltage of transistor 12 and the emitter voltage of transistor 14, thus providing a charge related to the difference between the base to emitter voltages, i.e., the ⁇ V be , of the transistors 12 and 14.
  • the voltage, V be will exhibit a negative temperature coefficient (NTC) .
  • NTC negative temperature coefficient
  • PTC positive temperature coefficient
  • an operational amplifier 42 has its negative input coupled to node 40 and its positive input coupled to the reference voltage V ag .
  • a feedback capacitor 44 is coupled between the output of operational amplifier 42 at node 46 and the negative input of the operational amplifier at node 40.
  • a switch 48 is coupled across feedback capacitor 44 with the control input thereof coupled to clock signal C provided by clock circuit 16. By periodically closing switch 48, the operational amplifier 42 is placed in unity gain, and any charge on capacitor 44 is removed.
  • the clock circuit 16 initially provides the clock signal A in a high state to close switches 30 and 36, and clock signal B in a low state to open switches 32 and 38. Simultaneously, the clock circuit 16 provides the clock signal C in a high state to close the switch 48.
  • feedback capacitor 44 is discharged, and, ignoring any amplifier offset, capacitors 28 and 34 are charged to the reference voltage, V ag , and the V be of the transistor 12, respectively.
  • the clock circuit 16 opens switch 48 by providing the clock signal C in a low state. Shortly thereafter, but still before the end of the precharge period, the clock 16 opens switches 30 and 36 by providing the clock signal A in the low state.
  • the clock circuit 16 closes switches 32 and 38 by providing the clock signal B in the high state.
  • the voltage on the terminals of capacitor 28 changes by -V be of transistor
  • this positive bandgap reference voltage, +V ref is made substantially temperature independent by making the ratio of capacitors 28 and 34 equal to the ratio of the temperature coefficients of ⁇ V be and V be .
  • a negative bandgap reference voltage, -V ref may be obtained by inverting clock signal C so that the precharge and valid output reference periods are reversed.
  • FIG 3 illustrates in schematic form, a modified form of amplifier circuit 22' which can be substituted for the amplifier circuit 22 of Figure 1 to substantially eliminate the offset voltage error.
  • Amplifier circuit 22' is comprised of the operational amplifier 42 which has its positive input coupled in parallel to feedback capacitor 44 and periodically discharges the feedback capacitor. However, one terminal of the feedback capacitor 44 is now connected via a switch 52 to the output of the operational amplifier 42 at node 46. Capacitor 44 is also coupled to an input signal, V IN , at node 40.
  • an offset storage capacitor 54 is coupled between node 40 and the negative input terminal of operational amplifier 42, and a switch 56 is connected between node 40 and the reference voltage V ag .
  • the clock circuit 16' generates the additional clock signals D and E, as shown in Figure 4 for controlling the switches 56 and 50, respectively, with the inverse of clock signal D controlling switch 52.
  • the bandgap reference circuit 10 has three distinct periods of operation. During the precharge period, the clock circuit 16' provides clock signals C, D, and E in the high state to close switches 48, 56 and 50 and open switch 52. During this period, capacitor 44 is discharged by switch 48.
  • the operational amplifier 42 is placed in unity gain by switch 50, and the offset storage capacitor 54 is charged to the offset voltage, V os , of the operational amplifier 42.
  • the clock circuit 16' Near the end of the precharge period, the clock circuit 16' provides clock signal E in the low state to open switch 50, leaving capacitor 54 charged to the offset voltage of the oeprational amplifier 42.
  • the clock circuit 16' provides clock signal D in the low state to open switch 56 and close switch 52. Since this switching event tends to disturb the input node 40, a short settling time is preferably provided before clock circuit 16' provides clock signal C in the low state to open switch 48. Thereafter, the charge stored on feedback capacitor 44 will be changed only by a quantity of charge coupled from the switched capacitor sections 13 and 20.
  • the reference voltage developed on the node 46 will be substantially free of any offset voltage error. If the offset capacitor 54 is periodically charged to the offset voltage, V os , the operational amplifier 42 is effectively autozeroed, with node 40 being the zero-offset input node.

Abstract

A temperature stable bandgap voltage reference source (10) utilizing two substrate bipolar transistors (12 and 14) biased at different emitter current densities. Switched capacitors (28 and 34) are used to input the Vbe and the (Alpha)Vbe of the transistors (12 and 14) (NTC and PTC voltages, respectively) into an amplifier (42) to provide a reference voltage VREF proportional to the weighted sum of the PTC and NTC voltages. Proper selection of the ratio of the switched capacitors (28 and 34) renders the reference voltage VREF substantially independent of temperature. In a modified form of the reference (10), the reference amplifier (22) is implemented by an auto-zeroed operational amplifier (42) which uses switched capacitor techniques and an integrated capacitor (44) to achieve the auto-zeroing function.

Description

SWITCHED CAPACITOR BANDGAP REFERENCE
BACKGROUND OF THE INVENTION
Field of the Invention
This invention relates generally to bandgap reference circuits and more particularly to CMOS bandgap reference circuits.
Description of the Prior Art
Typically, the best reference for a good reproducible, stable voltage below three volts has been the bandgap reference circuit. As discussed in Analysis and Design of Analog Integrated Circuits by Paul R. Gray and Robert G. Meyer (John Wiley and Sons, 1977, pp. 239-261), the base to emitter voltage Vbe, of a bipolar transistor exhibits a negative temperature coefficient with respect to temperature. Thus, the sum of the base to emitter voltage, Vbe, of a bipolar transistor and a differential voltage ΔVbe win be relatively independent of temperature when the sum voltage equals the energy gap of silicon. Such temperature stable references have been created by generating a Vbe and summing a ΔVbe of such value that the sum substantially equals the bandgap voltage of 1.205 volts.
A standard CMOS process can be used to fabricate open emitter NPN bipolar transistors for use in a bandgap reference circuit such as that taught in U.S. Patent Application No. 034513. To create a stable temperature independent CMOS bandgap voltage with amplifying means, such as an operational amplifier, two transistors of varying current density were used as emitter followers having resistors in their emitter circuits from which a differential voltage was obtained. An output voltage having a positive, negative or zero coefficient was thereby produced. Several factors in the CMOS circuit, however, affected the initial tolerance variation and temperature variation of the bandgap voltage. The dominant initial tolerance error was caused by the offset voltage associated with the operational amplifier being multiplied by the ratio of two resistors in the emitter circuit of the transistor with lowest current density. Further disadvantages of the prior art are problems with P-resistor matching and a 2:1 variation in the P-resistivity over temperature. Previous CMOS bandgap circuits also required a startup circuit.
Summary of the Invention
It is an object of the present invention to provide a bandgap reference utilizing substrate bipolar transistors and MOS transistors to provide a reference voltage which is substantially temperature stable and substantially independent of process variations.
It is a further object of the invention to provide a bandgap reference fabricated using a standard CMOS process and switched capacitor techniques, which sums the Vbe and ΔVbe of substrate bipolar transistors to derive a near zero temperature coefficient reference voltage.
According to an (P. A.) aspect of the invention, there are provided a first and a second substrate bipolar transistor wherein the emitter area of the first transistor is much larger than the emitter area of the second transistor. Since the second transistor is operated at a higher current density than the first transistor, the Vbe of the second transistor is greater than the Vbe of the first transistor. Using switched capacitors coupled to the emitters of the transistors, the base to emitter voltages of the devices are sampled. When the difference between the two sampled voltages are added in the correct proportion, the result is a voltage with a substantially zero temperature coefficient. The above and other objects, features and advantages of the present invention will be more clearly understood from the following detailed description taken in conjunction with the accompanying drawings.
Brief Description of the Drawings
Figure 1 is a schematic diagram illustrating one preferred embodiment of the invention. Figure 2 is a graphic timing diagram for the schematic embodiment shown in Figure 1.
Figure 3 is a schematic diagram illustrating another embodiment of the amplifier used in the present invention. Figure 4 is a graphic timing diagram for the schematic embodiment shown in Figure 3.
Description of the Preferred Embodiment
Shown in Figure 1, is a switched capacitor bandgap reference circuit 10 constructed in accordance with the preferred embodiment of this invention. The bandgap reference circuit 10 is comprised generally of first and second bipolar transistors 12 and 14, respectively, a clock circuit 16, a first switched capacitance circuit 18, a second switched capacitance circuit 20, and an amplifier circuit 22.
Each of the first and second bipolar transistors 12 and 14 has the collector thereof connected to a positive supply Vdd , the base thereof connected to a common reference voltage, say analog ground Vag, and the emitter thereof connected to a negative supply Vss via respective current sources 24 and 26. In the preferred form, the current sources 24 and 26 are constructed to sink a predetermined ratio of currents, and transistor 12 is fabricated with a larger emitter area than the transistor 14. Since the transistors 12 and 14 are biased at different current densities they will thus develop different base-to-emitter voltages, Vbe. Because the transistors 12 and 14 are connected as emitter followers, the preferred embodiment may be fabricated using the substrate NPN in a standard CMOS process.
In the first switched capacitance circuit 18, a capacitor 28 has an input connected via switches 30 and 32 to the common reference voltage Vag and the emitter of transistor 14, respectively. In the second switched capaci- tance circuit 20, a capacitor 34 has an input connected via switches 36 and 38 to the emitter of transistors 12 and 14, respectively. Capacitors 28 and 34 have the outputs thereof connected to a node 40. In the preferred embodiment, switches 30, 32, 36 and 38 are CMOS transmission gates which are clocked in a conventional manner by the clock circuit 16. Switches 30 and 36 are constructed to be conductive when a clock signal A applied to the control inputs thereof is at a high state, and non-conductive when the clock signal A is at a low state. In contrast, switches 32 and 38 are preferably constructed to be conductive when a clock signal B applied to the control inputs thereof is at a high state and non-conductive when the clock signal B is at a low state.
In this configuration, switches 30 and 32 will cooperate to charge capacitor 28 alternately to the base voltage of transistor 14 and the emitter voltage of transistor 14, thus providing a charge related to Vbe of transistor 14. Simultaneously, switches 36 and 38 cooperated to charge capacitor 34 alternately to the emitter voltage of transistor 12 and the emitter voltage of transistor 14, thus providing a charge related to the difference between the base to emitter voltages, i.e., the ΔVbe, of the transistors 12 and 14. As will be clear to those skilled in the art, the voltage, Vbe, will exhibit a negative temperature coefficient (NTC) . On the other hand, it is well known that the voltage ΔVbe exhibits a positive temperature coefficient (PTC). Thus, it will be clear that the weighted sum of these voltages, Vbe + kΔVbe, where K = C34/C28 may be made substantially temperature independent by appropriate selection of the ratio of capacitors 28 and 34.
In the amplifier circuit 22, an operational amplifier 42 has its negative input coupled to node 40 and its positive input coupled to the reference voltage Vag. A feedback capacitor 44 is coupled between the output of operational amplifier 42 at node 46 and the negative input of the operational amplifier at node 40. In the preferred form, a switch 48 is coupled across feedback capacitor 44 with the control input thereof coupled to clock signal C provided by clock circuit 16. By periodically closing switch 48, the operational amplifier 42 is placed in unity gain, and any charge on capacitor 44 is removed.
As shown in Figure 2, the clock circuit 16 initially provides the clock signal A in a high state to close switches 30 and 36, and clock signal B in a low state to open switches 32 and 38. Simultaneously, the clock circuit 16 provides the clock signal C in a high state to close the switch 48. During this precharge period, feedback capacitor 44 is discharged, and, ignoring any amplifier offset, capacitors 28 and 34 are charged to the reference voltage, Vag, and the Vbe of the transistor 12, respectively. A short time before the end of the precharge period, the clock circuit 16 opens switch 48 by providing the clock signal C in a low state. Shortly thereafter, but still before the end of the precharge period, the clock 16 opens switches 30 and 36 by providing the clock signal A in the low state. At the end of the precharge period and the start of a valid output reference period, the clock circuit 16 closes switches 32 and 38 by providing the clock signal B in the high state. At this time, the voltage on the terminals of capacitor 28 changes by -Vbe of transistor
14 and the voltage on the terminals of capacitor 34 changes by the difference between the base to emitter voltages of the transistors 12 and 14, (Vbe 12 - Vbe14). This switching event causes an amount of charge Q = Vbe14C28 + (Vbe12 - Vbe14)C34 to be transferred to capacitor 44 resulting in an output voltage of Vref = -1/C44 [-Vbe14C28 + (Vbe12 - Vbe14) C34] on node 46. In the preferred form, this positive bandgap reference voltage, +Vref is made substantially temperature independent by making the ratio of capacitors 28 and 34 equal to the ratio of the temperature coefficients of ΔVbe and Vbe. If desired, a negative bandgap reference voltage, -Vref, may be obtained by inverting clock signal C so that the precharge and valid output reference periods are reversed.
In general, the accuracy of the bandgap circuit 10 will be adversely affected by the offset voltage of the operational amplifier. Figure 3 illustrates in schematic form, a modified form of amplifier circuit 22' which can be substituted for the amplifier circuit 22 of Figure 1 to substantially eliminate the offset voltage error. Amplifier circuit 22' is comprised of the operational amplifier 42 which has its positive input coupled in parallel to feedback capacitor 44 and periodically discharges the feedback capacitor. However, one terminal of the feedback capacitor 44 is now connected via a switch 52 to the output of the operational amplifier 42 at node 46. Capacitor 44 is also coupled to an input signal, VIN, at node 40. In addition, an offset storage capacitor 54 is coupled between node 40 and the negative input terminal of operational amplifier 42, and a switch 56 is connected between node 40 and the reference voltage Vag. In this embodiment, the clock circuit 16' generates the additional clock signals D and E, as shown in Figure 4 for controlling the switches 56 and 50, respectively, with the inverse of clock signal D controlling switch 52. In this configuration, the bandgap reference circuit 10 has three distinct periods of operation. During the precharge period, the clock circuit 16' provides clock signals C, D, and E in the high state to close switches 48, 56 and 50 and open switch 52. During this period, capacitor 44 is discharged by switch 48. The operational amplifier 42 is placed in unity gain by switch 50, and the offset storage capacitor 54 is charged to the offset voltage, Vos, of the operational amplifier 42. Near the end of the precharge period, the clock circuit 16' provides clock signal E in the low state to open switch 50, leaving capacitor 54 charged to the offset voltage of the oeprational amplifier 42. A short time thereafter, the clock circuit 16' provides clock signal D in the low state to open switch 56 and close switch 52. Since this switching event tends to disturb the input node 40, a short settling time is preferably provided before clock circuit 16' provides clock signal C in the low state to open switch 48. Thereafter, the charge stored on feedback capacitor 44 will be changed only by a quantity of charge coupled from the switched capacitor sections 13 and 20. During this third period of circuit operation, labeled the valid output reference period, the reference voltage developed on the node 46 will be substantially free of any offset voltage error. If the offset capacitor 54 is periodically charged to the offset voltage, Vos, the operational amplifier 42 is effectively autozeroed, with node 40 being the zero-offset input node.
While the invention has been described in the context of a preferred embodiment, it will be apparent to those skilled in the art that the present invention may be modified in numerous ways and may assume many embodiments other than that specifically set out and described above. Accordingly, it is intended by the appended claims to cover all modifications of the invention which fall within the true spirit and scope of the invention.

Claims

1. A circuit 22 for producing a substantially temperature independent reference voltage VREF, the circuit 22 comprising: first and second bipolar transistor means 12 and 14 having a predetermined base voltage and biased at different current densities to develop first and second emitter voltages, respectively, on the emitters thereof; clock means 16 for alternately providing first and second non-overlapping clock signals; first switched capacitance means 28 coupled to said base voltage in response to the first clock signal A and to said first emitter voltage in response to the second clock signal B, for providing a first charge related to the Vbe of the first transistor means; second switched capacitance means 34 coupled to said second emitter voltage in response to the first clock signal A and to said first emitter voltage in response to the second clock signal B, for providing a second charge related to the difference in the Vbe of the first and second bipolar transistor means 12 and 14; and amplifier means 22 coupled to the first and second switched capacitance means 28 and 34 for providing a reference voltage VREF proportional to the sum of the first and second charges.
2. The circuit 22 of claim 1 wherein each of said switched capacitance means 28 and 34 comprises a capacitor and switching means 30, 32 and 36, 38, respectively, responsive to said clock signals A and B.
3. The circuit 22 of claim 2 wherein said amplifier means 22 comprise an operational amplifier 42, a feedback capacitor 44, and switching means 48 for periodically coupling the input and output portions of the feedback capacitor.
4. A method of producing a substantially temperature independent reference voltage VREF comprising the steps of: biasing first and second bipolar transistor means 14 and 12, respectively, having the same predetermined base voltage, at different current densities to develop first and second emitter voltages; providing first and second non-overlapping clock signals A and B; coupling an input portion of first capacitance means 28 to said base voltage in response to the first clock signal and to the first emitter voltage in response to the second clock signal B, whereby an output portion of said first capacitance means 28 couples a first charge related to the Vbe of the first transistor means; coupling an input portion of second capacitance means 34 to said second emitter voltage in response to the first clock signal A and to said first emitter voltage in response to the second clock signal B, whereby an output portion of said second capacitance means 34 couples a second charge related to the difference in the Vbe of the first and second transistor means 14 and 12, respectively; and amplifying the sum of the charges coupled from the output portions of the first and second capacitance means 28 and 34 to provide a reference voltage VREF Proportional to the sum of the first and second charges.
5. A circuit 22 for producing a substantially temperature independent reference voltage VREF, the circuit 22 comprising: first and second transistors 14 and 12 having the bases thereof coupled to a predetermined bias voltage VAG, the collectors coupled to a positive supply VDD and the emitters thereof open; biasing means 24 and 26 coupled between the emitters of the first and second transistors 14 and 12 and a negative supply VSS biasing said first and second transistors 14 and 12 at different current densities; a first capacitor 28 having a first portion coupled alternately to the predetermined bias voltage VAG amd emitter of the first transistor 14, for providing a first charge related to the Vbe of the first transistor 14; a second capacitor 34 having a first portion coupled alternately to the emitter of the first transistor 14 and the emitter of the second transistor 12, for providing a second charge related to the difference in the Vbe of the first and second transistors 14 and 12; and an amplifier 22 coupled to the first and second capacitors 28 and 34 for providing a reference voltage VREF proportional to the sum of the first and second charges.
6. The circuit 22 of claim 5 wherein the first portions of the first and second capacitors 28 and 34 are alternately coupled to the first and second transistors 14 and 12 by clocked switches in response to non-overlapping clock signals A and B.
7. The circuit 22 of claim 5 wherein the amplifier 22 comprises an operational amplifier 42, a feedback capacitor 44, and switching means 48 for periodically discharging the feedback capacitor.
PCT/US1982/000093 1981-02-03 1982-01-25 Switched capacitor bandgap reference WO1982002806A1 (en)

Priority Applications (3)

Application Number Priority Date Filing Date Title
DE8282900750T DE3273265D1 (en) 1981-02-03 1982-01-25 Switched capacitor bandgap reference
SG759/88A SG75988G (en) 1981-02-03 1988-11-15 Switched capacitor bandgap reference
HK17289A HK17289A (en) 1981-02-03 1989-03-02 Switched capacitor bandgap reference

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
US06/231,073 US4375595A (en) 1981-02-03 1981-02-03 Switched capacitor temperature independent bandgap reference
US231073810203 1981-02-03

Publications (1)

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WO1982002806A1 true WO1982002806A1 (en) 1982-08-19

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US (1) US4375595A (en)
EP (1) EP0070315B1 (en)
JP (1) JPS58500045A (en)
CA (1) CA1178338A (en)
DE (1) DE3273265D1 (en)
IT (1) IT1150382B (en)
SG (1) SG75988G (en)
WO (1) WO1982002806A1 (en)

Cited By (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2532115A1 (en) * 1982-08-19 1984-02-24 Western Electric Co CIRCUIT COMPRISING A VARIABLE TRANSCONDUCTANCE ELEMENT
EP0508360A1 (en) * 1991-04-12 1992-10-14 STMicroelectronics S.r.l. Sampled band-gap voltage reference circuit
WO1996003682A1 (en) * 1994-06-24 1996-02-08 Semcotec Handelsgesellschaft Mbh Temperature stabilising process
GB2308684A (en) * 1995-12-22 1997-07-02 Motorola Inc Switched capacitor voltage reference circuit
US7236047B2 (en) 2005-08-19 2007-06-26 Fujitsu Limited Band gap circuit

Families Citing this family (37)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4523107A (en) * 1982-04-23 1985-06-11 Motorola, Inc. Switched capacitor comparator
US4513207A (en) * 1983-12-27 1985-04-23 General Electric Company Alternating comparator circuitry for improved discrete sampling resistance control
US4588941A (en) * 1985-02-11 1986-05-13 At&T Bell Laboratories Cascode CMOS bandgap reference
US4736153A (en) * 1987-08-06 1988-04-05 National Semiconductor Corporation Voltage sustainer for above VCC level signals
US4896094A (en) * 1989-06-30 1990-01-23 Motorola, Inc. Bandgap reference circuit with improved output reference voltage
US5132556A (en) * 1989-11-17 1992-07-21 Samsung Semiconductor, Inc. Bandgap voltage reference using bipolar parasitic transistors and mosfet's in the current source
US5059820A (en) * 1990-09-19 1991-10-22 Motorola, Inc. Switched capacitor bandgap reference circuit having a time multiplexed bipolar transistor
US5280235A (en) * 1991-09-12 1994-01-18 Texas Instruments Incorporated Fixed voltage virtual ground generator for single supply analog systems
US5588673A (en) * 1994-02-01 1996-12-31 The Bergquist Company Membrane switch for use over a steering wheel airbag assembly
US5563504A (en) * 1994-05-09 1996-10-08 Analog Devices, Inc. Switching bandgap voltage reference
US5614816A (en) * 1995-11-20 1997-03-25 Motorola Inc. Low voltage reference circuit and method of operation
JP3262013B2 (en) * 1997-02-24 2002-03-04 三菱電機株式会社 Capacitive sensor interface circuit
US5796244A (en) * 1997-07-11 1998-08-18 Vanguard International Semiconductor Corporation Bandgap reference circuit
US5834926A (en) * 1997-08-11 1998-11-10 Motorola, Inc. Bandgap reference circuit
US5910726A (en) * 1997-08-15 1999-06-08 Motorola, Inc. Reference circuit and method
US6215353B1 (en) * 1999-05-24 2001-04-10 Pairgain Technologies, Inc. Stable voltage reference circuit
US6323801B1 (en) 1999-07-07 2001-11-27 Analog Devices, Inc. Bandgap reference circuit for charge balance circuits
US6060874A (en) * 1999-07-22 2000-05-09 Burr-Brown Corporation Method of curvature compensation, offset compensation, and capacitance trimming of a switched capacitor band gap reference
GB2359203B (en) 2000-02-09 2004-09-01 Mitel Semiconductor Ab CMOS Low battery voltage detector
US6529058B2 (en) 2001-01-11 2003-03-04 Broadcom Corporation Apparatus and method for obtaining stable delays for clock signals
US6535054B1 (en) * 2001-12-20 2003-03-18 National Semiconductor Corporation Band-gap reference circuit with offset cancellation
US6819163B1 (en) 2003-03-27 2004-11-16 Ami Semiconductor, Inc. Switched capacitor voltage reference circuits using transconductance circuit to generate reference voltage
US7161341B1 (en) * 2004-05-25 2007-01-09 National Semiconductor Corporation System, circuit, and method for auto-zeroing a bandgap amplifier
US7786792B1 (en) 2007-10-10 2010-08-31 Marvell International Ltd. Circuits, architectures, apparatuses, systems, and methods for low noise reference voltage generators with offset compensation
US8766602B1 (en) 2010-08-30 2014-07-01 Enerdel, Inc. Self protecting pre-charge circuit
CN102176188A (en) * 2011-03-30 2011-09-07 上海北京大学微电子研究院 Band-gap reference voltage producing circuit
TWI490456B (en) * 2011-04-29 2015-07-01 Elan Microelectronics Corp Differential Capacitance Sensing Circuit and Method
US8717005B2 (en) * 2012-07-02 2014-05-06 Silicon Laboratories Inc. Inherently accurate adjustable switched capacitor voltage reference with wide voltage range
US9063556B2 (en) * 2013-02-11 2015-06-23 Omnivision Technologies, Inc. Bandgap reference circuit with offset voltage removal
CN104375551B (en) * 2014-11-25 2017-01-04 无锡中感微电子股份有限公司 Band gap voltage generative circuit
US9342084B1 (en) 2015-02-20 2016-05-17 Silicon Laboratories Inc. Bias circuit for generating bias outputs
US9958888B2 (en) 2015-06-16 2018-05-01 Silicon Laboratories Inc. Pre-charge technique for a voltage regulator
CN105468077B (en) * 2015-12-28 2017-05-31 中国科学院深圳先进技术研究院 A kind of low-power consumption band gap reference
US10254177B2 (en) * 2016-09-14 2019-04-09 Nxp B.V. Temperature-to-digital converter
US10852758B2 (en) 2019-01-03 2020-12-01 Infineon Technologies Austria Ag Reference voltage generator
US11429125B1 (en) 2021-03-18 2022-08-30 Texas Instruments Incorporated Mitigation of voltage shift induced by mechanical stress in bandgap voltage reference circuits
CN115016589B (en) * 2022-06-01 2023-11-10 南京英锐创电子科技有限公司 Band gap reference circuit

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4071813A (en) * 1974-09-23 1978-01-31 National Semiconductor Corporation Temperature sensor
US4249122A (en) * 1978-07-27 1981-02-03 National Semiconductor Corporation Temperature compensated bandgap IC voltage references
US4295089A (en) * 1980-06-12 1981-10-13 Gte Laboratories Incorporated Methods of and apparatus for generating reference voltages
US4307333A (en) * 1980-07-29 1981-12-22 Sperry Corporation Two way regulating circuit
US4317054A (en) * 1980-02-07 1982-02-23 Mostek Corporation Bandgap voltage reference employing sub-surface current using a standard CMOS process

Family Cites Families (7)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4165478A (en) * 1977-09-21 1979-08-21 General Electric Company Reference voltage source with temperature-stable MOSFET amplifier
US4176308A (en) * 1977-09-21 1979-11-27 National Semiconductor Corporation Voltage regulator and current regulator
US4191900A (en) * 1978-01-27 1980-03-04 National Semiconductor Corporation Precision plural input voltage amplifier and comparator
US4210872A (en) * 1978-09-08 1980-07-01 American Microsystems, Inc. High pass switched capacitor filter section
US4287439A (en) * 1979-04-30 1981-09-01 Motorola, Inc. MOS Bandgap reference
US4263519A (en) * 1979-06-28 1981-04-21 Rca Corporation Bandgap reference
US4325018A (en) * 1980-08-14 1982-04-13 Rca Corporation Temperature-correction network with multiple corrections as for extrapolated band-gap voltage reference circuits

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
US4071813A (en) * 1974-09-23 1978-01-31 National Semiconductor Corporation Temperature sensor
US4249122A (en) * 1978-07-27 1981-02-03 National Semiconductor Corporation Temperature compensated bandgap IC voltage references
US4317054A (en) * 1980-02-07 1982-02-23 Mostek Corporation Bandgap voltage reference employing sub-surface current using a standard CMOS process
US4295089A (en) * 1980-06-12 1981-10-13 Gte Laboratories Incorporated Methods of and apparatus for generating reference voltages
US4307333A (en) * 1980-07-29 1981-12-22 Sperry Corporation Two way regulating circuit

Cited By (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
FR2532115A1 (en) * 1982-08-19 1984-02-24 Western Electric Co CIRCUIT COMPRISING A VARIABLE TRANSCONDUCTANCE ELEMENT
EP0508360A1 (en) * 1991-04-12 1992-10-14 STMicroelectronics S.r.l. Sampled band-gap voltage reference circuit
US5352972A (en) * 1991-04-12 1994-10-04 Sgs-Thomson Microelectronics, S.R.L. Sampled band-gap voltage reference circuit
WO1996003682A1 (en) * 1994-06-24 1996-02-08 Semcotec Handelsgesellschaft Mbh Temperature stabilising process
AT403532B (en) * 1994-06-24 1998-03-25 Semcotec Handel METHOD FOR TEMPERATURE STABILIZATION
US5945871A (en) * 1994-06-24 1999-08-31 National Semiconductor Corporation Process for temperature stabilization
GB2308684A (en) * 1995-12-22 1997-07-02 Motorola Inc Switched capacitor voltage reference circuit
GB2308684B (en) * 1995-12-22 2000-03-29 Motorola Inc Switched-capacitor reference circuit
US7236047B2 (en) 2005-08-19 2007-06-26 Fujitsu Limited Band gap circuit

Also Published As

Publication number Publication date
EP0070315A4 (en) 1983-06-17
IT1150382B (en) 1986-12-10
DE3273265D1 (en) 1986-10-23
JPS58500045A (en) 1983-01-06
IT8247697A0 (en) 1982-02-01
EP0070315A1 (en) 1983-01-26
US4375595A (en) 1983-03-01
EP0070315B1 (en) 1986-09-17
CA1178338A (en) 1984-11-20
JPH0412486B2 (en) 1992-03-04
SG75988G (en) 1989-03-23

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