WO1992004669A1

WO1992004669A1 - Improved imstrumentation system

Info

Publication number: WO1992004669A1
Application number: PCT/US1991/006442
Authority: WO
Inventors: L. Jonathan Kramer; Richard G. Hall
Original assignee: Drexelbrook Controls Inc.
Priority date: 1990-09-06
Filing date: 1991-09-06
Publication date: 1992-03-19

Abstract

An instrument system includes a voltage regulator (10) using as a pass element a field effect device (36) connected as an unity voltage gain amplifier and a feedback control amplifier (38) coupled to the pass element which is powered solely from the regulated voltage; a bridge (14) having a first switch (108) for coupling various load impedances to the bridge and a second switch (94) cooperating with the first switch to couple the bridge output to the selected load impedance; a phase shifting network coupled to the bridge (14) for providing a predetermined phase shift which is substantially independent of frequency over a range of frequencies; and/or an instrument system response time controlling circuit having a low pass filter (19) utilizing a variable synthesized capacitance (186) to effect changes in response time. The instrument system provides improved simplicity and reliability of operation, and is particularly useful as a radio frequency admittance responsive system for monitoring the condition of materials, such as the level of materials in a vessel.

Description

IMPROVED INSTRUMENTATION SYSTEM

Field Of The Invention

This invention relates to instrument systems, such as industrial instrumentation systems for monitoring of process parameters or other physical conditions. More particularly, this invention relates to instrumentation systems in which a process parameter is monitored by means of a radio frequency measurement. Still more particularly, this invention relates to radio frequency instrumentation systems in which the input to the system is an admittance, generated by an admittance-responding transducer which provides an output admittance that is related to a parameter .of interest. This invention also relates to circuits which, while particularly useful in such industrial instrumentation, may be advantageously employed in a wide variety of electronic systems.

Radio frequency instrumentation systems are well known. They find wide applicability in industrial instrumentation, such as in systems for monitoring the condition of materials. In such systems, a condition of interest, such as the level of materials in a vessel, ~ay be converted into an easily measured electrical quantity, such as an admittance, by an admittance-responding probe or sensor. The instrumentation system measures the electrical input at radio frequency in an admittance- responsive circuit to generate an output which bears a predetermined relationship to the input. The output may be used for a variety of indicating or control purposes.

Such instrument systems are well known in the art, typical examples being shown in U.S. Patent Ncs.

3,706,980; 3,746,975; 3,860,882; 3,993,947; 4,146,834; 4,232,300; 4,235,106; 4,363,030; 4,568,874; 4,723,122; 4,799,174. The present invention comprises improvements to such systems, and the disclosures of the referenced patents are incorporated herein by this reference.

While the systems described in the above- referenced patents are suitable for a wide variety of applications, they nonetheless have certain drawbacks. One such drawback is in their regulated power supplies, in which the feedback amplifiers which are responsive to the difference between a reference input and a signal derived from the regulated output are not powered from the regulated output. This leads to stability problems, supply rejection problems, and/or undue complexity of the regulator circuit.

Another difficulty with the above-referenced systems may arise from the fact that their admittance input is monitored by means of a bridge. As used herein, a "bridge" comprises any circuit in which an electrical quantity is compared to a reference, wherein the output of the circuit is responsive to the input and the reference. In order to match the input range of the bridge circuit to the output range of the admittance-responding transducer at the process conditions of interest, it is frequently necessary to adjust the bridge parameters by operation of various controls. In known radio frequency bridge-type instrument systems, the inclusion of such bridge- parameter-altering controls can result in inaccuracy, great expense, and undue complexity.

Another area in which the systems of above- referenced patents may suffer from drawbacks is in sensitivity to component tolerances and component changes.

In radio frequency operated systems, the frequency of operation is controlled at least in part by electronic components comprising the instrument system. In commercial production and application, the frequency of operation may be variable between nominally-identical systems due to component tolerances, and the frequency of operation of an individual system may be variable in response to changes in the components due to time, temperature or other such environmental factors or even in response to changes in the condition being monitored. The circuits of the above-referenced patents are sensitive in varying degrees to such frequency changes, which in some applications may result in unacceptably high errors.

Finally, in many applications of the systems of the above-referenced patents, it is desirable to provide a long and/or variable time of response of the system output to changes in the input. In the prior art circuits, the provision of such a long or variable response time has led to use of components whose values adversely affect intrinsic safety, stability, cost and/or reliability.

Summary Of The Invention

It is therefore a general object of the invention to provide an instrument system which does not suffer from the aforementioned drawbacks of the prior art systems.

It is another general object of the invention to provide a r; io frequency instrument system of improved simplicity, reliability, ruggedness, and inexpensiveness.

It is a more specific object of the invention to provide an instrument system having an improved power supply, and an improved power supply suitable for use in a variety of applications. It is another specific object of the invention to provide a bridge-type instrument system having improved controls for altering bridge parameters.

It is another specific object of the invention to provide an improved radio frequency instrument system having a reduced spurious output in response to frequency changes.

It is another specific object of the invention to provide an instrument system having a long and/or variable response time which is stable, inexpensive and easily implemented.

In accordance with the foregoing objects, the system of the present invention comprises a power supply having a feedback controlled amplifier which is powered from the regulated power supply output; a radio frequency bridge circuit comprising cooperative switching of bridge parameter control elements and the bridge output; a phase reference generating circuit which is highly insensitive to frequency variations; and/or a response time controlling circuit employing a synthetic capacitor. The foregoing and other objects and features of the present invention will be more fully understood with reference to the drawings and the following description thereof.

Brief Description Of The Drawings

Figure 1 is a block diagram of an instrument system in accordance with the present invention.

Figure 2 is a schematic diagram of a voltage regulating power supply circuit in accordance with the present invention. Figure 3 is a schematic of a diagram of a bridge circuit including the bridge load impedance switching and output switching scheme of the present invention.

Figure 4 is a schematic diagram of a phase reference generating circuit in accordance with the present invention.

Figure 5 is a phasor diagram illustrating the operation of the circuit of Figure 4 in one mode.

Figure 6 is a phasor diagram of the phase referenced generating circuit of Figure 4 when operated in another mode.

Figure 7 is a schematic diagram and phasor diagram of a prior art phase reference generating network.

Figure 8 is a schematic diagram of a low pass filter circuit including a synthesized capacitance.

Detailed Description Of The Preferred Embodiment

Figure 1 shows a block diagram of an instrument system, in particular a radio frequency admittance responsive system, in accordance with the present invention. Operating power supplied to the system at a power input 22 adapted to be coupled to a source of electrical power. A power supply 10 coupled to the input 22 and comprising a voltage regulator provides a regulated voltage output 34 which is shown coupled to oscillator 12 but may be used to power, directly or indirectly, the remainder of the circuitry. A radio frequency oscillator 12 energizes an admittance responsive bridge 14 ^• which responds to an input admittance ϊ, such as may be generated by an admittance-responding probe which is responsive to the level of materials in a vessel. The bridge output is typically too small to be directly detected, and so an error amplifier 16 may be coupled to the bridge output to amplify the bridge output signal prior to detection. Detection is accomplished by a detector 18, preferably by a phase sensitive detector, coupled to the bridge output such as by error amplifier 16, in accordance with a phase reference signal typically derived from the oscillator or the bridge. As shown in Figure 1, the phase reference input to the detector is provided by a phase reference generator 17 coupled to the oscillator. The detected signal provided by detector 18 is coupled to an output element 20, which generates an output 24 having a signal bearing a predetermined relationship to the input admittance. The output signal may be continuously variable or limited to discrete values or states. Desirably the instrument output is a two wire signal, such as a standard 4-20mA current signal, so that the instrument output is conducted to remote locations over a pair of conductors coupled to input 22 which supply power to the instrument system. In such a system, the instrument output 24 would be coupled to the power input 22.

A system such as is shown in Figure 1 is typically provided with means for controlling the response time of the output caused by changes in the input. Such a response time controlling means is shown in Figure 1 as a separate block comprising a low pass filter 19, but it will be understood that such a function may be considered as part of the detector or the output element and that its function may be implemented in other circuit blocks. The system of Figure 1 comprises a monitoring circuit for monitoring a physical condition in accordance with an input received from an admittance responding transducer which produces an output having a predetermined relationship to the input.

Figure 2 shows a schematic diagram of a voltage regulating power supply circuit in accordance with the present invention and suitable for use in the power supply 10 of Figure 1. The power supply circuit receives an input voltage B+ on conductor 30 which is unregulated with respect to a common potential on conductor 32. The circuit supplies a regulated output voltage on conductor 34 with respect to the common potential on conductor 32, which regulated voltage may be used to power, directly or indirectly, the remaining blocks of the circuit shown in Figure 1. Desirably, the unregulated potential B+ is supplied by the signal wires of a two wire D.C. current loop. However, in various applications, the regulated power supply circuit of Figure 2 may also be powered from a battery, a rectified A.C. voltage such as from a power transformer, or otherwise. The input to the regulator circuit may include additional components such as rectifiers to rectify an A.C. input or to prevent reverse polarity damage to the circuit, voltage limiting devices such as zener diodes for transient and/or input overvoltage protection, and capacitors to remove A.C. from the input to the regulator circuit. Such additional devices are well known in the art and are not shown in Figure 2.

Current flow into the voltage regulated conductor 34 is supplied from the 3+ potential by pass element 36. The pass element 36 is a field effect device, preferably a depletion mode device and preferably an MOS field effect transistor. For a regulator circuit providing a positive output voltage with respect to common, an n-channel device is used. An example of a suitable pass element is the depletion mode field effect transistor sold under the designation BSS129. It is connected in Figure 2 with its drain connected to B+, its source connected to the regulated output on conductor 34, and its control input, the gate, coupled to the output of a feedback control amplifier 38, which may be a type TLC271IP operational amplifier. Transistor 36 is thus coupled as a source follower, having substantially unity voltage gain from its gate to its source. The positive input of amplifier 38 is connected to voltage reference 46, which is desirably a stable device such as a type LM285 band gap reference. Voltage reference 46 is energized by current flowing through resistor 44 from the regulated supply. The negative input of amplifier 38 is connected to a voltage divider comprising resistors 40 and 42 connected between the regulated potential and the common potential, to provide a feedback input which varies in response to variations in the regulated output. A decrease in the regulated voltage causes a decrease in the feedback voltage applied to the negative input of amplifier 38, driving the output of amplifier 38 positive. This in turn raises the gate voltage applied to pass transistor 36, which increases the current flow through transistor 36 to provide feedback controlled regulation of the potential on conductor 34.

The use of a depletion mode field effect pass element provides substantial benefits in terms of simplicity of circuit design and effectiveness of operation. The regulator circuit is self-starting without the necessity of using a bypass element such as a resistor in parallel with the pass transistor, since the pass transistor is biased into conduction by pullup resistor 39. Such bypass elements can impose substantial burdens on circuit complexity and/or performance. Another advantage is that the feedback amplifier 38 may be powered from the regulated voltage. Because the threshold voltage of a depletion mode pass transistor 36 is negative, i.e., the gate must be negative with respect to the source to cut off channel conduction, the gate control voltage for transistor 36 need never rise above the regulated voltage. This permits the amplifier 38 to be powered from tr.e regulated voltage while its output is directly coupled to and at the same D.C. potential as the gate of the pass transistor 36, and full control of the pass transistor 36 may be effected while the output of amplifier 38 is maintained between its supply potentials. This eliminates complexity and stability problems associated with conventional circuits for coupling the feedback amplifier output to the pass element control input. In addition to providing simplicity, powering the feedback amplifier 38 from the regulated output 34 also provides substantially improved power supply rejection, as well as substantially restricting the range of input voltage to which the amplifier itself is exposed. This permits the use of commonly available, high performance, inexpensive amplifiers for the feedback amplifier 38.

The regulator circuit of Figure 2 also provides stability of operation with minimal complexity and expense. Conventional integrated circuit operational amplifiers suitable for use as amplifier 38 are typically unity gain stable. However, the pass element in conventional regulators may comprise an amplifier which is difficult to make stable or requires unwarranted complexity and expense to do so. Because the pass element 36 in the circuit of Figure 2 is connected as a source follower, it provides a gain which is substantially but slightly less than unity. Accordingly, the pass element of the present invention is inherently stable and is not subject to parasitic oscillation. Accordingly, since both the pass element and the feedback control amplifier are easily made unity gain stable, the entire regulator circuit is made intrinsically stable with minimal complexity and expense.

Resistor 50 provides additional assurances of stability by attenuating, in conjunction with the gate capacitance of transistor 36, any high frequency input voltage to the pass transistor. Capacitor 48 may be provided either to unity gain stabilize an amplifier 38 which is not intrinsically unity gain stable, or to establish a high frequency cutoff for amplifier 38. One or more decoupling or filter capacitors such as capacitor 60 may be provided across the regulated output to provide a low output impedance for the regulated supply.

The regulator circuit of Figure 2 is particularly advantageous for use in two wire operated, radio frequency monitoring systems. In a two wire system, the regulator circuit desirably introduces as little voltage drop as possible so that loop voltage may be made available for operating the instrument and for IR drops in the signal wires. The use of a field effect device as the pass element permits a minimal drop across the pass element; for instance, the preferred BSS129 device has a drain to source voltage drop of only .1 volt at a channel current of 20 mA, which is the maximum signal loop current of a typical two-wire instrument system. The regulator of Figure 2 is also highly desirable for use in radio frequency instrument systems, in which operation of an oscillator or other radio frequency circuitry may require substantial radio frequency current flow from the regulated potential. In the regulator circuit of Figure 2, such radio frequency current may be substantially supplied by capacitors such as capacitor 60, while the regulator circuit has substantial high frequency gain driving such capacitive loads and is nonetheless unity gain stable. Accordingly, with the regulator circuit of Figure 2, radio frequency current flow in the input lines is minimized. Redundant voltage limiting components such as zener diodes 54, 56, 58 may be provided to limit the voltage on the regulated output in order to assist in rendering the instrument system intrinsically safe and to protect circuitry coupled to the regulated output against overvoltage.

The circuit of Figure 2 may be modified to provide some, but not all, of the advantages set forth above by using an enhancement mode device instead of a depletion mode device for pass element 36. In this situation, means must be provided for coupling the output of amplifier 38, which is less than the regulated voltage, to the gate of pass element 36, which is greater than the regulated voltage. Such coupling may be achieved by coupling a current source 62 such as a current regulator diode and a voltage source 64 such as a zener diode between the unregulated potential 30 and the output of amplifier 38, and coupling the gate of pass element 36 to such current and voltage sources. In addition to added complexity and expense, this modification also has the disadvantage of requiring a higher input voltage to support a given regulated voltage in order to provide for the enhancement voltage of the pass element and the voltage drop across source 62.

While the regulator circuit of Figure 2 is particularly advantageous for use in radio frequency systems and fcr use in two wire instrument systems, its advantages of simplicity, stability, and effectiveness will render it useful in a wide variety of applications requiring a regulated potential.

Figure 3 is a schematic of a bridge circuit incorporating an improved circuit for altering the bridge parameters in accordance with the present invention. The bridge shown in Figure 3 is a transformer-coupled bridge, in which a transformer winding 70 is energized by an oscillator 12. Winding 70 is tapped to provide a first winding portion 72 and a second winding portion 74. These winding portions function as radio frequency voltage sources coupled in series between conductors 79 and 80 and form a fixed side of the bridge circuit, with conductor 78 coupled to the tap of transformer 70 providing a generally fixed bridge reference output voltage. Typically in an admittance responsive system for monitoring the condition of materials, and as shown, a conductor 80 coupled to winding 74 is connected to ground. It will be understood that other means for providing sources comprising a fixed side of the bridge may be employed. A variable side of the bridge comprises an admittance divider network in parallel with the fixed side of the bridge and comprising the admittance to be measured, represented in Figure 3 by capacitor 84, and a reference admittance represented in Figure 3 by capacitor 82, coupled in series at a conductor 86 with admittance 84. The admittance 84 is typically provided by an admittance-responding sensor, such as is illustrated in the patents referenced herein above. The output of the bridge circuit is the signal which is coupled to downstream circuit elements for further processing, e.g. the signal presented to the inputs of error amplifier 16. The bridge output is a differential signal generated with respect to the reference potential present on conductor 78 so as to vary in response to changes in admittance 84. As is known in the art, such a bridge is desirably operated near balance, i.e. when the bridge output voltage is substantially zero. This balance condition exists when the ratio of the voltages generated by windings 72 and 74 equals the admittance ratio of admittances 82 and 84. Typically admittance 82 is made variable, so that its value may be selected to cause the bridge to be balanced at a predetermined admittance of input admittance 84. This enables the bridge to be "zeroed", for instance to balance the bridge and set the instrument's output to a zero value at a predetermined level cf materials in a vessel.

The bridge circuit is desirably provided with terminals 88, 90, and 92 to provide a shielded coupling via a three-conductor cable to an admittance-responding sensor at a remote location. Such an arrangement is illustrated in the previously referenced patents.

The bridge gain, i.e. the change in bridge output voltage resulting from a given change in admittance 84, is determined by a variety of factors including the value of admittance 84, the value of admittance 82, and the value of the load admittance on the bridge output. The bridge gain is quite nonlinear with large deviations from bridge balance. The bridge gain is also variable with variations in the measured admittance 84, the reference admittance 82, and the bridge load admittance. In order to linearize the bridge output and stabilize it against changes in the above-referenced admittances, a large load admittance is typically disposed across the bridge between the fixed output and the variable output. Such a load admittance provides bridge stability and linearity at the expense of bridge gain, and accordingly typically requires the use of an error amplifier to amplify the bridge output signal to usable levels.

In order for a bridge type instrument system to be useful with a wide variety of input admittance ranges, the overall gain of the system must be made variable or alterable. This is typically accomplished at least in part by providing an alterable bridge load admittance to vary the bridge gain. In the circuit of Figure 3, as in the prior art, alteration of bridge gain is effected by a switch 108 which couples selected load admittances 96, 98, 100, 102, and 104, which are typically and as shown capacitors, between the fixed side of the bridge and the variable side of the bridge. It should be noted that no capacitor is switched in circuit by switch 108 in the position shown; rather, capacitor 106 representing the smallest bridge load admittance is permanently connected in circuit at the input to error amplifier 16. Such a configuration may aid in stabilizing the operation of the error amplifier.

The bridge gain selection circuit of Figure 3 differs from the prior art in that the bridge output (i.e. the error amplifier input) is not directly coupled to bridge conductor 86. Such a direct connection has led to certain difficulties in the prior art, which are overcome by the circuit of Figure 3. In particular, the spurious component of bridge load admittance caused by the admittance selecting switch 108, whether it is mechanical or electronic, can lead to substantial errors. For instance, a radio frequency instrument system operating at 100 kHz may be required to accommodate a full scale input admittance 84 of 40,000 pF. For acceptable stability and linearity, such an input admittance range may require a bridge load admittance 104 of 4.7 uF (microfarads) which at the specified measuring frequency has a reactance of about .34 ohm. The series resistance of mechanical switches (contact resistance) or electronic switches (on resistance) may easily be of the same or greater magnitude. In prior art bridge load switching systems, such conditions introduce a substantial amplitude variation and phase shift into the bridge output signal which is unrelated to the measured admittance. Moreover, this spurious error may be highly variable both in magnitude and phase with time, temperature, or other conditions.

Such a spurious output is minimized by the circuit of Figure 3. The bridge of Figure 3 includes an additional switch 94 which is coupled to bridge load admittance selecting switch 108. Switch 94 couples the input of error amplifier 16 (i.e. the bridge output) to the selected bridge load capacitor 96-104 where such capacitor is connected to switch 108. Accordingly, only the signal which is present across the selected bridge load admittance, and not the spurious signal which may be present across the internal impedance of the switch 108, is coupled to the input of error amplifier 16. The internal impedance of the switch, whether a contact resistance of a mechanical switch or an on resistance of electronic switch, is effectively placed in series with admittances 82 and 84. While the internal resistance of switch 108 may be appreciable compared to the values of the bridge load admittances 96-104, it is typically negligible compared to the admittances of reference admittance 82 or input admittance 84. Accordingly, in the circuit of the present invention, both the magnitude and phase of the error introduced by the internal impedance of switch 108 is negligible.

Figure 4 shows a schematic diagram of the phase reference generating network of the instrument system of the present invention. In typical radio frequency admittani-. monitoring systems, such as those shown in the above-referenced patents, the bridge output signal is coupled to the signal input of a phase sensitive detector, the output of which is responsive to the magnitude and phase of the input signal. As is disclosed in the prior art, such as in U.S. Patent 3,746,975, such phase sensitive detection permits a radio frequency admittance responsive instrument to be responsive to certain conditions and/or to be unresponsive to certain other conditions. Two detection phases are particularly useful in monitoring the condition of materials by an admittance- responding probe. When the detection phase is 0°, the instrument responds only to the capacitive component of the input admittance. This permits the system to ignore spurious resistive signals, such as arise from the presence of moisture in a granular insulating material. A 45° detection phase permits an instrument system to be substantially unresponsive to the presence of a resistive coating on an insulated sensor. The phase to which the phase-sensitive detector is responsive is determined by a phase reference input generated by a phase reference generator and supplied to the phase sensitive detector. Typically, the phase reference generator in an admittance- responsive system receives an input signal having a known phase with respect to the bridge, being derived either from the bridge itself or from the oscillator which energizes it. The phase reference generator generates one or more output signals, typically square waves, at the measuring frequency and bearing a predetermined phase relationship to the input of the phase reference generator.

While generating a phase reference at 0° may in principle be easily accomplished by applying the bridge- related signal directly to a high gain amplifier, a phase shifting network of some sort must be used to generate a phase reference at 45°. The use of such a phase shifting network introduces the possibility of substantial errors. For instance, the phase shift may be provided by an impedance or voltage divider comprising a resistive component and a reactive component. However, the .phase shift provided by such a network is dependent on frequency. In most admittance responsive systems the frequency of operation is variable with environmental and operating conditions, changes in which can result in substantial errors. In practice, the situation is complicated even when providing a 0° phase reference. Typical error amplifier circuits, amplifiers used in phase reference generators, and phase sensitive detectors provide an appreciable inherent phase shift or delay time at the frequency of operation. Such intrinsic phase shift must be compensated by the phase reference generating network so that the output of the phase sensitive detector provides the desired relationship to the bridge output. This generally requires introduction of phase shift even when a 0° net phase relationship is desired.

The phase reference generating circuit of Figure 4 provides phase shifts which are substantially independent of frequency over the frequency ranges typically encountered in the application of the present invention to admittance monitoring of the condition of materials. For instance, the network shown in Figure 4 may provide less than 1° of phase change in response to frequency variation in the range of 92 kHz - 108 kHz.

The phase reference generating circuit of Figure 4 comprises two portions: a three-stage amplifier comprising inverters 120, 122, and 124, and a phase shifting network receiving a bridge-related input signal and providing a phase shifted output signal to the amplifier. D.C. input bias for the amplifier input is provided by low pass filtering of the amplifier output, effected by resistor 132 and capacitor 134. This feedback establishes a D.C. voltage across capacitor 134 which is a function of the duty cycle of the amplifier output, and provides amplifier feedback to maintain the duty cycle at the desired value of 50%. The phase input is provided to the phase shift network through coupling capacitor 136, which is coupled to a radio frequency source of known phase such as the oscillator or the bridge to provide an input signal ^V _TM« ^Tne phase shifting network provides phase shift resulting in 0° detection or 45° detection, as selected by switch or jumper selector 138. In each condition, the circuit operates similarly in accordance with the general principles of this aspect of the invention, but somewhat differently. Operation of the phase shift network is best understood with reference to the phasor diagrams of Figures 5 and 6, discussed below. Prior to review thereof, it may be helpful to consider the operation of a nominally 45° phase shift network comprising a CR network, as has been used in the prior art.

Such a network and a phasor diagram illustrating its operation is depicted in Figure 7. The input voltage phasor V_TN is considered to have a 0° angle. The input voltage V_™ generates voltages across the capacitor and resistor as represented by phasors V-, and V_R, respectively, which are inherently at right angles to each other. In the circuit shown, the output phasor is the phasor V_R. As the frequency of V_IN varies from zero to infinity, the head of the output voltage phasor V_ will follow the indicated semicircle from one end point at the origin to another end point at the head of phasor V_T|J. At all frequencies the resistor voltage phasor V_R will be at right angles to the capacitor voltage phasor V_. For a network having a phase shift of 45° at frequency f , the voltage across the resistor and capacitor may be represented by V_R, and , . A reduction in frequency to frequency f- will result in resistor and capacitor voltages represented by phasors V_R2 and V_C2. This results in a phase shift and corresponding phase error E. * The error results from the fact that the output voltage phasor VR is not tangent to the indicated semicircle at any nonzero frequency, and at any nonzero frequency the derivative of the output phase with respect to frequency is not zero. The phase shift network of the present invention substantially reduces such phase error induced by frequency shift.

Figure 5 shows a phasor diagram illustrating the operation of the phase shifting network of Figure 4 in accordance with the present invention, when the 45° mode of operation is selected. Potentiometer 152 and resistor 154 form a phase adjusting network enabling the phase to be trimmed to account for component tolerances and the like. Its operation will be discussed later, and the immediately followinq analysis does not take into account the presence of this network.

Whereas in the prior art phase shift network in

Figure 7 the resistive load on the phase shifting capacitor is coupled to A.C. common, in the circuit of Figure 4 the phase shifting capacitor 150 is returned to a voltage divider comprising resistors 140 and 142. These resistors form a Thevenin source having a source impedance equal to the parallel combination of resistors 140 and 142 and a source voltage determine*^' by the driving voltage V_TN and by the divider ratio established by resistors 140 and 142. It is the presence of such a second source which enables the phase reference generating network of the present invention to have a phase shift which is substantially independent of frequency over an appreciable range of frequencies. As illustrated in Figure 5, the Thevenin source provides a voltage phasor V_TH which is in phase with the input phasor V,_N. The voltage across the Thevenin equivalent source resistance is represented by phasor v^" _RE0 and the voltage across capacitor 15.0 is represented by phasor V . These phasors must lie at right angles to each other, and accordingly the head of phasor V_nRE„Q~ describes the indicated semicircle, moving from the head of V--,_H to the head of V_IN as frequency ranges from zero to infinity. The output voltage of the phase shifting network, present on conductor 148, is represented by the phasor V_Q extending from the origin to the head -of phasor _RE0. The angle of phasor V_ with respect to the input phasor V_TN is the amount of phase shift introduced by the phase shifting network. When the output phasor V_π is substantially tangent to the semicircle, change in the phase of V_Q caused by frequency change is minimized, and at tangency, the derivative of the output phase with respect to frequency is zero. The condition of substantial tangency may be obtained by appropriate selection of the nominal frequency of operation, the source voltage and source impedance of the second source, and the value of capacitor 150.

Potentiometer 152 and resistor 154 provide a means for adjusting the source voltage and source resistance of the Thevenin source. Accordingly, by adjusting potentiometer 152, the phase shift may be trimmed to the desired value.

Figure 6 shows a diagram illustrating the operation of the phase shifting network of Figure 4 when the 0° mode of operation is selected. In this mode of operation, capacitor 146 and resistor 144 are coupled into the phase shift network, and resistors 140 and 142 are nonfunctional. Resistor 154 and potentiometer 152, together with resistor 144, form a Thevenin equivalent source having a source voltage and source impedance determined by the values of resistors 154, 144, and potentiometer 152 and the setting of potentiometer 152. The voltage of this Thevenin source is represented by phasor ^V _{R tj}« Capacitor 150 and capacitor 146 likewise form a Thevenin source having a source voltage and source capacitance determined by their values. Phasor VC_rI_τ,H„ represents the voltage of the Thevenin source formed by capacitors 150 and 146. Phasors _RE0 and _CEQ represent the voltages across the equivalent source resistance and equivalent source capacitance of the Thevenin sources; these phasors must lie at right angles. Accordingly, the head of the output voltage phasor ^" ₀ will traverse the indicated semicircle from the head of _RTH to the head of ^V _CTH ^as frequency ranges from zero to infinity. For the reasons described above with respect to Figure 5, the change in phase of the output voltage phasor with respect to frequency may be minimized by making the output voltage phasor substantially tangential to the semicircle, which may be effected by appropriate choice of frequency and component values in the phase shifting network. At tangency, the derivative of output phase with respect to frequency is zero.

It will be noted that phasor V_Q is not at 0° with respect to the input phasor V-_N, although the mode of operation is described as 0°. This is because some phase shift must typically be provided in the phase shift network to accommodate leads or lags occurring in other blocks of the circuit, such as the in error amplifier, the phase reference generator amplifier, and the phase sensitive detector. For instance, in the 0° mode a nominal phase shift of 10° may be required; likewise, to achieve 45° detection, a phase shift of 55° may be required.

It may be desirable to purposely depart from a nominal condition of exact tangency in order to accommodate the particular causes of frequency variation from the nominal which are expected. Some changes, such as caused by temperature-induced change in the value of frequency determining oscillator components, typically provide a relatively symmetric change in frequency over the range of conditions of interest, e.g. f + X%. Other changes provide a frequency shift primarily or only in one direction. For example, when the oscillator includes a frequency-determining resonant circuit which comprises the bridge, increasing measured capacitance from zero provides only a decrease in operating frequency. The overall error caused by such a non-symmetric frequency range may be minimized by a departure from nominal tangency, in the present example by shifting the output voltage phasor in the direction of higher frequency.

The output of the phase shift network, on conductor 148, is typically a sine wave, whereas a square wave typically is desired to drive the detector. Therefore, the output of the phase shifting network is applied to an amplifier comprising amplifiers 120, 122, and 124. These amplifiers desirably comprise CMOS inverters, and a type CD4007 integrated circuit may be configured to provide such amplifiers. The output of the first amplifier 120 is a substantially square wave, which is supplied to further amplifiers 122, and 124 to improve the rise time of the square wave phase reference signal. Either the output of the second amplifier or the third amplifier, or both, may be used to operate a phase sensitive detector.

Figure 8 shows a schematic diagram of a circuit providing a variable response time including a long response time for an instrument system of present invention such as is shown in Figure 1. The circuit is interposed between the phase sensitive detector and the output circuit and may appropriately be considered a part of either of them or a separate circuit block. The response time controlling circuit is implemented as a low pass filter having an input for receiving signals relating to a condition being monitored and an output providing output signals which are low pass filtered with respect to the input signals. The output of the phase sensitive detector is supplied to the input 180 of a low pass filter comprising a series impedance formed by resistor 182 between the filter input on conductor 180 and the filter output on conductor 184. An impedance shunting the output is provided, embodied in Figure 8 as a reactance including capacitor 204. It will be understood that while a resistive series impedance and capacitive shunt impedance as shown will implement the low pass function desired in this context, other impedances may be employed which will provide the same or other desired transfer functions. Capacitor 204, which in the circuit shown is always in circuit, provides a minimum time constant for the circuit.

The time constant of the low pass filter typically dominates the overall response time of the instrument, and in the prior art may be made variable by varying values of resistor 182 or capacitor 204. In many applications of admittance responsive instruments it is considered desirable for the response time to be variable from times on the order of milliseconds, so as to be able to respond to rapidly changing conditions of interest, to tens of seconds, to avoid spurious responses to slowly changing conditions such as waves in a vessel which it may be desirable to ignore. In the prior art, varying the instrument response time by varying resistor 182 or capacitor 204 is subject to certain drawbacks. Varying resistor 182 provides a varying load on the phase sensitive detector and a varying input impedance to the output stage, which can have deleterious effects on instrument operation. Varying capacitor 204 to provide long time delays on the order of tens of seconds can require extremely large values of capacitor 204, and such capacitors are typically large, expensive, unstable and may require complex and expensive schemes to protect them for intrinsic safety purposes. The circuit of Figure 3 avoids such drawbacks by use of a synthesized capacitance or synthetic capacitor, i.e. a circuit which responds in the manner of a capacitor whose value is determined by circuit parameters. The synthetic capacitor circuit of Figure 8 comprises operational amplifiers 188 and 190, capacitor 186, and resistors 192-202. Operation of the synthetic capacitor is as follows.

Capacitor 186 is coupled at one terminal to the output of the low pass filter at conductor 184 and its other terminal to the output of an inverting amplifier whose input is coupled to conductor 184. The effective capacitance of capacitor 186 in the circuit is thus multiplied by (1+A), where A is the gain of the inverting amplifier, to form a synthesized capacitance.

A second amplifier such as amplifier 188, connected an a unity gain noninverting buffer, may be provided. The input of amplifier 188 is coupled to the output conductor 184. The output of amplifier 188 is connected to potentiometer 200, connected as a variable attenuator, providing at the wiper of potentiometer 200 a fraction of the output voltage on conductor 184. The voltage fraction determined by the position of the wiper of potentiometer 200 is applied to amplifier 190, connected as an inverting amplifier whose gain is determined by the ratio of resistor 196 to resistor 202. Accordingly, the overall gain of the circuit, measured from the output of the low pass filter to the output of amplifier 190, may be varied from zero to the resistance ratio of resistors 196 and 202 by varying potentiometer 200. The synthesized capacitance generated by capacitor 186 and the amplifier circuit may thus encompass a wide range and large values without the use of large discrete components for capacitor 204 or 186. For instance, resistor 182 may have a value of 147 Kohms, and capacitor 186 may have a value of 1.5 uF, yielding a time constant of .22 seconds. Relatively small, stable, and inexpensive capacitors having this value are readily available. Amplifier 190 may be configured with a gain of 40, thus multiplying the effective value of capacitor 186 by a factor of 41. With the 1.5 uF value of capacitor 186, the effective value of the capacitance synthesized when the signal is unattentuated by potentiometer 200 is 61.5 uF, which would provide a time constant of about 9 seconds if resistor 182 has a value of 147 Kohms. Discrete capacitors having such a large value are generally large, expensive, unstable, and/or not readily available.

Resistor 198 is provided to balance the input impedances of amplifier 190. Amplifiers 188 and 190 may be type T1078, and desirably are powered from a dual supply having positive and negative voltages with respect to common. Resistor 194, coupled to the negative input of amplifier 190 and to a biasing potential V such as the positive supply potential, establishes the output voltage of amplifier 190 when no input is provided from potentiometer 200. Desirably, resistor 194 biases the output so that it is near one supply potential when there is no input, so that maximum excursion of the output in response to input signal is available. Resistor 192 limits the output current which may be required from amplifier 190.

While in principal extremely large capacitance values may be synthesized with the circuit of Figure 8, in practice the multiplication factor is limited by the available gain of the amplifier, i.e. the available output voltage excursion of amplifier 190 divided by the range of input signal expected at conductor 180.

It will be understood that while the response time controlling circuit of Figure 8 is particularly useful in radio frequency admittance responsive systems, it will also be applicable in connection with other instrument systems and other circuits.

While an improved instrument system may desirably contain the improved regulator, bridge load switching scheme, phase reference generator, and low pass filter, it will be understood that certain applications may require all of these improvements, but that in other applications various combinations of them may still desirably be employed. It will also be understood that various of the oscillators, error amplifiers, detectors, and output circuits disclosed in the patents referred to herein may be employed to provide these functions in the block diagram of Figure 1 in combination with the improvements disclosed herein.

While a preferred embodiment of the present invention has been shown and described, variations will no doubt occur to those skilled in the art without departing from the spirit and scope of the invention described in the foregoing specification and the following claims.

Claims

What is claimed is:

1. An instrument system comprising:

a power input adapted to be coupled to a source of electrical power;

a voltage regulator coupled to said power input and generating a regulated output having a regulated voltage, said voltage regulator including a field effect pass element coupled between said power input and said regulated output, said pass element including a control input for controlling the current through said pass element, said pass element including an amplifier having substantially unity voltage gain between said control input and said regulated output, said regulator further including a feedback amplifier powered from said regulated voltage and having an output coupled to said control input; and,

a monitoring circuit, coupled to said regulated output voltage, having an input for receiving an input signal representing a physical condition, said monitoring circuit producing an output signal having a predetermined relationship to said input signals.

2. A system according to claim 1, wherein said pass element comprises a depletion mode field effect device.

3. A system according to claim 1, wherein said pass element comprises a MOSFET.

4. A system according to claim 1, wherein said voltage regulator further comprises a voltage reference and means for generating a feedback signal which is a function of the regulated voltage, said feedback amplifier having inputs coupled to said voltage reference and said feedback generating means and having an output responsive to said feedback amplifier inputs.

5. A system according to claim 4, wherein said feedback amplifier is powered solely from said regulated output.

6. A system according to claim 4, wherein said control input is operated at substantially the D.C. potential of said feedback amplifier output.

7. A system according to claim 4, wherein said voltage reference includes a bandgap reference.

8. A system according to claim 1, wherein said instrument system is a two-wire instrument system, said monitoring circuit output signal being coupled to said power input.

9. A system according to claim 1, wherein said monitoring circuit input signal is an admittance, and said instrument system is admittance responsive.

10. A system according to claim 1, wherein said monitoring circuit operates at radio frequency.

11. An instrument system comprising a bridge having an input which is responsive to a physical condition, 'an output which is responsive to said input, and means for coupling a variable load admittance to said bridge, wherein said bridge output comprises said load admittance but not said coupling means.

12. A system according to claim 11, wherein said bridge input is an admittance.

13. A system according to claim 11, wherein said bridge operates at radio frequency.

14. A system according to claim 11, wherein said variable load admittance comprises a plurality of admittance elements and said coupling means comprises a first switch coupled to said admittance elements for selectively coupling said admittance elements to said bridge.

15. A system according to claim 14, wherein said bridge comprises a second switch coupled to said admittance elements for selectively coupling said admittance elements to said bridge output.

16. A system according to claim 15, wherein at least one of said first and second switches comprises an electronic switch.

17. A system according to claim 15, wherein at least one of said first and second switches comprises a mechanical switch.

18. A system according to claim 11, wherein said load admittance is substantially capacitive.

19. A system comprising a first radio frequency source and a phase shifting network coupled to said first source for generating a phase shifted output signal having a predetermined phase relationship to said first source at said radio frequency, wherein the derivative of the phase of said output signal with respect to frequency is substantially zero at said radio frequency.

20. A system according to claim 19, wherein said phase shifting network comprises a second radio frequency source and a voltage dividing network including resistive and reactive components, said voltage dividing network being coupled to said first and second sources.

21. A system according to claim 20, wherein said second source comprises a Thevenin equivalent source including said first source and said voltage dividing network.

22. A system according to claim 21, wherein said phase shifting network comprises a third radio frequency source including a Thevenin equivalent source which includes said first source and a voltage dividing network.

23. A system according to claim 20, wherein said second source includes a first resistor and a second resistor coupled in series to said first source.

24. A system according to claim 23, wherein said phase shifting network includes said first and second resistors and a capacitor coupled in parallel with one cf said resistors.

25. A system according to claim 24, wherein said output signal of said phase shifting network is the signal at the junction of said first and second resistors and said capacitor.

26. A system according to claim 20, wherein said second source includes a first capacitor and a second capacitor coupled in series to said first source.

27. A system according to claim 26, wherein said phase shifting network includes said first and second capacitors and a resistor coupled in parallel with one of said capacitors.

28. A system according to claim 20, further comprising means for adjusting the phase of said output signal.

29. A system according to claim 28, wherein said phase adjusting means comprises a potentiometer coupled to said first source.

30. A system according to claim 19, further comprising an amplifier having an input coupled to the output of said phase shifting network.

31. A system according to claim 19, further comprising a radio frequency bridge, wherein said first radio frequency source is coupled to said bridge.

32. A system according to claim 31, further comprising a phase sensitive detector having a signal input coupled to said bridge, a phase reference input for controlling the phase at which detector input signals are detected, and an output which is responsive to the amplitude and phase of the input signal, wherein the output of said phase shifting network is coupled to said phase reference input.

33. A system according to claim 32, wherein the phase shift provided by said phase shift network causes said phase sensitive detector to be primarily responsive to input signals substantially in phase with said bridge.

34. A system according to claim 32, wherein the phase shift provided by said phase shift network causes said phase sensitive detector to be primarily responsive to input signals at substantially 45° with respect to said bridge.

35. An instrument system comprising a monitoring circuit having an input for receiving input signals representing a physical condition to be monitored and producing an output responsive to said input signal, and a low pass filter having an output and having an input coupled to said monitoring circuit output, wherein said low pass filter comprises a synthesized capacitance.

36. A system according to claim 35, wherein said low pass filter comprises an impedance coupled in series between the input and the output of said low pass filter, and said synthesized capacitance comprises a capacitor having a first terminal coupled to the output of said low pass filter.

37. A system according to claim 36, wherein said synthesized capacitance includes a first amplifier having an output coupled to said capacitor at a second terminal of said capacitor.

38. A system according to claim 37, wherein said first amplifier includes an input which is coupled to the output of said low pass filter.

39. A system according to claim 38, wherein said low pass filter includes a second amplifier having an input which is coupled to the output of said low pass filter and having an output which is coupled to the ^• input of said first amplifier.

40. A system according to claim 39, wherein said second amplifier comprises a unity gain buffer amplifier.

41. A system according to claim 38, further comprising means for adjusting the gain of said first amplifier, wherein said gain is the ratio of the output of said first amplifier to the output of said low pass filter.

42. A system according to claim 41, wherein said gain adjusting means comprises a potentiometer which provides a variable attenuation of the input to said first amplifier.

43. A system according to claim 37, further comprising a power supply generating a potential difference between a pair of supply conductors, wherein said first amplifier includes a pair of power supply inputs coupled to said supply conductors and means for biasing the output of said amplifier to a quiescent potential near the potential of one of said supply conductors.

44. A voltage regulator comprising a common terminal; an input terminal for receiving an unregulated input voltage with respect to said common terminal; an output terminal having an output voltage which is regulated with respect to said common terminal; a pass element coupled between said input terminal and said output terminal, said pass element including a field effect device having a control input for controlling the current through said pass element between said input terminal and said output terminal, said field effect device being connected as an amplifier having substantially unity voltage gain between said control input and said output terminal; and a control amplifier having a feedback input coupled to said output terminal and an output coupled to said pass element control input for feedback control of said regulator output voltage, wherein said control amplifier is powered solely by the regulator output.

45. A regulator according to claim 44, wherein said field effect device is a depletion mode field effect device.

46. A regulator according to claim 45, wherein said field effect device is an MOS device.

47. A regulator according to claim 44, further comprising a reference voltage source, wherein said control amplifier includes an input coupled to said reference voltage source, and said control amplifier output is responsive to the difference between said control amplifier inputs.

48. A low pass filter comprising an input terminal for receiving an input signal; an output terminal for providing an output signal which is low pass filtered with respect to said input signal; an impedance coupled in series between said input and output terminals; an amplifier having an amplifier input coupled to said output terminal and having an amplifier output; and a capacitor coupled between said amplifier output and said low pass filter output terminal.

49. A low pass filter in accordance with claim 48, further comprising means for varying the gain of said amplifier.

50. A low pass filter in accordance with claim

49, further comprising a buffer amplifier having a buffer amplifier input coupled to said low pass filter output terminal and a buffer amplifier output coupled to said amplifier input of said first-mentioned amplifier.