WO1996031947A2 - Offset-compensated linear rf detector - Google Patents
Offset-compensated linear rf detector Download PDFInfo
- Publication number
- WO1996031947A2 WO1996031947A2 PCT/FI1996/000179 FI9600179W WO9631947A2 WO 1996031947 A2 WO1996031947 A2 WO 1996031947A2 FI 9600179 W FI9600179 W FI 9600179W WO 9631947 A2 WO9631947 A2 WO 9631947A2
- Authority
- WO
- WIPO (PCT)
- Prior art keywords
- detector
- transistor
- output
- voltage
- circuit
- Prior art date
Links
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3211—Modifications of amplifiers to reduce non-linear distortion in differential amplifiers
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03F—AMPLIFIERS
- H03F3/00—Amplifiers with only discharge tubes or only semiconductor devices as amplifying elements
- H03F3/45—Differential amplifiers
- H03F3/45071—Differential amplifiers with semiconductor devices only
- H03F3/45479—Differential amplifiers with semiconductor devices only characterised by the way of common mode signal rejection
Definitions
- the Invention relates to a linear RF detector comprising a detector part having a detector diode biased with bias current, and a linearizer part having an operation amplifier to the inverting input of which is connected both the output of the detector part and the feedback path coming from the output of the operation amplifier.
- the level of a received radio signal is detected, but the detector is also used for detecting the level of a transmitted radio signal.
- This application relates to the latter kind of detector.
- Requirements set for an RF detector include good linearity, high speed and zero output voltage when the input voltage is zero. Accuracy of detection should not suffer if the detector part and the linearizer part are placed on different circuit boards.
- a known detector is shown in fig. 1.
- the detector part comprises a detector diode DI, a capacitor C and a resistor R 0
- the linearizer part comprises an operation amplifier 0A1 having a diode D2 and a resistor R in its feedback path.
- the signal whose level is to be detected is represented by generator V j Cos ⁇ , where V t is the amplitude to be detected.
- V k is a direct-current voltage used for biasing diode DI.
- the first term of the formula is the desired part depending on the RF input voltage.
- the second term is a constant offset voltage
- the third term is a varying offset voltage depending on the temperatures and device matching of the diodes.
- the part of output voltage V 0 that is dependent on voltage V i is given by a non-linear function f, where I 0 and I x are Bessel functions, and is obtained from formula (2)
- the prior art detector circuit illustrated by fig. 1 has several advantages.
- First, according to formula (2) the diode bias current I b has theoretically, i.e. if the diode functions as an ideal diode, no effect on the part f(V A , V ⁇ l ) of the output voltage V 0 that is dependent on the input voltage.
- the bias current will have to be set at a large enough value to enable fast charge/discharge of the circuit capacitances.
- resistor R 0 does not affect the output voltage in the transfer of the input voltage.
- the detector is fast, if R 0 and C are sufficiently small.
- the fourth advantage is that the linear dynamic range of the detector is about 50 dB.
- the lowest input level to a 50 ⁇ impedance is -20 dBm (sensitivity dV o /dVi has dropped to half of its nominal value) and the highest input level is +30 dBm, depending on the break down voltage of diode DI and supply voltage of the operation amplifier.
- the main disadvantage of the above-described known detector circuit concerns the output offset voltages, the values of which are predictable only if diodes DI and D2 are matched devices and have the same temperature, so they should preferably be on the same silicon chip. However, it is often advantageous to place the detector and the linearizing amplifier on different circuit boards. It should also be noted that if diodes DI and D2 are in the same package, RF energy will easily leak through diode D2 into the linearizing amplifier disturbing its operation. This is difficult to prevent, since D2 is in the feedback path of a fast amplifier, in the vicinity of which no filtering is allowed.
- fig. 2 A way of eliminating the offset voltage in a circuit according to fig. 1 is shown in fig. 2.
- offset compensation is used outside the feedback path of the linearizing amplifier.
- Diodes Dla and D2a are matched devices and have the same temperature, and they are preferably in the same package.
- Biasing voltage V k is obtained by the use of a diode D2 that is similar to diodes Dla and Dlb. This makes the bias current independent of the forward voltage and the temperature of the diodes.
- the offset voltage of matched diodes Dla and Dlb at the output of the first operation amplifier 0A1 is 2*V k .
- the circuit of fig. 2 has exactly the same drawbacks as the circuit of fig. 1: the detector and the linearizing amplifier must be placed on the same circuit board, which raises EMC (Electro-Magnetic Compatibility) problems when the RF input level is high (more than 10 dBm).
- EMC Electro-Magnetic Compatibility
- the RF energy leaks through diode Dlb into linearizer 0A1 and is rectified in the p-n junctions of diode Dlb and/or of the operation amplifiers. It is almost impossible to arrange an effective lowpass filter in the vicinity of Dlb, because the diode is in the feedback path of a fast amplifier.
- An extra operation amplifier reduces the speed of the detector and raises the price, especially since the detector must be fast.
- the object of the present invention is to provide a linear detector circuit that does not have the drawbacks involved in the offset voltages of the known detector.
- the object is thus to provide a detector whose output offset voltages have been eliminated and in which the detector and the linearizing amplifier may have different temperatures and which does not involve RF leakage into the linearizing amplifier.
- the object is achieved with the detector circuit disclosed in claim 1.
- the output offset voltage is rendered independent of the biasing voltage of the detector diode.
- the detector part and the linearizer part can thus be made physically separate.
- two transistors are connected such that the sum of their base-emitter voltages is essentially the same as the output offset voltage.
- one transistor circuit is preferably in a darlington configuration such that it generates an output voltage that compensates for the offset voltage generated by the transistor in the feedback path.
- fig. 1 shows a known linearizing detector
- fig. 2 illustrates a known compensation method
- fig. 3 illustrates a compensation principle according to the invention
- fig. 4 illustrates compensation of offset voltages
- fig. 5 illustrates compensation of offset voltages in a feedback path
- fig. 6 illustrates compensation of the offset voltage by an operation amplifier
- fig. 7 illustrates a modification of the compensation of the offset voltage.
- Fig. 3 illustrates the principle of the invention in a simplified form. In practice, the circuit will have to be supplemented with the values stated below.
- biasing voltage source V k has been transferred to the non-inverting input of the amplifier, so that the RF voltage source can be grounded directly.
- diode Dlb which is preferably identical to and located on the same silicon chip as detector diode Dla (diode DI in fig. 1), is used for transferring voltage V k such that bias current I b becomes independent of the device and temperature.
- a suitable value for the bias current is e.g. 50 ⁇ A.
- diode D2 of fig. 1 in the feedback path of the operation amplifier has been replaced with transistor Q 2 , fig. 3.
- the replacement is possible since the collector current vs. base-emitter current of the transistor obeys the same law as the current vs. voltage of the diode.
- the reference level of output voltage V 0 is earth ( ⁇ base voltage of the transistor) instead of direct current V k that affects at the inputs of the amplifier.
- the circuit of fig. 3 does not, however, function in exactly the way described above, for the collector voltage of the transistor is lower than the base voltage by voltage V k .
- the base voltage must thus be dropped.
- This is implemented by adding transistor Q 3a to the base circuit of transistor Q 2 as shown in fig. 4.
- the drop in the base voltage is the base-emitter voltage of the added transistor Q 3a , which is operated with collector current I 3a .
- the current is obtained by connecting the emitter of the added transistor Q 3a through resistor R6 to negative voltage source -V B .
- the offset voltage is eliminated by raising the output voltage by an equal amount. This is performed by using the base-emitter voltage of darlington transistors Q 2 _' Q 3b having collector currents I 2b and I 3b so that formula (4) holds true:
- resistor values of fig. 4 will not be discussed here. Determining of these values is obvious to a person skilled in the art. Resistors connected to supply voltage sources -V B and +V B can be replaced with current sources. Components Dla and Rl can be interchanged without affecting the operation of the circuit, so that a diode pair with a common anode can be used. In that case, however, most of the RF current will flow through capacitor C2, whereby it must be of a low loss type and grounded as directly as possible. Resistor R 7 , which connects the output of the operation amplifier to supply voltage -V B , is not essential to the operation of the circuit but can be used for reducing dissipation of the operation amplifier.
- the circuit of fig. 4 gives a negative output voltage V 0 .
- a positive output voltage is obtained by inverting the diodes, replacing the NPN transistors with PNP transistors, and changing the polarities of supply voltages -V B and +V B .
- the above circuit will function sufficiently well in most applications, although it has some minor defects. If necessary, most of the defects can be eliminated by adding to circuit complexity.
- the error caused by the different temperatures of the detector and the linearizer part in that part of the output voltage which is dependent on the input voltage is not eliminated. The error, however, is zero when the input voltage is zero, and at most it is in the order of dozens of millivolts when the input voltage is high.
- the diode bias current I b and currents I 3a and I 2b are derived from the negative voltage, whereas the compensating current I 3b is obtained from the positive voltage. This increases the sensitivity to non- symmetrical supply voltage variations.
- the ratio of the collector current of transistor Q 2b to diode bias current I b is not very well defined and is dependent on the temperature due to imperfect matching of diodes D la and D lb , input offset voltage of the operation amplifier, and the temperature and device dependencies of the current gain of transistor Q 3b . If the resulting output offset voltage is too high, it can be eliminated by tuning, e.g. by making resistor R4 adjustable.
- FIG. 6 shows another embodiment of the inven ⁇ tion where the output impedance is zero.
- Another operation amplifier 0A2 is used therein.
- This embodiment utilizes a usually undesired characteristic of the simplest possible differential operation amplifier: non- inverting amplification is twice as great as the inverting amplification. Only one transistor Q 4 is thus needed to simulate the base-emitter voltage of two transistors.
- Fig. 7 shows yet another embodiment, in which the defect of the circuit of fig. 4 is taken into account, i.e. the defect that the circuit does not function correctly if current amplification of transistor Q 3b is too small.
- the base current of transistor Q 3b thus forms too large a part of current I 2b .
- the problem can be solved by turning transistor Q 2b into a 'super emitter follower' by adding one PNP transistor Q4.
Abstract
Description
Claims
Priority Applications (7)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
EP96908150A EP0767989B1 (en) | 1995-04-05 | 1996-04-02 | Offset-compensated linear rf detector |
US08/750,324 US5987312A (en) | 1995-04-05 | 1996-04-02 | Offset-compensated linear RF detector having a detector part and a linearizer part |
DE69613531T DE69613531T2 (en) | 1995-04-05 | 1996-04-02 | LINEAR OFFSET COMPENSATED HIGH FREQUENCY DETECTOR |
AU51495/96A AU685359B2 (en) | 1995-04-05 | 1996-04-02 | Offset-compensated linear RF detector |
JP8530016A JPH11504774A (en) | 1995-04-05 | 1996-04-02 | Offset compensated linear RF detector |
AT96908150T ATE202662T1 (en) | 1995-04-05 | 1996-04-02 | LINEAR OFFSET COMPENSATED HIGH FREQUENCY DETECTOR |
NO965186A NO965186L (en) | 1995-04-05 | 1996-12-04 | Deviation-compensated linear RF detector |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
FI951623A FI97655C (en) | 1995-04-05 | 1995-04-05 | Linear RF detector with offset compensation |
FI951623 | 1995-04-05 |
Publications (2)
Publication Number | Publication Date |
---|---|
WO1996031947A2 true WO1996031947A2 (en) | 1996-10-10 |
WO1996031947A3 WO1996031947A3 (en) | 1997-01-09 |
Family
ID=8543188
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/FI1996/000179 WO1996031947A2 (en) | 1995-04-05 | 1996-04-02 | Offset-compensated linear rf detector |
Country Status (10)
Country | Link |
---|---|
US (1) | US5987312A (en) |
EP (1) | EP0767989B1 (en) |
JP (1) | JPH11504774A (en) |
CN (1) | CN1073760C (en) |
AT (1) | ATE202662T1 (en) |
AU (1) | AU685359B2 (en) |
DE (1) | DE69613531T2 (en) |
FI (1) | FI97655C (en) |
NO (1) | NO965186L (en) |
WO (1) | WO1996031947A2 (en) |
Families Citing this family (12)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5742845A (en) | 1995-06-22 | 1998-04-21 | Datascape, Inc. | System for extending present open network communication protocols to communicate with non-standard I/O devices directly coupled to an open network |
WO2002084862A1 (en) * | 2001-04-11 | 2002-10-24 | Koninklijke Philips Electronics N.V. | High duty cycle offset compensation for operational amplifiers |
KR100452918B1 (en) | 2002-04-12 | 2004-10-14 | 한국디엔에스 주식회사 | Spin-etcher with thickness measuring system |
US6980116B2 (en) * | 2002-12-20 | 2005-12-27 | Motorola, Inc. | Method for failure detection in a radio frequency device |
US6825715B2 (en) * | 2003-05-02 | 2004-11-30 | Biode, Inc. | Temperature compensated, high efficiency diode detector |
US6998918B2 (en) * | 2004-04-08 | 2006-02-14 | Sige Semiconductor (U.S.), Corp. | Automatic current reduction biasing technique for RF amplifier |
JP4549743B2 (en) * | 2004-06-07 | 2010-09-22 | 富士通セミコンダクター株式会社 | Temperature sensor circuit and calibration method thereof |
CN101688889B (en) * | 2007-05-14 | 2012-08-15 | 希泰特微波公司 | RF detector with crest factor measurement |
JP2009105726A (en) * | 2007-10-24 | 2009-05-14 | Panasonic Corp | High frequency power detection circuit and radio communications equipment |
JP6616810B2 (en) * | 2017-08-01 | 2019-12-04 | アンリツ株式会社 | Radio terminal reception characteristic measuring system and measuring method |
US10658980B2 (en) * | 2018-08-22 | 2020-05-19 | Honeywell International Inc. | Modulating input device having a full wave rectifier |
CN110971229B (en) * | 2019-12-26 | 2020-08-25 | 郑州科技学院 | Electronic signal calibration system |
Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4460873A (en) * | 1982-11-19 | 1984-07-17 | Control Data Corporation | Active differential output direct current offset voltage compensation circuit for a differential amplifier |
EP0367707A2 (en) * | 1988-10-31 | 1990-05-09 | International Business Machines Corporation | A circuit arrangement for adjusting offset voltages associates with operational amplifiers |
US5097223A (en) * | 1990-05-22 | 1992-03-17 | Analog Devices, Inc. | Current feedback audio power amplifier |
US5132637A (en) * | 1991-03-25 | 1992-07-21 | Harris Corporation | RF power amplifier system having improved distortion reduction |
EP0638995A1 (en) * | 1993-08-09 | 1995-02-15 | AT&T Corp. | A DC coupled amplifier FED by an RF detector |
Family Cites Families (7)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US3758865A (en) * | 1972-01-18 | 1973-09-11 | Gen Motors Corp | Bias voltage generator for the voltage-responsive tuning elements in an electronically tuned radio receiver |
US4068187A (en) * | 1976-03-19 | 1978-01-10 | Hitachi, Ltd. | Audio-frequency power amplifiers |
US4502015A (en) * | 1982-03-31 | 1985-02-26 | General Electric Company | Diode detector with linearity compensating circuit |
JPH0263206A (en) * | 1988-08-29 | 1990-03-02 | Toshiba Corp | Current mirror circuit |
IT1244210B (en) * | 1990-12-20 | 1994-07-08 | Sgs Thomson Microelectronics | UNIT-GAIN FINAL STAGE PARTICULARLY FOR MONOLITHICALLY INTEGRABLE POWER AMPLIFIERS |
DE4311411A1 (en) * | 1993-04-07 | 1994-10-13 | Philips Patentverwaltung | Amplifier arrangement |
JP3167608B2 (en) * | 1995-12-18 | 2001-05-21 | 日本マランツ株式会社 | Wireless device |
-
1995
- 1995-04-05 FI FI951623A patent/FI97655C/en not_active IP Right Cessation
-
1996
- 1996-04-02 EP EP96908150A patent/EP0767989B1/en not_active Expired - Lifetime
- 1996-04-02 AU AU51495/96A patent/AU685359B2/en not_active Ceased
- 1996-04-02 DE DE69613531T patent/DE69613531T2/en not_active Expired - Lifetime
- 1996-04-02 US US08/750,324 patent/US5987312A/en not_active Expired - Lifetime
- 1996-04-02 WO PCT/FI1996/000179 patent/WO1996031947A2/en active IP Right Grant
- 1996-04-02 AT AT96908150T patent/ATE202662T1/en not_active IP Right Cessation
- 1996-04-02 CN CN96190474A patent/CN1073760C/en not_active Expired - Fee Related
- 1996-04-02 JP JP8530016A patent/JPH11504774A/en active Pending
- 1996-12-04 NO NO965186A patent/NO965186L/en not_active Application Discontinuation
Patent Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US4460873A (en) * | 1982-11-19 | 1984-07-17 | Control Data Corporation | Active differential output direct current offset voltage compensation circuit for a differential amplifier |
EP0367707A2 (en) * | 1988-10-31 | 1990-05-09 | International Business Machines Corporation | A circuit arrangement for adjusting offset voltages associates with operational amplifiers |
US5097223A (en) * | 1990-05-22 | 1992-03-17 | Analog Devices, Inc. | Current feedback audio power amplifier |
US5132637A (en) * | 1991-03-25 | 1992-07-21 | Harris Corporation | RF power amplifier system having improved distortion reduction |
EP0638995A1 (en) * | 1993-08-09 | 1995-02-15 | AT&T Corp. | A DC coupled amplifier FED by an RF detector |
Also Published As
Publication number | Publication date |
---|---|
EP0767989A1 (en) | 1997-04-16 |
CN1154183A (en) | 1997-07-09 |
DE69613531T2 (en) | 2002-03-28 |
FI97655B (en) | 1996-10-15 |
NO965186D0 (en) | 1996-12-04 |
WO1996031947A3 (en) | 1997-01-09 |
AU685359B2 (en) | 1998-01-15 |
EP0767989B1 (en) | 2001-06-27 |
ATE202662T1 (en) | 2001-07-15 |
US5987312A (en) | 1999-11-16 |
JPH11504774A (en) | 1999-04-27 |
FI951623A0 (en) | 1995-04-05 |
AU5149596A (en) | 1996-10-23 |
FI97655C (en) | 1997-01-27 |
DE69613531D1 (en) | 2001-08-02 |
CN1073760C (en) | 2001-10-24 |
NO965186L (en) | 1997-02-04 |
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