WO1996042160A2

WO1996042160A2 - Transmission system using transmitter with phase modulator and frequency multiplier

Info

Publication number: WO1996042160A2
Application number: PCT/IB1996/000508
Authority: WO
Inventors: Leo Kool; Wilhelmus Hendrikus Gerardus Deguelle; Johannes Yzebrand Tichelaar
Original assignee: Philips Electronics N.V.; Philips Norden Ab
Priority date: 1995-06-08
Filing date: 1996-05-28
Publication date: 1996-12-27
Also published as: JPH11500882A; KR100395584B1; EP0775410A2; KR970705273A; CN1090861C; US5903609A; CN1159268A; WO1996042160A3

Abstract

In a transmission system for transmitting digital symbols according to a phase constellation, a transmitter (2) is provided with a phase modulator (8) and a frequency multiplier (18) to obtain a modulated carrier signal. To prevent degradation of the modulated carrier signal, the input constellation of the frequency multiplier (18) has at least one additional phase state with respect to the output constellation of the frequency multiplier. Advantageously, the number of phase states of the constellation at the input of the frequency multiplier (18) is N times the number of phase states of the constellation at the output of the frequency multipplier (18).

Description

Transmission system using transmitter with phase modulator and frequency multiplier

The invention is related to a transmission system comprising a transmitter for transmitting via a transmission medium to a receiver, a carrier modulated by a sequence of digital symbols according to an output constellation having a predetermined number of phase states, said transmitter comprising a cascade connection of a phase modulator and a frequency multiplier.

The invention is further related to a transmitter for transmitting digital symbols modulated on a carrier.

A system according to the preamble is known from the Japanese Patent Application laid open under No. 53-110355. Transmission systems for transmitting digital symbols modulated on a carrier according to a constellation with a predetermined number of phase states are used for a large number of applications. Examples of such applications are mobile radio, radio relay systems, and broadcast systems like Microwave Video Distribution Systems (MVDS). The used constellation can vary from BPSK (Binary Phase Shift Keying), QPSK (Quaternary Phase Shift Keying) to 16,64 and even 256 QAM (Quadrature Amplitude Modulation).

A usual way of generating such a modulated carrier is to modulating a carrier having a fairly low intermediate frequency according to the desired constellation. The desired RF signal with a frequency that can be in the GHz range (e.g. 27 GHz, 40 GHz) is obtained by mixing the modulated carrier with a local oscillator signal having a suitable frequency. If the desired frequency is substantially higher than the intermediate frequency complex filters are required to suppress the undesired sideband in die output signal of the mixer. If the desired frequency is larger than 20 GHz, building blocks like amplifiers and mixers are required that are able to handle these high frequencies. Such building blocks are difficult to manufacture and are consequently rather expensive. In the transmission system according to the previously mentioned Japanese

Patent Application a cascade connection of a phase modulator and a frequency multiplier is used. The phase modulator modulates a carrier according to a constellation with two phase states τ/2 radians apart. By using a frequency doubler, a signal being modulated according to a constellation with two phase states τ radians apart is obtained. This signal is a common BPSK signal. By using a frequency multiplier to obtain the desired RF signal, only the frequency multiplexer must be able to operate on the desired frequency. The remaining building blocks like amplifiers and mixer have only to operate on a fraction of the desired frequencies, and can consequently be less expensive.

Simulations of the transmission system according to the principle of me Japanese Patent Application shows a substantial degradation of the bit error rate in comparison with a system that does not use a frequency multiplexer for obtaining the final constellation.

The object of the present invention is to provide a transmission system according to the pre-amble in which no substantial degradation of the bit error rate occurs. Therefore the invention is characterized in the phase modulator is arranged for modulating the sequence of digital symbols on a carrier according to a further signal constellation having a number of phase states being larger than the predetermined number of phase states.

The invention is based on the recognition that the degradation of the bit error rate is caused by the asymmetry of the further signal constellation. By introducing at least one additional phase state to the further signal constellation the symmetry can be restored. This is now further explained using a simple example. The case of generating a QPSK signal using a phase modulator and a frequency multiplier is considered. It is assumed that the phase states in the output constellations are 0, ?r/2, x, 3χ/2. The corresponding phase states in the further constellation are 0, x/4, x/2, 3χ/4.

If in the output constellation a change from the phase state 3 /2 to the phase state is 0 is required, this can only be realised by a change from the corresponding phase state 3 /4 of the further constellation to the phase state 0 of the further constellation. Such a change results in a phase trajectory from 3x/4 via χ/2, x/4 to 0 in the further constellation. In the output constellation a corresponding phase trajectory from 3x/2 via x, χ/2 to 0 is obtained in stead of a direct change from 3 /2 to 0. Due to the longer phase trajectory additional intersymbol interference will occur leading to a decreased bit error rate. It can also be expected that the frequency spectrum will be broader than necessary.

If in this situation according to the inventive concept an additional phase state x is added to the further constellation, a change from 3x/2 to 0 in the output constellation can be obtained by a change from 3x/4 to x in me further constellation. Such a change of phase states in the further constellation loads to a direct change from 3x/2 to 0 in the output constellation. Consequently the performance of the transmission system will improve. An embodiment of the invention is characterized in that the number of phase states of the further constellation is at least to the product of the frequency multiplication factor of the frequency multiplier and the number of phase states of the output constellation. Using a further constellation with this number of phase states will result in an output constellation in which every conceivable change of the phase state can be obtained without passing other phase states in the output constellation.

A further embodiment of the invention is characterized in that the phase modulator comprises selection means for selecting a phase state of me further constellation corresponding to phase state in the output constellation in such a way that the phase change in the further constellation is minimal.

By selecting a phase state in the further constellation that minimized the phase shift in the further constellation results in a minimum intersymbol interference of the output signal of the transmitter. A further embodiment of the invention is characterised in that the transmitter comprises a first Nyquist filter for filtering the input signal of the frequency multiplier, the receiver comprises a second Nyquist filter for filtering the received signal, and in that the shape of the transfer function of the first Nyquist filter and the second Nyquist filter are different. In digital transmission systems in which digital symbols are modulated on a carrier the symbols to be transmitted are represented by square wave signals. If these square wave signals are directly modulated on a carrier the bandwidth of the modulated signal will be excessive. In order to limit the bandwidm of the modulated signal, the signal representing the digital symbols are filtered by a filter. It is also possible to filter the modulated signal. To avoid intersymbol interference by the filtering, the transfer function of the filter has to satisfy the so called Nyquist 1 criterion.

In general the Nyquist filtering is performed by a first Nyquist filter in the transmitter and a second Nyquist filter in the receiver. The cascade of the Nyquist filters now has to satisfy the Nyquist 1 criterion. In such a splitting of the filter operation the shape of the first Nyquist filter and the second Nyquist filter are equal. Simulations have shown that if a phase modulator connected in cascade with a frequency multiplier the use of equal shapes for the first and second Nyquist filter does not result in a minimum intersymbol interference. By using different shapes for both Nyquist filters a better performance can be obtained. In general is to be expected mat the bandwidth of the first Nyquist filer is lower than the bandwidth of the second Nyquist filter.

A further embodiment of the invention is characterised in mat the first Nyquist filter is a Raised Cosine filter with a roll off factor between 0.9 and 1, and in that the second nyquist filter is a Raised Cosine filter with a roll off factor between 0.3 and 0.4.

Simulations have shown that using such filters in me transmitter and in the receiver results in a negligible adverse effect of intersymbol interference on the bit error rate.

The invention will now be explained with reference to the drawings.

Herein shows:

Figure 1 a transmission system according to the invention; Figure 2 a first example of a further constellation to be used with a frequency doubler to obtain a QPSK signal; Figure 3 a second example of a furmer constellation to be used with a frequency doubler to obtain a QSPK signal;

Figure 4 a first example of a further constellation to be used with a frequency tripler to obtain a QPSK signal;

Figure 5 a second example of a further constellation to be used with a frequency tripler to obtain a QPSK signal;

Figure 6 an example of a further constellation to be used with a tripler to obtain a 16-QAM signal.

In the transmission system according to Figure 1 , an input of the transmitter, carrying the input symbols a_k, is connected to a first input of a phase modulator 8. An output of an oscillator 6, with an output frequency of 1,5 GHz is connected to a second input of the phase modulator 8. An output of the modulator 8 is connected to an input of a bandpass filter 10. The output of the bandpass filter 10 is connected to a first input of a mixer 12. An output of an oscillator 14, wim an output frequency of 12 GHz is connected to a second input of the mixer 12.

An output of the mixer 12 is connected to an input of a bandpass filter 14. The output of the bandpass filter 14 is connected to an input of an amplifier 16. The output of the amplifier 16 is connected to an input of a frequency multiplexer, being here a frequency tripler 18. The output of the frequency tripler 18 is connected to antenna 22 via a bandpass filter 20.

An antenna 24 is connected to an input of a receiver 26. At the output of the receiver 26 the reconstructed digital symbol a_k are available. The carrier signal generated by the oscillator 6 is modulated in phase according to a further constellation according to the invention by the phase modulator 8. The output signal of the phase modulator is filtered by the bandpass filter 10 in order to limit the bandwidth of the phase modulated signal.

The transfer function of the bandpass filter 10 is a so-called square root Nyquist filter. The transfer function of such a filter fulfils the following requirement:

In (1) is C a constant, T the symbol period, a the so-called roll off factor, f the frequency and f_a the frequency of the signal generated by the local oscillator 6. It is known that the use of this type of filter reduces the bandwidth of me output signal of the phase modulator 8 without introducing intersymbol interference at instants t=nT if the receiver uses a similar filter. The adverse effect of intersymbol is substantially zero for a filter having α= l in the transmitter and a filter having α=0.35 in the receiver. In general a value of between 0.9 and 1 for the filter in the transmitter, and a value of for the filter in the receiver is a good choice.

The output signal of me bandpass filter 10 is mixed with the output signal of the oscillator 14 by the mixer 12. The output signal of the mixer 12 now has a frequency of 13,5 GHz. The bandpass filter 14 removes the undesired sideband at 10,5 GHz and passes only the desired signal with a frequency of 13,5 GHz. This 13,5 GHz signal is amplified in the amplifier 16 and subsequently tripled in the tripler 18. The output signal of the tripler 18 is filtered by means of the band pass filter 20, to remove all output components but the desired output signal at 40,5 GHz. The 40,5 GHz signal is radiated by the antenna 22.

The antenna 24 receives the signal emitted by the transmitter 2. The output signal of the antenna 24 is demodulated and detected in the receiver 26. At the output of the receiver 26 the reconstructed symbols a_k are available. In the phase modulator according to Fig. 2, the symbols are applied to an input of a finite state machine 160. The output of the finite state machine S, carrying an output signal indicating the phase state actually to be transmitted is connected to address inputs of a read only memory 162 and to address inputs of a read only memory 164. The output of the read only memory 162 is connected to a first input of a digital multiplier 166. An output of a digital cosine generator 170 is connected to a second input of me digital multiplier 166. The output of the read only memory 164 is connected to a first input of a digital multiplier 168. An output of a digital sine generator 172 is connected to a second input of the digital multiplier 168. The output of the digital multiplier 166 is connected to an input of a digital to analog converter 176, and the output of the digital multiplexer 168 is connected to an input of a digital to analog converter 174.

The output of the digital to analog converter 176 is connected to a first input of a mixer 178. The output of the digital to analog converter 174 is connected to a first input of a mixer 180.

The signal from the oscillator 6 in Fig. 1 is applied to a second input of the mixer 178 and to an input of a x/2 phase shifter 184. An output of the x/2 phase shifter is connected to a second input of me mixer 180.

An output of the mixer 178 is connected to a first input of an adder 182 and to an output of the mixer 180 is connected to a second input of the adder 182. At the output of the adder 182 a modulated signal is available.

In the modulator according to Fig. 2 the finite state machine 160 provides in response to the symbols a_k an output signal representing the phase state in the further constellation according to me invention associated with the symbol a_k to be transmitted. The output signal of the finite state machine 160 is used to address the read only memories 162 and 163. The read only memory 162 provides the real part of the derived phase state in the further constellation, and the read only memory 164 provides the imaginary part of the desired phase state in the further constellation. The output signal of the read only memory 162 is multiplied by a cosine signal generated by the cosine generator 170. In order to simplify the multiplier the cosine signal can be represented by a sequence + 1, -, -1, + 1 ... The use of such a representation allows a substantial reduction of the complexity of the multiplier 166 because it has only to provide multiplications with +1, -1 and 0, which can be easily implemented.

The output signal of the read only memory 164 is multiplied by a sine signal generated by the sine generator 172. The sine generator 172 generated a sequence 0,

+ 1, 0, -1, 0, +1 being 90° out of the phase with respect to the output signal of the cosine generator 170. Consequently at the outputs of the multipliers 166 and 168 a quadrature signal is available, representing a carrier modulated according to the further constellation. By means of the digital to analog converters 176 and 174, the digital quadrature signal is converted into an analog quadrature signal. The quadrature signal represented by the output signals of the digital-analog converters 174 and a 176 is converted to a frequency of 1,5 GHz by the quadrature mixer constituted by the mixers 178 and 180, the adder 182 and the phase shifter 184. The constellation according to Fig. 3 is arranged to obtain a quadrature phase shift keying signal at the output of a frequency doubler. According to the inventive idea an additional phase state 28 is added to the phase states 30. 32, 34 and 36. The phase states 30, 32, 34 and 36 correspond to a phase 3x/8, x/8, -x/8 and -3τ/8 with respect to the positive real axis. After frequency doubling this results into phase states 3x/4, =x/4 and - 3x/4 respectively, being in accordance to a QPSK signal.

To prevent a degradation of the bit error rate at a transition from phase state 30 to phase state 36 the additional phase state 28 is introduced, to enable to obtain the same phase state in the output QPSK constellation without a large phase jump in the further constellation. In Fig. 4 a further constellation is which each conceivable phase state change is possible. For each phase state in the output constellation there are two corresponding states in the further constellation. E.g. for an input sequence 00 there are the phase states 32 and 40, leading bom to a phase state corresponding to x/4 in the output constellation. The phase state in the further constellation to choose depends on the previous phase state in the furmer constellation. The phase state in the further constellation chosen is that one which is nearest to the previous phase state in the further constellation are nearest to a previous phase state in the further constellation the choice is made randomly. This can be implemented by making the choice dependent on the output signal of a (binary) random generator. The above mentioned rules can be implemented in a finite state machine of which the new (phase) state in dependence on the previous (phase) state and the input signal and the random signal is presented in the table below. In said table the logical value "x" means a "don't care" OLD PHASE STATE INPUT RANDOM NEW PHASE STATE

00 0 32

00 1 40

01 X 42

28 10 X 30

11 X 28

00 X 32

01 0 34

01 1 42

30 10 X 30

11 X 28

00 X 32

01 X 34

10 X 30

32 11 0 36

11 1 28

00 X 32

01 X 34

10 0 38

34 10 1 30

11 X 36

00 0 40

00 X 32

01 X 34

36 10 X 38

11 X 36

00 X 40

01 0 42

01 1 34

38 10 X 38

11 X 36

00 X 40

01 X 42

10 X 38

40 11 0 28

11 1 36

00 X 40

01 X 42

10 0 30

42 10 1 38 The implementation of such a finite state machine is known to those skilled in the art. Because there are 8 output states, the output signal of the finite state machine can be represented by 3 bits.

In Fig. 5 a further constellation is shown, by which a QPSK signal can be obtained by using a frequency tripler. The phase states are here spaced x/12 apart. In general is this φ/n in which n is the frequency multiplication factor and in which ψ is the desired phase shift in the output constellation. In the further constellation according to Fig. 5 an additional phase state 43 is introduced in order to deal with a transition from 3 x/4 to 5x/4 in the output constellation, without needing to pass other phase states in the output constellation.

In Fig. 6 a further constellation to be used with a frequency tripler is shown which leads to an output constellation in which every conceivable phase change is possible. Because this further constellation has to be used with a frequency tripler, the number of phase states in the further constellation is 12 rather than 8 which was the number of phase states in a similar constellation to be used with a frequency doubler.

For each phase state in the output constellation where are now three corresponding states in the further constellation according to Fig. 5. E.g. for an input sequence of 00, there are the phase states 46, 54 and 62 in the further constellation, all leading to a phase state x/4 in the output constellation. The criterion for choosing a phase state in the further constellation is the same as the criteria used with the furmer constellation according to Fig. 4. Below the state transition table for initial state 42 is given

OLD PHASE STATE INPUT RANDOM NEW PHASE STATE

00 X 46

00 1 62

01 X 64

42 10 X 44

11 X 42

The phase state transition table can be easily expanded for all initial phase states by using the criterion explained above.

The output constellation according to Fig. 7 is a 16 QAM constellation. The binary numbers between brackets are the values of four subsequent bits corresponding to the phase-amplitude state.

Fig. 8 shows a further constellation to be used with a tripler, corresponding to the output constellation. In general for a phase amplitude state in the further constellation one can write:

r_f = a ⁿι (2 )

k- 2τr

Ψf (3)

^~n n

In (2) r_f is the amplitude corresponding to the respective phase-amplitude state in the further constellation, a is a proportionality constant, n is the frequency multiplication factor and r₀ the amplitude corresponding to the phase-amplitude state in the output constellation.

In (3) ( is the phase corresponding to the respective state of the further constellation, φ₀ the phase corresponding to the respective phase of the output constellation and k an integer between 1 and k. The further constellation according to Fig. 8 is derived from the output constellation according to Fig. 7 by applying (2) and (3) using n=2. the phase-amplitude states in the first and fourth quadrant are the phase-amplitude states with a value of k equal to 0 and the phase-amplitude states in the second and third quadrant correspond to k equal to 1. As can be seen form Fig. 8, for each phase-amplitude state in the output constellation two corresponding phase amplitude states exist. These phase-amplitude states are positioned symmetrically with respect to the origin. The choice between two possible phase-amplitude states in the further constellation can be made according to the criterion disclosed above. Below the state transition table for initial phase-amplitude state 106 is shown.

The state transition table can be easily expanded for the complete constellation according to the criterion mentioned above.

Claims

1. Transmission system comprising a transmitter for transmitting via a transmission medium to a receiver, a carrier modulated by a sequence of digital symbols according to an output constellation having a predetermined number of phase states, said transmitter comprising a cascade connection of a phase modulator and a frequency multiplier, characterised in that the phase modulator is arranged for modulating the sequence of digital symbols on a carrier according to a further signal constellation having a number of phase states being larger than the predetermined number of phase states.

2. Transmission system according to claim 1 , characterized in that the number of phase states of the further constellation is at least to the product of the frequency multiplication factor of the frequency multiplier and the number of phase states of the output constellation.

3. Transmission system according to claim 1, or 2, characterized in that the phase modulator comprises selection means for selecting a phase state of the further constellation corresponding to phase state in the output constellation in such a way that the phase change in the further constellation is minimal.

4. Transmission system according to claim 3, characterized in that the selection means are arranged for selecting the phase state of the further constellation randomly out of a number of phase states causing all a minimal phase change in the further constellation.

5. Transmission system according to one of the previous claims, characterised in that the transmitter comprises a first Nyquist filter for filtering the input signal of the frequency multiplier, the receiver comprises a second Nyquist filter for filtering the received signal, and in that the shape of the transfer function of the first Nyquist filter and the second Nyquist filter are different.

6. Transmission system according to claim 5, characterised in that the first

Nyquist filter is a Raised Cosine filter with a roll off factor between 0.9 and 1, and in that the second nyquist filter is a Raised cosine filter with a roll off factor between 0.3 and 0.4.

7. Transmitter for transmitting a carrier modulated by a sequence of digital symbols according to an output constellation having a predetermined number of phase states, said transmitter comprising a cascade connection of a phase modulator and a frequency multiplier, characterised in that the phase modulator is arranged for modulating the sequence of digital symbols on a carrier according to a further signal constellation having a number of phase states being larger than the predetermined number of phase states.

8. Transmitter according to claim 7, characterized in that the number of phase states of the further constellation is at least to the product of the frequency multiplication factor of the frequency multiplier and the number of phase states of the output constellation.

9. Transmitter according to claim 7 or 8, characterized in that the phase modulator comprises selection means for selecting a phase state of the further constellation corresponding to phase state in the output constellation in such a way that the phase change in the further constellation is minimal.

10. Transmitter according to claim 9, characterized in that the selection means are arranged for selecting the phase state of the further constellation randomly out of a number of phase states causing all a minimal phase change in the further constellation.