WO2001015334A1

WO2001015334A1 - Coupling stage for a data transmission system for low voltage networks

Info

Publication number: WO2001015334A1
Application number: PCT/EP2000/008019
Authority: WO
Inventors: Jochen Mast; Lothar Heinemann; Miermann Rainer; Michael Arzberger
Original assignee: Abb Patent Gmbh
Priority date: 1999-08-26
Filing date: 2000-08-17
Publication date: 2001-03-01
Also published as: BR0013620A; NZ515475A; HUP0201602A2; SK2542002A3; TR200200479T2; DE19940544A1; NO20020402D0; CZ2002597A3; NO20020402L; ZA200108987B; PL353183A1; IL146376A0; HUP0201602A3; CN1371555A; EP1206846A1

Abstract

The invention relates to a device for coupling a high-frequency transmission signal (uHF) into a low voltage network (L1, N). An output amplifier (202) is provided. A differential mode voltage (uDiff) which is produced at a summing point (209) and represents an input voltage is supplied to the output amplifier between the transmission signal (uHF) and the feedback signal (urück). A network (206) is switched downstream in relation to the output amplifier (202) for adapting to the network impedance (ZL). Means (208) for detecting a voltage (umess) that is proportional to the network output current (iM) are arranged at the output of the network (206). The detected voltage (umess) is supplied to a controller module (207). The output signal of said controller module (207) is the feedback signal (urück).

Description

Coupling stage for a data transmission system for low-voltage networks

description

The invention relates to a device for coupling high-frequency useful signals for bidirectional data transmission on a low-voltage network.

Due to the deregulation of the energy market and the associated energy value-added services, the interest in the direct transmission of data between the energy supply companies and the various end customers has increased significantly. The bidirectional data transfer required for this can advantageously take place via the low-voltage network itself. This type of data transmission is referred to as power line communication (PLC). The data transmission system can be represented schematically by a star-shaped arrangement. At the node of the star is an intelligent control unit, the so-called intelligent network controller (= INC). In the end points, the signals for the bidirectional data transfer are each generated or processed by a transmitter / receiver unit. This' transceiver unit is also referred to as a transceiver (= TR). In practice, the star point usually corresponds to the 50 Hz distribution transformer station, the end points are usually found near the end customer's house connection.

In order to be able to use the low-voltage network as a transmission medium, a modulator / demodulator (= modem) is used in the INC as well as in each TR, where the digital user data is prepared in a suitable manner for the message channel. The modem in turn essentially consists of a part for signal processing, where the modulation / demodulation takes place, and a network coupling stage, with which the analog output signal is impressed on the low-voltage network and also received.

According to "German Institute for Standardization: DIN EN 50065-1: Signal transmission on electrical low-voltage networks in the frequency range 3 kHz to 148.5 kHz, VDE-Verlag, Berlin", the entire frequency band approved for communication on low-voltage distribution networks for energy supply companies ranges from 9 to 95 kHz. As explained in "Arzberger, M .: Data communication on electrical distribution networks for extended energy services, dissertation University of Karlsruhe, 1997", is based on the special properties of the message channel, the so-called frequency hopping modulation method (= FH modulation) with, for example, 4 discrete frequencies within a frequency range of e.g. Currently used 40 to 80 kHz. The INC periodically checks the accessibility of the individual TR and processes data traffic if necessary (= polling procedure).

In principle, two different variants for coupling and decoupling signals to the low-voltage network are conceivable, namely serial and parallel coupling and decoupling. A bidirectional data transmission in the frequency range from 40 to 80 kHz can be realized more cost-effectively with the parallel coupling and decoupling, since the output stage for decoupling the signals is simply via a sufficiently voltage-proof capacitor (to keep the 230 V / 50 Hz mains voltage away from the modem) and possibly connected in parallel to the supply network via a transformer for electrical isolation. The coupling and decoupling takes place either between a phase and the neutral conductor or, e.g. in networks without a neutral conductor, between two phases. The coupling and decoupling between a phase and the neutral conductor is usually preferred for practical reasons because the 50 Hz mains voltage, which is disruptive from a transmission point of view, is only 230 V compared to 400 V when coupling and decoupling between two phases.

The coupling of the signals to the low-voltage network requires significantly more circuitry than is the case when the signals are decoupled. The coupling of the signals must therefore also be given greater importance, since the greater potential for further cost savings with improved properties is seen here. Therefore, only the coupling of the signals is considered and discussed below. The actually always present stage for the decoupling of the signals is only mentioned if this appears necessary in the sense of the description of the invention.

The lines from the distribution transformer station to the house connections of the end users are partly designed as underground cables, partly as overhead lines, with several transitions from underground cables to overhead lines and vice versa can occur on the way from the distribution transformer station to the end user. In addition, branch points occur since not every end user is connected to the distribution transformer station via a separate cable connection. Compared to overhead lines, underground cables have a larger capacitance and a smaller inductance coating, which results in a significantly lower wave impedance of the underground cable. For this reason, a voltage division takes place at a transition from overhead line to underground cable, which makes a large contribution to the overall very high attenuation values. These joints are also the reason why that the distribution network is not reciprocal as a message channel, but that the behavior depends on the direction of communication. The damping properties of the news channel change over the course of the day, depending on how strongly and softly the low-voltage network is loaded by connected consumers (in particular devices with input-side EMC filters, such as primary clocked power supplies from TV sets etc.). In order to keep the negative effects of these properties on the reliability of the communication system low, it is desirable that the maximum permitted signal amplitude is always available at the network feed point in the entire useful frequency range, regardless of the current load condition.

The mostly strongly inductive, but possibly also capacitive coupling or access impedance can vary within wide limits at the network entry point of the INC or the TR. The magnitude of the impedance is frequency-dependent and, depending on the type of cabling and the load on the network and for each discrete transmission frequency, is between less than one ohm up to a hundred ohms. This impedance forms the load for the output amplifier of the INC and the TR. The lower the impedance, the more apparent power is required to impress a certain signal amplitude on the existing mains voltage. This apparent power must be provided by the power supply of the output amplifier as active power and is largely implemented as power loss in the output amplifier. The cause of this problem is to be found in the mismatching of the access impedance and source impedance of the switching network. It would be desirable if the output power to be applied by the output amplifier were pure active power. Only then is it possible to achieve an optimal design of the amplifier and its power supply with the lowest possible power.

Due to the frequency dependencies of the input impedance on the one hand and the impedance of the actual coupling network on the other hand, the waveform of the FH modulation signal is distorted. This leads to transient and decay processes during the transition between the discrete frequencies and must be avoided as far as possible with regard to interference-free data transmission.

In today's commercially available systems, the output amplifier together with the power supply causes a large share of the costs for the entire PLC system, so that the previously high costs of the entire system can be significantly reduced by optimal design of the network coupling stage. In addition, commercially available systems do not meet the above-mentioned basic technical requirements. The invention is therefore based on the object of specifying a coupling stage with which the overall system meets the above-mentioned elementary technical requirements and as a result of which a significant improvement in efficiency can be achieved with low manufacturing costs.

This object is achieved by a coupling network with the features specified in claim 1. Advantageous refinements are specified in further claims.

A further description of the invention and its advantages is given below using exemplary embodiments which are illustrated in the drawing figures.

Show it:

Fig. 1 block diagram of a conventional coupling stage according to the prior art

Fig. 2 block diagram of the coupling stage according to the invention

Fig. 3 circuit diagram of a common implementation of the coupling stage customary in the prior art

Fig. 4 circuit diagram of a first possible implementation of the invention

Einkoppelstufe

Fig. 5 circuit diagram of a second possible implementation of the invention

Einkoppelstufe

6a, b, c comparative representation of the transient response of the transmission signal according to the prior art and according to the coupling-in stage according to the invention

According to the block diagram shown in FIG. 1, the configuration customary in the prior art for feeding a transmission signal U _HF basically consists of a module for signal processing 101, in which the modulation / demodulation takes place, a module which contains an output amplifier 102, a module 103 for the power supply of the output amplifier 102 and the circuit for the signal processing 101, a module for coupling the signals 104 to the low-voltage network and a module for the output Coupling of the received signal 105. The modules 104 and 105 usually also each contain a transformer for potential isolation and for adapting the signal amplitude. The module 101 supplies a voltage U _H F as an analog output variable which is amplified by the output amplifier 102. The decoupled received voltage u _{RX is} to be regarded as the input variable of the module 101. The module 101 also handles the digital data transfer to the rest of the circuit. The output variable of the overall circuit forms the voltage u _L , which is then applied quasi above the network impedance Z _L. With u ui, u ₂ input and output voltages of the power supply module 103 are designated. The alternating voltage of the low-voltage network is designated by u _N. The output voltage of amplifier 102 is _labeled u _amP .

In the block diagram of the network coupling stage according to the invention shown in FIG. 2, three further modules 206, 207 and 208 are added to the block diagram shown in FIG. 1, with which the basic technical requirements mentioned at the outset are met. The modules denoted by 201 to 205 can be constructed in the same way as is given in the prior art for the modules 101 to 105 shown in FIG. 1. Module 206 is arranged between the module for output amplifier 202 and the module for coupling signals 204. With this network, which preferably consists of passive components, an adaptation of the impedance with which the output amplifier 202 is loaded (= matching network) is achieved. The network

206 is expediently designed such that in the worst case to be expected (this is the one in which the power to be applied by 202 and thus also by 203 is otherwise the highest), an ohmic load occurs at the output of amplifier 202. In this way, the power requirement of the output amplifier 202 and thus also the output power of the power supply 203 to be provided can be further reduced. The controller module 207 includes a circuit through which a regulation of the output voltage u _L without knowledge of the Einkoppelimpedanz Z _L can be obtained. This regulation can take place either directly (by measuring the output voltage u _L with the coupling stage arranged at mains potential) or indirectly by measuring a current (for example the output current of the output amplifier i _M or the current of the series-connected network for adapting the impedance). 2 shows an example of how the indirect regulation of the output voltage u _L can take place. The input variable of the controller module

207 forms the current i _{M of} the network 206 mapped by the transducer 208 to a voltage u _{m ess} to adapt the impedance. The output voltage of module 206 is denoted by u _M. The output variable u - ^ - ^ of the controller module 207 is one of the

Current of the network to adjust the impedance and thus the output voltage u equivalent voltage. As is common in control engineering, this voltage is compared with the AC voltage UF supplied by module 201. The differential voltage upi _f formed by means of an addition point 209 is then at the input of the output amplifier. available, which adjusts itself depending on the impedance of the network Z _L. It should also be mentioned that the module 206 with the network for matching the impedance lies within the control loop shown.

The arrangement shown in FIG. 2 consequently provides an output voltage u _{L of} almost constant amplitude, regardless of the frequency and the most varied impedances of the network, with at the same time a low power requirement of the output amplifier 202 and thus a low power requirement of the power supply 203.

Fig. 3 shows a circuit diagram of a common implementation according to the prior art. The basic problem of the known circuit is that the output amplifier 102 outputs a signal voltage of constant amplitude independent of the coupling impedance Z _L. The dynamically low output impedance of the amplifier 102 is greatly increased by the impedance Z _Q formed by the coupling capacitors C ^ and C ^ as well as the transformer T (with frequency-dependent short-circuit resistance and leakage inductance), which has parasitic properties. The coupling capacitor C _k - | is required to keep the 230V / 50Hz mains voltage u _N away from the output amplifier. The choice of the capacitance value of C ^ and C ^ is a compromise. On the one hand, the value should be as low as possible in order to limit the 50 Hz current through the output amplifier. On the other hand, the impedance of the capacitors C _k . and C ^ at the lowest signal frequency to be transmitted must not be too large, since it forms a frequency-dependent voltage divider with the coupling impedance Z _L and thus reduces the coupled signal amplitude. In particular at the low signal frequencies, where comparatively high coupling powers are required due to the particularly low amount of the coupling impedance, the high-pass character of the arrangement is noticeably noticeable. The capacitor C _k2 ensures that a DC _{component of} the output voltage of the amplifier, possibly due to the offset of the amplifier, does not magnetize the transformer in one direction and thus saturates the transformer.

The increased source impedance Z _Q results in a frequency and iast dependent

Voltage division between Z _L and Z _Q , which is why the amplitude of the output voltage u fluctuates very strongly. In such an arrangement according to the prior art, the impedance Z _{Q of} the coupling stage must be low in comparison to the lowest impedance of the network be so that the described voltage division is not too important. Even if this succeeds, it is not possible with this circuit to reproducibly adjust the amplitude of the input transmit signal over a larger frequency range.

FIG. 4, on the other hand, shows a first possible implementation of the coupling-in stage according to the invention with indirect regulation of the output signal u _L. In this implementation variant, the primary current i _{M of} the transformer T is used as a measure of the output voltage. This principle is based on the fact that the voltage at the line impedance Z can be approximately calculated from the integral of the current i _M with known values of the coupling capacitors C ^ and C- ^ and the parameters of the transformer. Simplified, it is assumed here that the 50 Hz voltage u _N - as with the

State-of-the-art circuit - drops at the coupling capacitor and that the corresponding mesh closes on the network side via the main inductance of the transformer. That is, by knowing the current i _M on the primary side of the transformer, it is possible to draw an approximate conclusion about the voltage u _L across the coupling impedance. The actual voltage signal can be reproduced from the current i _M if the output signal u _m mess of a current measuring device 208, which can be designed, for example, as a resistor or current transformer, is initially amplified in block 207 (indicated by the operational amplifier OP1 with the gain factor k) and then subjected to high-pass filtering in order to eliminate the remaining portion of the 50 Hz current. For example, this is done in FIG. 4 by means of R _{HP 1} , RH P ₂ . C _{HP 1} , C _{H P2} , switched as 2nd order high pass OP2. Finally, the current signal is fed to an integrator (OP3, R. ^ C ^. The resulting image u _{ώck of} the output voltage u _L and the output voltage u _{M of} a matching network 206 are weighted by means of R _x and R ₂ and summed up by the target Output voltage u _HF subtracted to form the control difference u _D ^ This takes place in the input part of the power amplifier 202, which is advantageously designed as a differential amplifier.

The matching network 206 is generally shown in the form of a T equivalent circuit diagram. In the simplest conceivable case, it can consist of only a single series inductance, which is connected to the series connection of the capacitors C _k - | and C _k2 of module 204.1 forms a series resonance circuit whose resonance frequency in the transmission band is close to the lowest frequency to be transmitted. Such a series inductance can either be a discrete component or else due to the leakage inductance of a transformer for level fit. In contrast to the prior art shown in FIG. 3, the leakage inductance of such a transmitter is not a disadvantage in the concept proposed in FIG. 4, rather a previously disruptive parasitic property of a component is advantageously used. This is essentially achieved by incorporating the adaptation network 206 into the inner loop of the two-loop control loop (so-called cascade control), which is the arrangement proposed in FIG. 4.

5 shows a second possible implementation of the coupling-in stage according to the invention. The entire coupling stage is at network potential. If the transformer for potential isolation in the output circuit is dispensed with, the module for grid coupling 204.2 is considerably simplified and contains only one coupling capacitor C ^. This not only saves the time-consuming and expensive transformer; on top of that a more exact reproduction of the voltage u _{L can be} achieved by the regulator module 207. In this variant, the load current i _M is used directly to calculate the output voltage u _L. For this purpose, analogous to FIG. 4, the current measuring device 208, here a simple shunt resistor R _cs , is used to convert the current i _M into a voltage ^u m ess. This is again amplified by OP1, filtered by OP2 and integrated by OP3. The controlled variable u _{M of} the inner control loop is scaled by R _l7, the feedback variable of the outer circle u _ώck by R ₂ . This weighted sum is subtracted from the desired output voltage u _HF to form the control difference u _{D: iff} , which is again advantageously carried out in the power amplifier 202. The properties of the cascade control come into effect in this variant as well as in the arrangement shown in FIG. 4. If potential isolation from the rest of circuit 201 is necessary, this can be done, for example, at the output of module 201, where u _HF is present. The advantage over the prior art (FIG. 3) is that only small powers occur at this point, so that a small-sized, inexpensive transformer can be used. In addition, its parasitic properties play only a minor role at the proposed installation location. It would also be conceivable to use an optocoupler instead of the transmitter.

6a to 6c finally show an example of a simulation result of the transient process of the transmission signal u _L. The vibration packet shown in FIG. 6a with 3 different frequencies, but of identical amplitude, should be impressed as precisely as possible on the unknown network impedance u _L with an amplitude of 2 V. FIG. 6b shows the time profile of the output voltage u _L when a coupling stage according to the prior art (FIG. 3) is used: There are strong transient distortions, and the amplitudes of the individual transmission frequencies are very different. The highest and the lowest transmission frequency do not reach the desired level at the entry point, which limits the range and reliability of the data transmission. Due to resonance effects, as can also be clearly seen in FIG. 6 b, uncontrolled amplitude increases can also occur at certain frequencies. Such behavior is, however, unacceptable from the standpoint of the EN50065-1 standard, which defines maximum amplitudes for such power line communication systems: depending on the impedance properties of the entry point, the permitted transmission levels can sometimes be exceeded considerably. This can only be prevented if the transmission amplitude is chosen so low from the outset that it is impossible to exceed it. However, this results in drastic losses in the range and reliability of the PLC system.

The results shown in FIG. 6c look different when the proposed novel coupling stage is used: the desired constant and very reproducible amplitude of the transmitted signal of 2 V is achieved, disturbing transient processes are largely suppressed. With this arrangement, the transmission levels specified in EN50065-1 can actually be exhausted without having to accept losses in level at the entry point or exceeding certain frequencies. The adaptation network 206 shown in FIGS. 2, 4 and 5 simultaneously succeeds in minimizing the power to be supplied by the transmission amplifier 202.

It follows from the above that the objectives

a) to increase the reliability of the entire power line data transmission system while b) simultaneously reducing the power loss of the transmission amplifier and the necessary power supply unit power,

can be achieved by using the proposed coupling device for power network communication systems.

Claims

claims

1. Device for coupling a high-frequency transmission signal (U _HF ) into a low-voltage network (L1, N), wherein

a) an output amplifier (202) is arranged, to which a differential voltage (uoi _tf ) formed at an addition point (209) between the transmission signal (U _HF ) and a feedback signal (u _mck ) is fed as the input voltage,

b) a network (206) is connected downstream of the output amplifier (202) for adaptation to the network impedance (Z _L ),

c) means (208) for detecting a voltage (u _mess ) proportional to the network output current (iM) are arranged at the output of the network (206), and

d) the detected voltage (u _me ss) is fed to a control module (207), the output signal of which is the feedback signal (u _ruc ).

2. Device according to claim 1, characterized in that the network (206) is constructed with passive components.

3. Device according to one of the preceding claims, characterized in that the controller module (207) contains means for performing amplification, high-pass filtering and an integration function.

4. Device according to one of the preceding claims, characterized in that the network (206) is followed by a coupling module (204.1) which contains a transformer (T) with primary and secondary coupling capacitors (C _k ι, C _k2 ).

5. Device according to one of claims 1 to 3, characterized in that the network (206) is followed by a coupling module (204.2) which contains only one coupling capacitor (Ck.).