WO2002025846A2 - Calibration of a transmit branch and/or a receive branch of a quadrature transmitter and/or transceiver - Google Patents
Calibration of a transmit branch and/or a receive branch of a quadrature transmitter and/or transceiver Download PDFInfo
- Publication number
- WO2002025846A2 WO2002025846A2 PCT/EP2001/010312 EP0110312W WO0225846A2 WO 2002025846 A2 WO2002025846 A2 WO 2002025846A2 EP 0110312 W EP0110312 W EP 0110312W WO 0225846 A2 WO0225846 A2 WO 0225846A2
- Authority
- WO
- WIPO (PCT)
- Prior art keywords
- branch
- transmit
- low frequency
- frequency component
- compensation
- Prior art date
Links
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B17/00—Monitoring; Testing
- H04B17/20—Monitoring; Testing of receivers
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
- H04B1/38—Transceivers, i.e. devices in which transmitter and receiver form a structural unit and in which at least one part is used for functions of transmitting and receiving
- H04B1/40—Circuits
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B17/00—Monitoring; Testing
- H04B17/10—Monitoring; Testing of transmitters
- H04B17/101—Monitoring; Testing of transmitters for measurement of specific parameters of the transmitter or components thereof
- H04B17/104—Monitoring; Testing of transmitters for measurement of specific parameters of the transmitter or components thereof of other parameters, e.g. DC offset, delay or propagation times
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04B—TRANSMISSION
- H04B17/00—Monitoring; Testing
- H04B17/10—Monitoring; Testing of transmitters
- H04B17/11—Monitoring; Testing of transmitters for calibration
- H04B17/14—Monitoring; Testing of transmitters for calibration of the whole transmission and reception path, e.g. self-test loop-back
Definitions
- the present invention relates to a transmitter or transceiver, more particularly to calibration of a transmit branch of such a transmitter or transceiver.
- a receive branch may be calibrated as well.
- Such a transmitter or transceiver operates in the so-called 2.4 GHz ISM band, for instance, or can be any other suitable transmitter or transceiver.
- the US Patent No. 5,793,817 discloses DC offset reduction in a transmitter.
- the transmitter has up-converters and an rf power amplifier through which a transmitter output signal is provided to an antenna.
- the transmitter has a feedback loop.
- the feedback loop derives a portion of the output signal, and divides the derived signal into phase related feedback paths.
- Each of the feedback path has frequency-down converters.
- the DC offset is measured at inputs of the up-converters when the feedback around a linearization loop is reduced to zero without altering the dc offsets produced at the outputs of the frequency down-converters.
- Subtractors subtract the measured DC offsets from feedback loop error signals. Such a DC nulling removes the effects of carrier feedthrough of the down-conversion mixers, thereby improving the resulting carrier feedthrough of the transmitter.
- a method of calibrating a transmit branch of a transmitter comprising: measuring a low frequency component in a high frequency output signal of said transmit branch, said low frequency component being present when said transmit branch is uncalibrated; from said measured low frequency component, deriving a first compensation signal and injecting said first compensation signal into an in-phase branch of said transmit branch, and deriving a second compensation signal and injecting said second compensation signal into a quadrature branch of said transmit branch; sweeping said first and second compensation signals and adapting said first and second derived compensation signals on the basis of said low frequency component; from said sweeping, obtaining minimum values of said measured low frequency component; and successively setting said first and second compensation signals at said minimum values.
- said compensation signals are injected after transmit filters in respective in-phase and quadrature branches of the transmit branch so that optimal carrier leakage compensation is achieved.
- different relative signal strengths of the high frequency output signal, a local oscillator signal, and sideband signals in a still uncalibrated transmitter are optimally taken into account when calibrating the transmitter.
- Such relative signal strength may vary depending on the silicon process used to manufacture the transmitter.
- carrier leakage compensation is done at full output power, and, thereafter, gain and phase error compensation is done at an output power level substantially lower than an output level of expected maximum carrier leakage.
- carrier leakage compensation is done at an output power level substantially lower than an output level of expected maximum carrier leakage, and, thereafter, gain and phase error compensation is done at full power.
- a transceiver that, in addition to the transmit branch, comprises a receive branch
- the method also comprises calibrating of the receive branch, by sweeping a gain setting in said receive branch while measuring an error signal, by obtaining a still further minimum value of said error signal during said sweeping of said gain in said receive branch, and by setting said gain in said receive branch to a value corresponding to said still further minimum value.
- Fig. 1 is a block diagram of a transceiver according to the present invention.
- Fig. 2 is a first embodiment of a power detector for use in a transceiver according to the present invention.
- Fig. 3 is a second embodiment of a power detector for use in a transceiver according to the present invention.
- Fig. 4 is a third embodiment of a power detector for use in a transceiver according to the present invention.
- Fig. 5 illustrates local oscillator leakage calibration according to the invention.
- Fig. 6 illustrates local oscillator leakage
- Figs. 7A-7D show sweeping of transmitter parameters for transmitter calibration.
- Figs. 8A-8B further illustrate transmitter calibration.
- Figs. 9A-9C illustrate receiver calibration.
- FIG. 1 is a block diagram of a transceiver 1 according to the present invention.
- the transceiver 1 operates in the so-called 2.4 GHz ISM (Industrial, Scientific and Medical) band, and is a so-called zero-IF transceiver that receives and transmits at the same frequency so that only a single tuned oscillator is needed.
- the transceiver 1 comprises a receive branch 2 and a transmit branch 3.
- the receive branch 2 comprises a low noise amplifier (LNA) 4 that is coupled to an antenna 5 via a filter 6 and a transmit/receive switch 7.
- the LNA 4 is coupled to a pair of quadrature mixers 8 and 9 in respective in-phase and quadrature receive branches.
- the mixer 8 is AC-coupled to a low pass filter 11.
- the mixer 9 is coupled to a low pass filter 13.
- the in-phase receive branch has a adjustable amplifier 14 to adjust the gain of the receive branch 2, and an injector 15 to inject a DC- voltage in order to compensate for DC-offsets
- the quadrature receive branch has an injector 16 to inject a DC-voltage to compensate for DC-offsets.
- the injectors 15 and 16 may be adders.
- an adjustable amplifier may be included in the quadrature receive branch.
- the transmit branch 3 of the transceiver 1 comprises transmit filters 20 and 21.
- the in-phase and quadrature transmit branches comprise respective injectors 22 and 23 to inject DC-voltages for calibrating the transmit branch 2, and further an adjustable amplifier 24 in the quadrature transmit branch.
- an adjustable amplifier may be included in the in-phase receive branch.
- both the in-phase and quadrature transmit branches may comprise adjustable amplifiers.
- the injectors 22 and 23 are coupled to transmit power amplifiers 28 and 29.
- the transmit power amplifier 29 is coupled to the transmit/receive switch 7.
- the transceiver 1 further comprises means 40 for measuring a low frequency component in a high frequency output signal of the transmit branch 3.
- the means 40 comprises a power detector 41, a high pass filter 42, and an AC-voltage detector 43.
- the power detector 41 provides a constant DC-signal so that the AC-voltage detector 43 does not produce an output signal.
- the power detector 41 provides a varying DC-signal, i.e. a low frequency component, so that the AC-voltage detector 43 produces an output signal.
- the transceiver 1 further comprises a voltage controlled oscillator 50 that is controlled by a phase locked loop (PLL) 51.
- the VCO 50 provides a local oscillator signal to the respective mixers 8 and 25 in the in-phase receive and transmit branches, and, through an adjustable phase shifter 52, a phase shifted local oscillator signal to the respective mixers 9 and 26 in the quadrature receive and transmit branches.
- the adjustable phase shifter 52 is adjustable around a ninety degrees nominal phase shift.
- the transceiver 1 further comprises a controller 60.
- the controller 60 comprises a processor 61, a non- volatile memory 62 for storing non- volatile data such as program and calibration data, and RAM 63 for storing volatile data.
- the controller 60 further comprises analog-to-digital converters (ADC) 64, 65, and 66 for sampling received and down-converted signals provided by the injectors 15 and 16, and for sampling an output signal provided by the AC-voltage detector 43, respectively.
- the controller 60 further comprises digital-to-analog converters (DAC) 67 and 68 for providing respective in-phase and quadrature transmit signals to the transmit filters 20 and 21, and digital-to-analog converters 69, 70, 71, 72, 73, 74, and 75 for respectively providing adjustment or calibration signals to the injectors 15 and 16, the amplifier 14, the injectors 22 and 23, the amplifier 24, and the phase shifter 52.
- ADC analog-to-digital converters
- DAC digital-to-analog converters
- Fig. 2 is a first embodiment of the power detector 41.
- the power detector 41 comprises an AM-demodulator formed by a diode 90, a resistor 91 and a capacitor 92.
- the AM-demodulator provides an output voltage that is proportional to the square root of the RF output power provided by the power amplifier 28.
- Fig. 3 is a second embodiment of the power detector 41.
- the power detector 41 comprises a micro-strip coupler 93 and a Schottky diode 94.
- the power detector provides an output voltage that is proportional to the RF output power provided by the power amplifier 28.
- Fig. 4 is a third embodiment of the power detector 41.
- the power detector 41 comprises a mixer 95, a resistor 96 and a capacitor 97.
- the power detector provides an output voltage that is proportional to the RF output power provided by the power amplifier 28.
- Fig. 5 illustrates local oscillator leakage calibration according to the invention.
- a four phase quadrature signal constellation is a perfect circle 100 as indicated by the dashed signal, i.e. the amplifier 28 provides constant RF power as indicated by constant radius r. Due to carrier leakage from the VCO 50 through the mixers 25 and 26, in an uncalibrated transmit branch, the signal constellation, for calibrated gain and phase, shifts as indicated by a solid circle 101 with a new center.
- the shifted circle 101 results in the transmit branch having different DC-components in the in-phase and quadrature transmit branches, V DC _ I and V DC _Q- Because radius r is then no longer constant, the RF power at the amplifier 28 significantly changes in proportionality with the square of radius r. Carrier leakage is compensated by injecting proper calibration voltages in the injectors 22 and 23. In case of still uncalibrated gain and phase in the transmit branch 3, the circle adopts the form of an ellipse so that also gain and phase calibration is needed.
- Fig. 6 illustrates local oscillator leakage in the frequency domain. Shown is a spectrum of f RF and carrier leakage signal f ⁇ >, for a single tone transmitted signal.
- Figs. 7A-7D show sweeping of transmitter parameters for transmitter calibration. In Figs. 7A and 7B, respective sweeping of signals Vi and V 2 to the injectors 22 and 23 is shown. The DACs 72 and 73 are respectively set to N ⁇ , op t and N 2> opt . Herewith carrier leakage is compensated for. In order to achieve better calibration, sweeping and setting of Vi and V 2 is repeated at least once. In Figs. 7C and 7D, respective sweeping of signals ⁇ G1 and ⁇ to the amplifier 24 and the phase shifter 52 is shown. The DACs 74 and 75 are respectively set to ⁇ G1, opt and ⁇ , opt. Herewith gain and phase imbalances in the transmit branch 3 are compensated for.
- Figs. 8 A-8B further illustrate transmitter calibration with still uncalibrated gain and phase imbalances.
- the sideband signal power P S B due to uncalibrated gain and phase imbalances is much weaker than the carrier leakage signal power P O , both signals being shown with respect to P RF .
- first low leakage is compensated for.
- full power is applied to the transmit branch 3, resulting in signal powers P' S B, P' LO and P'R F .
- Figs. 9A-9C illustrate receiver calibration, receiver calibration being done after transmitter calibration.
- the signal constellation for transmitted signals is optimal.
- the processor 61 closes the switch 80 and performs receiver calibration.
- V er ror average(V ⁇ _Q).
- the DAC 72 sweeps the gain signal ⁇ G2 and sets an optimum gain at ⁇ G2, opt. Gain sweeping can be done starting from a high gain, a low gain, or a nominal gain.
- DC-offsets can accordingly be set by the DACs 69 and 70, at V 3>opt and V 4j0p t- In one scenario, V 3 and V are set at some expected maximum DC-offset below full power reception, and, thereafter, at full power reception, the calibrated gain. For more accuracy, calibration may be repeated at least once.
Abstract
Description
Claims
Priority Applications (3)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
EP01972017A EP1364476A2 (en) | 2000-09-20 | 2001-09-06 | Calibration of a transmit branch and/or a receive branch of a quadrature transmitter and/or transceiver |
JP2002528939A JP2004509555A (en) | 2000-09-20 | 2001-09-06 | Calibration of the transmitting and / or receiving branches of a quadrature transmitter and / or receiver |
KR1020027006323A KR20020059745A (en) | 2000-09-20 | 2001-09-06 | Calibration of a transmit branch and/or a receive branch of a quadrature transmitter and/or transceiver |
Applications Claiming Priority (2)
Application Number | Priority Date | Filing Date | Title |
---|---|---|---|
US66592500A | 2000-09-20 | 2000-09-20 | |
US09/665,925 | 2000-09-20 |
Publications (2)
Publication Number | Publication Date |
---|---|
WO2002025846A2 true WO2002025846A2 (en) | 2002-03-28 |
WO2002025846A3 WO2002025846A3 (en) | 2003-10-02 |
Family
ID=24672120
Family Applications (1)
Application Number | Title | Priority Date | Filing Date |
---|---|---|---|
PCT/EP2001/010312 WO2002025846A2 (en) | 2000-09-20 | 2001-09-06 | Calibration of a transmit branch and/or a receive branch of a quadrature transmitter and/or transceiver |
Country Status (4)
Country | Link |
---|---|
EP (1) | EP1364476A2 (en) |
JP (1) | JP2004509555A (en) |
KR (1) | KR20020059745A (en) |
WO (1) | WO2002025846A2 (en) |
Cited By (2)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7305024B2 (en) * | 2003-08-29 | 2007-12-04 | Texas Instruments Incorporated | Method of fixing frequency complex up-conversion phase and gain impairments |
US8055205B2 (en) | 2005-04-22 | 2011-11-08 | Mstar Semiconductor, Inc. | Assessing the performance of radio devices |
Families Citing this family (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
JP4786335B2 (en) * | 2005-12-26 | 2011-10-05 | 京セラ株式会社 | Communication system, receiver and transmitter |
JP4918927B2 (en) | 2006-04-21 | 2012-04-18 | 日本電気株式会社 | Signal processing circuit |
US8385458B2 (en) | 2006-08-08 | 2013-02-26 | Nec Corporation | Signal processing circuit and signal processing method |
CN101743730B (en) | 2007-07-10 | 2014-02-12 | 日本电气株式会社 | Signal processor and signal processing method |
KR101681045B1 (en) * | 2010-11-22 | 2016-12-01 | 삼성전자주식회사 | Apparatus and method for calibration in wireless comunication system |
Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5574994A (en) * | 1994-07-15 | 1996-11-12 | Uniden Corporation | Method of correcting carrier leak in a transmitter |
US5584059A (en) * | 1993-06-30 | 1996-12-10 | Motorola, Inc. | DC offset reduction in a zero-if transmitter |
US5628059A (en) * | 1994-07-15 | 1997-05-06 | Uniden Corporation | DC offset circuit for cartesian loop |
EP0801465A1 (en) * | 1996-04-12 | 1997-10-15 | Continental Electronics Corporation | Radio transmitter apparatus |
US5793817A (en) * | 1995-10-24 | 1998-08-11 | U.S. Philips Corporation | DC offset reduction in a transmitter |
-
2001
- 2001-09-06 WO PCT/EP2001/010312 patent/WO2002025846A2/en not_active Application Discontinuation
- 2001-09-06 EP EP01972017A patent/EP1364476A2/en not_active Withdrawn
- 2001-09-06 KR KR1020027006323A patent/KR20020059745A/en not_active Application Discontinuation
- 2001-09-06 JP JP2002528939A patent/JP2004509555A/en active Pending
Patent Citations (5)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US5584059A (en) * | 1993-06-30 | 1996-12-10 | Motorola, Inc. | DC offset reduction in a zero-if transmitter |
US5574994A (en) * | 1994-07-15 | 1996-11-12 | Uniden Corporation | Method of correcting carrier leak in a transmitter |
US5628059A (en) * | 1994-07-15 | 1997-05-06 | Uniden Corporation | DC offset circuit for cartesian loop |
US5793817A (en) * | 1995-10-24 | 1998-08-11 | U.S. Philips Corporation | DC offset reduction in a transmitter |
EP0801465A1 (en) * | 1996-04-12 | 1997-10-15 | Continental Electronics Corporation | Radio transmitter apparatus |
Cited By (2)
Publication number | Priority date | Publication date | Assignee | Title |
---|---|---|---|---|
US7305024B2 (en) * | 2003-08-29 | 2007-12-04 | Texas Instruments Incorporated | Method of fixing frequency complex up-conversion phase and gain impairments |
US8055205B2 (en) | 2005-04-22 | 2011-11-08 | Mstar Semiconductor, Inc. | Assessing the performance of radio devices |
Also Published As
Publication number | Publication date |
---|---|
KR20020059745A (en) | 2002-07-13 |
JP2004509555A (en) | 2004-03-25 |
WO2002025846A3 (en) | 2003-10-02 |
EP1364476A2 (en) | 2003-11-26 |
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