WO2004105291A2

WO2004105291A2 - Ultra-wideband high data-rate communication apparatus and associated methods

Info

Publication number: WO2004105291A2
Application number: PCT/US2004/015060
Authority: WO
Inventors: Kazimierz Siwiak
Original assignee: Time Domain Corporation
Priority date: 2003-05-13
Filing date: 2004-05-13
Publication date: 2004-12-02
Also published as: WO2004105291A3

Abstract

An RF transmitter includes a reference signal generator, a signal generator, and a mixer. The reference signal generator provides a reference signal that has a prescribed or desired frequency. The signal generator provides an operating signal in response to a selection signal. The operating signal has a frequncy that equals the frequency of the reference signal multiplied by a number. The mixer mixes the operting signal with another signal to generate a transmission signal. An RF receiver includes a first mixer, a second mixer, an integrator/sampler, and a signal generator. The first mixer receives as its inputs an input RF signal and a second input signal, and mixes its input signals to genrate a mixed signal. The integrator/sampler receives the mixed signal and processes it to provde an output signal. The integrator/sampler receives the mixed signal and processes it to provide an output signal. The signal generator provides an operating signal in response to a selection signal. The operating signal has a frequency equal to the frequency of a reference signal, multiplied by a number. The second mixer mixes the operating signal with a template signal to generate the second input signal of the first mixer.

Description

ULTRA-WIDEBAND HIGH DATA-RATE COMMUNICATION APPARATUS

AND ASSOCIATED METHODS

Cross-Reference to Related Applications

This patent application is a continuation-in-part of, claims priority to, and

incorporates by reference U.S. Patent Application Serial No. 10/206,648, Attorney

Docket No. TDCO:015, titled "High Data-Rate Communication Apparatus and

Associated Methods," filed on July 26, 2002.

Furthermore, this patent application claims priority to, and incorporates by

reference, the following patent documents:

• Provisional U.S. Patent Application Serial No. 60/451,538, Attorney

Docket No. TDCO:016PZ1, filed on March 3, 2003, and titled "Ultra-

Wideband High Data-Rate Communication Apparatus and Associated

Methods"; • Provisional U.S. Patent Application Serial No. 60/401,711, Attorney

Docket No. Time.l66-P, filed on August 7, 2002, and titled "High Data-

Rate Communication Apparatus and Associated Methods"; and

• Provisional U.S. Patent Application Serial No. 60/402,677, Attorney

Docket No. Time.l70-P, filed on August 12, 2002, and titled "High Data-

Rate Communication Apparatus and Associated Methods."

Technical Field

This patent application relates generally to communication apparatus and, more

particularly, to ultra- wideband (UWB) high data-rate (HDR) communication apparatus.

Background

The proliferation of wireless communication devices in unlicensed spectrum and

the ever increasing consumer demands for higher data bandwidths has placed a severe

strain on those frequency spectrum bands. Examples of the unlicensed bands include the

915 MHz, the 2.4 GHz Industrial, Scientific and Medical (ISM) band, and the 5 GHz

Unlicensed National Information Infrastructure (UNII) bands. New devices and new

standards emerge continually, for example, the IEEE 802.11b, IEEE 802.11a, IEEE

802.15.3, HiperLAN/2 standards. The emergence and acceptance of the standards has

placed, and continues to place, a further burden on those frequency bands. Coexistence among the many systems competing for radio-frequency (RF) spectrum is taking on an

increasing level of importance as consumer devices proliferate.

Persons skilled in the art know that the available bandwidth of the license-free

bands (and the RF spectrum available generally) constrains the available data bandwidth

of wireless systems. Furthermore, data-rate throughput capability varies proportionally

with available bandwidth, but only logarithmically with the available signal-to-noise

ratio. Hence, to transmit increasingly higher data rates within a constrained bandwidth

requires the use of complex communication systems with sophisticated signal modulation

schemes.

The complex communication systems typically need significantly increased

signal-to-noise ratios, thus making the higher data rate systems more fragile and more

easily susceptible to interference from other users of the spectrum and from multipath

interference. The increased susceptibility to interference aggravates the coexistence

concerns noted above. Furthermore, regulatory limitations v thin given RF bands

constrain the maximum available signal-to-noise ratio and therefore place a limit on the

data-rate throughput of the communication system. A need therefore exists for a high

data-rate communication apparatus and system that can readily coexist with other existing

wireless communication systems, and yet can robustly support relatively high data-rates

in a multipath environment. Summary

One aspect of the invention relates to communication apparatus, such as

communication transmission apparatus and communication receiver apparatus. In one

exemplary embodiment, an RF transmitter according to the invention includes a reference

signal generator, a signal generator, and a mixer.

The reference signal generator provides a reference signal that has a prescribed or

desired frequency. The signal generator provides an operating signal in response to a

selection signal. The operating signal has a frequency that equals the frequency of the

reference signal multiplied by a number. More particularly, in some embodiments, the

number may constitute an integer number, whereas in other embodiments, the number

may constitute a non-integer number, as desired. The mixer mixes the operating signal

with another signal to generate a transmission signal.

In another exemplary embodiment, an RF receiver according to the invention

includes two mixers, a first mixer and a second mixer. The receiver further includes an

integrator/sampler and a signal generator.

The first mixer receives as its inputs an input RF signal and a second input signal.

The first mixer mixes its input signals to generate a mixed signal. The integrator/sampler receives the mixed signal and processes it to provide an output signal. The signal

generator provides an operating signal in response to a selection signal. The operating

signal has a frequency equal to the frequency of a reference signal, multiplied by a

number. More particularly, in some embodiments, the number may constitute an integer

number, whereas in other embodiments, the number may constitute a non-integer number,

as desired. The second mixer mixes the operating signal with a template signal to

generate the second input signal of the first mixer.

Brief Description of the Drawings The appended drawings illustrate only exemplary embodiments of the invention

and therefore should not be considered as limiting its scope. The disclosed inventive

concepts lend themselves to other equally effective embodiments. In the drawings, the

same numeral designators used in more than one drawing denote the same, similar, or

equivalent functionality, components, or blocks.

FIG. 1 shows several power spectral density (PSD) profiles in various

embodiments according to the invention.

FIG. 2 illustrates exemplary signal waveforms corresponding to a high data-rate

UWB apparatus. FIG. 3 depicts an exemplary embodiment of a high data-rate UWB transmitter

according to the invention.

FIG. 4 shows exemplary waveforms corresponding to a high data-rate UWB

transmitter according to the invention.

FIG. 5 illustrates an exemplary embodiment of high data-rate UWB receiver

according to the invention.

FIG. 6 depicts exemplary waveforms corresponding to a high data-rate UWB

receiver according to the invention.

FIG. 7 shows the timing relationship among various signals in a high data-rate

UWB transmitter according to the invention.

FIG. 8 illustrates exemplary desired or prescribed PSD profiles that correspond to

the two modes of operation in illustrative embodiments according to the invention.

FIG. 9 shows a PSD profile for an exemplary embodiment of the invention that

uses higher-order harmonics. FIG. 10 illustrates an illustrative PSD profile in an exemplary embodiment

according to the invention.

FIG. 11A shows one cycle of an exemplary output signal of a transmitter in a

UWB communication apparatus according to the invention.

FIG. 1 IB illustrates one cycle of another exemplary output signal of a transmitter

in a UWB communication apparatus according to the invention.

FIG. 12 depicts a timing relationship between several signals in an exemplary

embodiment according to the invention.

FIG. 13 shows several PSD profiles for an illustrative embodiment according to

the invention.

FIG. 14 illustrates several PSD profiles for other exemplary embodiments

according to the invention.

FIG. 15 depicts PSD profiles for other illustrative embodiments according to the

invention. FIG. 16 shows PSD profiles for other exemplary embodiments of communication

systems or apparatus according to the invention.

FIG. 17 illustrates an exemplary embodiment according to the invention of a

communication system that incorporates mode switching.

FIG. 18 depicts illustrative chipping sequences for use in communication systems

and apparatus according to the invention.

FIG. 19 shows an exemplary embodiment 19 of a differential receiver according

to the invention.

FIG. 20 illustrates a set of offset quadrature phase shift keyed (OQPSK) UWB

signals in an exemplary embodiment according to the invention.

FIG. 21 depicts a set of chipping signal waveforms in an exemplary embodiment

according to the invention.

FIG. 22 shows an exemplary embodiment of a transmitter according to the

invention that uses independently modulated harmonic signals. FIG. 23 illustrates an exemplary embodiment of a receiver according to the

invention for receiving independently modulated harmonic signals.

FIG. 24 depicts a sample waveform in an illustrative embodiment according to the

invention.

FIG. 25 shows a Fourier transform of the signal in FIG. 24.

FIG. 26 illustrates sample waveforms in an exemplary embodiment of a

transmitter according to the invention.

FIG. 27 depicts an exemplary in-phase channel pulse as a function of time in an

illustrative embodiment according to the invention.

FIG. 28 shows the magnitude of the spectrum of the signal in FIG. 27.

FIG. 29 illustrates an exemplary quadrature channel pulse as a function of time in

an illustrative embodiment according to the invention.

FIG. 30 depicts the magnitude of the spectrum of the signal in FIG. 29. FIG. 31 shows two signals as a function of time in illustrative embodiments

according to the invention.

FIG. 32 illustrates the spectra resulting from using the signal shaping shown in

FIG. 31.

FIG. 33 depicts two signals as a function of time in other illustrative embodiments

according to the invention.

FIG. 34 shows the spectra resulting from using the signal shaping shown in FIG.

33.

Detailed Description

This invention contemplates high data-rate communication apparatus and

associated methods. Communication apparatus according to the invention provide a

solution to the problems of coexisting communication systems, and yet providing

relatively high data-rates. Note that wireless or radio communication systems according

to the invention provide relatively high data-rates in "hostile" propagation environments,

such as multipath environments. Furthermore, as described below, one may apply the

inventive concepts described here to land-line communication systems, for example,

communication systems using coaxial cables, or the like. In one exemplary embodiment according to the invention, a high data-rate UWB

data transmission system uses a binary phase shift keying (BPSK) modulation of a carrier

frequency, known to persons of ordinary skill in the art with the benefit of the description

of the invention. One obtains the power spectral density (PSD) at frequency /of such a

system as:

where j denotes the reference clock frequency, and n represents the number of carrier

cycles per chip. In other words,

# of carrier cyles n = I chip

A chip refers to a signal element, such as depicted in FIG. 11A or FIG. 1 IB. Put another

way, a chip refers to a single element in a sequence of elements used to generate the

transmitted signal. The transmitted signal results from multiplying the sequence of chips

(the chip sequence) by a spreading code, i.e., the code that spreads the transmitted signal spread over a relatively wide band. Multiple chips in proportion to a desired energy level

per bit encode each data bit.

In this embodiment, the modulation chipping rate is commensurate with the

carrier frequency. Put another way, n is a relatively small number. In illustrative

embodiments, for example, n has a value of less than ten, such as 3 or 4. In other

illustrative embodiments, one may use n in the range of 1 to 500, or 1 to 42. Using the

latter range of values of n, one may achieve a UWB bandwidth of 500 MHz or greater, up

to a frequency limit of approximately 10.6 GHz, as prescribed in the Federal

Communications Commission's (FCC) Part 15 rules.

As persons of ordinary skill in the art with the benefit of the description of the

invention understand, one may use other positive integer values of n, as desired.

Generally speaking, the choice of the values of n depend on one's definition of ultra-

wideband. Depending on a desired bandwidth, one may select appropriate values of n, as

desired.

The value of n (rounded up to an integer value) corresponds to approximately the

desired center operating frequency divided by one half the desired bandwidth. In other

words,

or

where /, and Δ/ denote, respectively, the center operating frequency and the desired

bandwidth. For instance, the above example of the FCC's definition of UWB results in

values of n in the range of 1 to 42. More specifically, a 500-MHz-wide UWB system

operating below (by half the bandwidth) the current FCC Part 15 limit frequency of 10.6

GHz results in:

or „ = [41.4^"] = 42.

The FCC has also allowed UWB signals of at least a 500-MHz bandwidth in the

frequency range of 22-29 GHz, which corresponds to an upper value of n = 116,000.

Thus, persons skilled in the art with the benefit of the description of the invention may

choose virtually any appropriate ranges of values for n, depending on the performance

and design specifications and requirements for a given application. Note that generally

the signal bandwidth varies inversely with the value of n.

FIG. 1 illustrates several PSD profiles for various values of n (the number of

carrier cycles per chip). PSD profile 11 corresponds to n - \, whereas PSD profile 12

and PSD profile 13 correspond, respectively, to n = 2 and n = 3. Note that as the value of

n increases, the bandwidth of the modulated signal decreases. Note further that, in a

UWB system that one wishes to constrain to a predetermined maximum PSD (e.g., PSD

characteristics prescribed by a regulatory authority), one seeks to achieve as flat a

spectrum as possible in order to maximize the total transmitted power in a predetermined

bandwidth.

In such a system, one likewise seeks to choose a transmission bandwidth

independent of the modulation rate in order to maximize the total transmitted power. As

persons of ordinary skill in the art appreciate, in conventional BPSK systems, the PSD profile is not flat even in the highest bandwidth case, where n = \. Furthermore, the

bandwidth depends on the chip rate, as manifested by the parameter n. The dependence

of the bandwidth on the parameter n may be undesirable for a variety of reasons, such as

difficulty or failure to meet prescribed regulatory or design specifications.

For illustrative purposes, FIG. 2 depicts various signals corresponding to a BPSK

transmission system. Carrier signal 21 may include only a fundamental frequency.

Alternatively, rather than a continuous sine-wave signal, carrier signal 21 may include

other waveforms, as described below. FIG. 2 also shows a pseudo-random noise (PN)

sequence 22. Note that the waveforms in FIG. 2 correspond to a communication system

with one chip per RF cycle (i.e., n — 1), and 4 chips per data bit.

The third waveform in FIG. 2 corresponds to data bits 23. Beginning at time 27

and ending at time 28, PN sequence 22 codes data bits 23. The coding of data bits 23

results in signal 24. Signal 24 modulates carrier 21 to generate modulated signal 25.

Signal 26 acts a gating signal. Put another way, the commumcation system transmits

modulated signal 25 while the gating signal 26 is active (during the active portion of

signal 26). Modulated signal 25 has a spectrum substantially the same as spectrum 11 in

FIG. 1 (i.e., the case where the parameter n has a value of unity). One may determine the data-rate or data throughput of the communication system

from various system parameters. For example, assume that the carrier signal has a

frequency of 4 GHz, and that the system operates with one chip per RF cycle (i.e., n = \)

and 4 chips per data bit. Given those parameters, persons of ordinary skill in the art who

have the benefit of the description of the invention readily appreciate that the system

provides a 1-gigabit-per-second (Gb/s) data rate.

One exemplary embodiment of a high data-rate UWB system according to the

invention includes a high data-rate UWB transmitter and a high data-rate UWB receiver.

FIG. 3 shows an exemplary embodiment of high data-rate UWB transmitter 4 according

to the invention.

Transmitter 4 includes reference clock 41 (a reference clock generator), timing

controller 42, data buffer 43, PN generator 45 (a pseudo-random noise sequence

generator), data/PN combiner 46, mixer 47, antenna 48, and harmonic generator 49.

Reference clock 41 generates a signal with a desired frequency. The frequency of

reference clock 41 corresponds to a carrier frequency for transmitter 4. Thus, the

frequency of reference clock 41 corresponds to the desired carrier frequency. One may

implement reference clock 41 in a number of way and by using various techniques that

fall within the knowledge of persons skilled in the art with the benefit of the description

of the invention. Reference clock 41 couples to harmonic generator 49. Based a clock signal it

receives from reference clock 41, harmonic generator 49 generates one or more

harmonics of the carrier frequency (the frequency of clock reference 41). For example,

given a clock frequency/,, a second harmonic signal at the output of harmonic generator

49 has a frequency 2-f_c, and so on, as persons skilled in the art with the benefit of the

description of the invention understand. Harmonic generator 49 generates the one or

more of harmonics synchronously with respect to the reference clock (i.e., the one or

more harmonics are synchronized to the reference clock).

Note that one may realize harmonic generator 49 in a number of ways, for

example, comb line generators, as persons of ordinary skill with the benefit of the

description of the invention understand. As another example, one may use phase-locked

loops, as desired. As other examples, one may employ an oscillator followed by digital

divider circuitry. By dividing a signal of a given frequency by various integers, one may

obtain the one or more harmonics. In connection with such an implementation, one may

use fractional-N synthesizers, as desired.

Furthermore, one may use a variety of circuitry and techniques to synchronize the

one or more harmonics to the reference clock. Such circuitry and techniques fall vvithin

the knowledge of persons of ordinary skill in the art who have the benefit of the description of the invention. As an example, a comb line generator may provide

synchronization of the one or more harmonics to the reference clock.

Mixer 47 receives the one or more harmonics from harmonic generator 49. Mixer

47 mixes the one or more harmonics of the carrier frequency with a signal (described

further below) that it receives from data/PN combiner 46. Mixer 47 provides the

resulting signals to antenna 48. Antenna 48 propagates those signals into the

transmission medium. In illustrative embodiments, antenna 48 may constitute a wide¬

band antenna.

Examples of wide-band antennas include those described in the following patent

documents: U.S. Patent No. 6,091,374; U.S. Patent Application Serial No. 09/670,792,

filed on September 27, 2000; U.S. Patent Application Serial No. 09/753,244, filed on

January 2, 2001; U.S. Patent Application Serial No. 09/753,243, filed on January 2, 2001;

and U.S. Patent Application Serial No. 09/077,340, filed on February 15, 2002; and U.S.

Patent Application Serial No. 09/419,806, all assigned to the assignee of the present

application. Furthermore, one may use wide-band horn antennas and ridged horn

antennas, as desired. As yet another alternative, one may employ a differentially driven

wire segment as a simple, effective, wide-band radiator. In addition, one may use other

suitable wide-band antennas, as persons of ordinary skill in the art who have the benefit

of the description of the invention understand. Note that some antennas are of the "constant gain with frequency" types, and

result in systems that have frequency dependent propagation characteristics. Other

antennas, for example, horn antennas, are of the "constant aperture" variety, and produce

frequency-independent propagation behavior. To use harmonics with relatively high

frequencies, exemplary embodiments according to the invention use "constant aperture

with frequency" antennas, although one may employ other types of antenna, as persons of

ordinary skill in the art who have the benefit of the description of the invention

understand.

Reference clock 41 also couples to timing controller 42. Timing controller 42

clocks the data in data buffer 43. Note that timing signals from timing controller 42 also

clock PN generator 45. Data buffer 43 receives its input data from data port 44. A PN

sequence from PN generator 45 modulates the data from data buffer 43 by using data/PN

combiner 46, in a manner that persons of ordinary skill in the art with the benefit of the

description of the invention understand. PN encoded data from data/PN combiner 46

modulates the one or more harmonics in mixer 47. In illustrative embodiments according

to the invention, data/PN combiner 46 constitutes an exclusive-OR (XOR) gate, although

one may use other suitable circuitry, as persons of ordinary skill in the art with the benefit

of the description of the invention understand. In illustrative embodiments, one may use filters at the output of harmonic

generator 49 to adjust the amplitudes of the one or more harmonics so that have

substantially the same value. Note, however, that in other embodiments according to the

invention, one may use unequal amplitudes, as desired. By using unequal amplitudes,

one may control the amount of energy in the transmitted signals at particular frequencies

or bands of frequencies.

Unequal amplitudes affect the amount of energy in various parts of the

corresponding PSD profile. For example, reduced (or eliminated) amplitudes result in

reduced energy in corresponding frequency bands. (FIG. 16 shows an example of such a

system, where one desires to radiate less energy in band 267 so as to improve coexistence

with radio systems operating within that band.)

FIG. 4 illustrates exemplary waveforms corresponding to high data-rate UWB

transmitter 4. Signal 421 corresponds to the output of harmonic generator 49. Signal 422

corresponds to a relatively short PN sequence of 4 chips per data bit. Signal 423

illustrates a relatively short data sequence. Signal 429, shown to provide more timing

detail for transmitter 4, constitutes the output signal of reference clock 41.

Persons of ordinary skill in the art who have the benefit of the description of the

invention appreciate that, depending on the application, chip sequences longer than 4 chips per bit may be desirable. For example, one may use such chip sequences when the

transmission medium constitutes an RF channel with substantial multipath, or when one

desires more energy per data bit (at the cost of the data throughput rate).

Generally, one may use as few as one chip per bit to obtain the maximum data

rate, as desired. Furthermore, one may employ as many as tens of thousands of chips per

bit in order to obtain "integration" gain at the cost of data rate. Thus, the range for the

number of chips per bit may be very broad, as desired, depending on the design and

performance specifications for a particular application, as persons skilled in the art

understand. For example, in illustrative embodiments according to the invention, one

may generally use 1 to 200 chips per bit, as desired. As another example, in

embodiments that comply with IEEE 802.15, one typically desires data rates as high as

480 Mb/s, corresponding to a few chips per bit, and as low as 11 Mb/s, implying

approximately several hundred chips per bit.

invention appreciate that the number of the PN chips per data bit is a measure of coding

gain useful in mitigating against interference and against multipath impairments. Thus,

using a larger number of chips per data bit provides one mechanism for reducing the

effects of interference and multipath. As noted above, one may implement data/PN combiner 46 using an exclusive-OR

gate. Signal 424 depicts the result of an exclusive-OR operation on signals 422 and 423.

Modulated RF signal 425 results from combining signal 421 and signal 424 in mixer 47.

Timing signal 426 depicts the transmission time for the sequence of data bits 423.

FIG. 5 illustrates an exemplary embodiment of high data-rate UWB receiver 5

according to the invention. Receiver 5 includes reference clock 53, tracking loop 52,

integrator/sampler 51, PN generator 55, data/PN combiner 56, mixer 57, antenna 58, and

harmonic generator 59. Similarly named blocks and components in receiver 5 may have

similar structure and operation as the corresponding blocks and components in transmitter

4 depicted in FIG. 3.

Referring to FIG. 5, in high data-rate UWB receiver 5, receiving antenna 58

couples received modulated signal 425 (shown as the signal coupled to the transmission

medium in FIG. 3, with an exemplary waveform depicted in FIG. 4) to mixer 57. Mixer

57 supplies its output signal to integrator/sampler 51. Integrator/sampler 51 integrates the

output signal of mixer 57 to deliver recovered data bit signal 563 as data output 54.

Mixer 57 also receives template signal 567. Data/PN combiner 56 generates

template signal 567 from an output of PN generator 55 and harmonic generator 59. In

illustrative embodiments according to the invention, data/PN combiner 56 constitutes an exclusive-OR (XOR) gate, although one may use other suitable circuitry, as persons of

ordinary skill in the art with the benefit of the description of the invention understand.

Harmonic generator 59 operates in a similar manner as harmonic generator 49 in FIG. 3,

and may have a similar structure or circuitry.

A tracking loop 52, well known in the art, controls reference clock 53 and PN

generator 55. Tracking loop 52 controls the timing of PN generator 55 for proper signal

acquisition and tracking, as persons of ordinary skill in the art with the benefit of the

description of the invention understand. Reference clock 53 provides reference clock

signal 569 to PN generator 55 and harmonic generator 59.

Note that one may implement tracking loop 52 in a variety of ways, as desired.

The choice of implementation depends on a number of factors, such as design and

performance specifications and characteristics, as persons skilled in the art understand.

Tracking loop 52 operates in conjunction with template signal 567 to provide a locking

mechanism for receiving a transmitted signal (template receiver or matched template

receiver), as persons skilled in the art who have the benefit of the description of the

invention understand.

Mixer 57 mixes the signal received from antenna 58 with template signal 567 to

generate signal 568. Integrator/sampler 51 integrates signal 568 to generate recovered data signal 563. Integrator/sampler 51 drives tracking loop 52, which controls signal

acquisition and tracking in high data-rate UWB receiver 5.

FIG. 6 illustrates exemplary waveforms corresponding to high data-rate UWB

receiver 5. Signal 562 constitutes the output of PN generator 55. Signal 561 conesponds

to the output of harmonic generator 59, whereas signal 567 is the output signal of data PN

combiner. Signal 568 constitutes the output signal of mixer 57, which feeds

integrator/sampler 51. Signal 563 is the output signal of integrator/sampler 51. Finally,

signal 569, shown to provide more timing detail for receiver 5, constitutes the output

signal of reference clock 53.

FIG. 7 shows further details of the timing relationship among various signals in

the high data-rate UWB transmitter 4. Waveform 75 conesponds to the signals in the

transmission medium (i.e., propagated from antenna 48). Waveform 76 shows the

transmission periods, i.e., periods of time during which transmitter 4 transmits. Finally,

waveform 73 illustrates data bit stream 73 during transmission periods 76. Waveform 79

depicts the clock tick marks for timing reference with respect to the other waveforms in

FIG. 7.

In other embodiments according to the invention, one may operate high data-rate

UWB transmitter 4 in either of two modes, depending on a selected or prescribed parameter. Each mode may generate a particular or prescribed PSD profile by using

particular or prescribed harmonic orders (i.e., the choice of the harmonics of the carrier to

use for each mode). By selecting a particular mode, one may operate transmitter 4 such

that it produced output signals that conform to a particular PSD profile or meet prescribed

conditions (as set forth, for example, by a regulatory authority, such as the FCC).

FIG. 8 depicts two exemplary desired or prescribed PSD profiles that conespond

to the two modes of operation in such embodiments. A transmitter according to the

invention may produce outputs that conform to a selected one of predetermined PSD

amplitude profile mask 80 and predetermined PSD amplitude profile mask 81. In an

embodiment of such a transmitter, the frequency of the reference clock (i.e., the

frequency of reference clock 41 in FIG. 3) is approximately 1.8 GHz. Accordingly, the

second and third harmonics appear at approximately 3.6 GHz and 5.4 GHz, respectively.

In a first mode of operation conforming to PSD amplitude profile mask 80, one

modulates the 3.6 GHz carrier (the second harmonic of the reference clock frequency)

with one chip per two RF carrier cycles. Furthermore, one modulates the 5.4 GHz carrier

(the third harmonic of the reference clock frequency) with one chip per three RF cycles.

In this mode of operation, the transmitter has a chipping rate of 1.8 giga-chips per second.

The transmitter produces a transmitted PSD profile 83. Note that transmitted PSD profile 83 has a substantially flat shape, and conforms to PSD mask 80 (i.e., it remains under

PSD mask 80).

In a second operating mode, one suppresses the second harmonic while

modulating the third harmonic 1.80-GHz clock (i.e., the harmonic appearing at 5.6 GHz)

at a rate of one chip per four RF cycles. As a result, the transmitter has a chipping rate of

1.35 giga-chips per second.

Note that one may implement embodiments according to the invention that

include more than two operating modes, as desired. For example, one may provide a

UWB apparatus that includes m operating modes, where m denotes an integer larger than

unity. One may implement such a system in a variety of ways, as persons of ordinary skill

in the art with the benefit of the description of the invention understand. For example,

one may use a bank of selectable harmonic filters (i.e., selectable choice of which

harmonic orders to use) to select any combination of one or more harmonics. Such a

UWB radio apparatus may selectively avoid interference from or with other radio systems

operating in the same band or bands. Note that in illustrative embodiments according to

the invention, one may consider "one or more of m harmonics" as a form of modulation

in addition to the polarity modulation (i.e., BPSK modulation). Although the description above refers to the second and third harmonic, persons

of ordinary skill in the art who have the benefit of the description of the invention

appreciate that one may use other harmonics, as desired. Put another way, in each

operating mode, one may employ additional harmonics beyond the third harmonic.

Using additional harmonics increases the total transmitted power, while simultaneously

conforming to the prescribed respective masks (i.e., remaining under the PSD masks).

FIG. 9 shows a PSD profile for an exemplary embodiment of the invention that

uses higher-order harmonics. Transmitted PSD profile 91 conesponds to modulated third

and fourth harmonics of a 1.1 -GHz reference clock. PSD profile 91 assumes modulation

at the rate of 1.1 giga-chips per second.

If one desired more transmitted power, one may employ the third through seventh

harmonics. Doing so results in transmitted PSD profile 93. Note that both PSD profile

92 and PSD profile 93 have substantially flat shapes. Note further that both PSD profile

92 and PSD profile 93 conform to a prescribed or desired PSD amplitude profile mask

90. Thus, by using a number of harmonics of the reference clock frequency that have an

appropriate order, one may implement communication systems with particular output

power profiles that conform to prescribed PSD profiles, as desired. Note that one may use an appropriate clock reference frequency and associated

harmonics to provide co-existence with other devices that use a particular RF band or

spectrum. For example, in other embodiments according to the invention, the clock

reference parameters and the harmonic carriers are selected so that the PSD of the high

data rate UWB transmissions coexist with wireless devices operating in the 2.4 GHz ISM

band and in the 5 GHz UNII bands.

More specifically, in such embodiments, the reference clock has a frequency of

approximately 1.1 GHz. Furthermore, the transmitter uses as ^"carrier frequencies

modulated at the reference clock rate of approximately 1.1 GHz both the third and fourth

harmonics of the reference clock frequency (i.e., 3.3 GHz and 4.4 GHz, respectively).

FIG. 10 shows an exemplary PSD profile for such an embodiment of the

invention. Transmission PSD profile 101 fits between the 2.4 GHz ISM band 102 and

the 5 GHz UNII bands 103, satisfying a desired level of coexistence. Note that the

communication system can still support a relatively high data-rate. For example, if one

uses 10 PN chips to comprise one data bit, the resulting data rate is 110 megabits per

second (Mb/s).

Signal harmonics may be added with a selectable, desired, or designed degree of

freedom regarding relative phase of the carriers. For example, in a communication system according to the invention that uses the third and fourth harmonics, one may generally

represent the time signals x(t), the sum of the carrier harmonics, by:

x(t) = sin(2^-3/_rt) + sin(2;r-4/_rt + φ),

where/- represents the reference clock frequency and φ denotes a selectable or prescribed

phase angle between 0 and 2π radians. Note that in exemplary embodiments according to

the invention, one may realize the phase angle by using a filter, as persons of ordinary

skill in the art with the benefit of the description of the invention understand.

Note that in exemplary embodiments according to the invention, one may use

various values of φ, as desired, where 0 ≤ φ≤ 2π. FIG. 11A illustrates one cycle of an

exemplary output signal 121 A of a transmitter in a UWB communication system

according to the invention. Signal 121 A conesponds to φ= π. Starting point 122 and

ending point 123 coincide with the chip boundaries, as illustrated, for example, by signal

421 and chip signal 422 (output signal of PN generator) in FIG. 4.

Furthermore, note that one may represent output signal x(t) by using cosines, as

desired. In other words, xi(t) = cos(2π-3-f_rt) + cos(2π-4f_rt + ),

where f_r represents the reference clock frequency and φ denotes a selectable or

prescribed phase angle between 0 and 2π radians (inclusive of the end points). FIG. 1 IB

shows one cycle of another exemplary output signal 12 IB of a transmitter in a UWB

communication system according to the invention. Output signal 121B has starting point

122B and ending point 123B.

Persons skilled in the art with the benefit of the description of the invention

appreciate that It will be appreciated that signals *(t) and x((t) constitute orthogonal

signals. One may therefore use signals x(t) and x (t) to implement quadrature phase shift

keying (QPSK) modulation, as described below.

Note that signals 121 A and 12 IB have relatively small signal levels at both their

starting points (i.e., 122A and 122B, respectively) and their ending points (i.e., 123A and

123B). Exemplary embodiments according to the invention switch signals ON and OFF

at those relatively small signal levels. Doing so tends to avoid switching transients that

with imperfect switching might alter the resulting spectrum undesirably. In illustrative embodiments according to the invention, one may represent the

harmonic carriers by a composite signal S that constitutes a summation of sinusoidal

and/or cosinusoidal signals, i.e.,

S(t) = ∑ s {2π-nf_r(t-s)} ,

where the summation extends over the range of harmonics n desired (i.e., it spans the

order of the desired harmonics, from the lowest to the highest). Put another way, the

composite signal S constitutes a sum of harmonic carriers over a selected range, n. Note

that one may also add cosine harmonics to implement a quadrature UWB communication

apparatus.

As noted above, in some embodiments, n may range from 3 to 4 (conesponding to

a UWB communication apparatus operating in a desired 3.1 GHz to 5.2 GHz frequency

range). FIG. 12 shows the timing relationship between several signals in such an

embodiment according to the invention, with n = 3. Signal 139 depicts a reference clock

signal, included to facilitate presentation of the timing relationship between the various

signals. Signal 131 conesponds to composite signal S, described above. Signal 132

denotes the sinusoidal signal the harmonics of which result in composite signal 131.

Reference clock signal 139 conesponds to the positive-going zero-crossings of sinusoidal

signal 132. Note that time displacement s offsets the chipping signal from the carrier signal.

More specifically, time displacement s appears as an offset between reference clock

signal 139 (or sinusoidal signal 132) and the chipping signals.

FIG. 12 shows signals conesponding to several values of time displacement s.

Each time displacement s signifies the offset between reference clock signal 139 (or

sinusoidal signal 132) and one of chipping signal 133, chipping signal 134, and chipping

signal 135, respectively. Specifically, chipping signal 133 conesponds to a time

displacement s of zero. Chipping signal 134 and chipping signal 135 denote,

respectively, time displacements of 0.25 and 0.5, respectively.

invention appreciate that, because of symmetry, negative values of s give the same results

as positive values of s. Hence, the description of the invention refers to the magnitude of

s, or \s\. Also, note that, although FIG. 12 illustrates the chipping sequence "101" as an

example for the sake of illustration, persons skilled in the art with the benefit of the

description of the invention understand that one may generally use a desired PN

sequence. FIG. 13 illusfrates several PSD profiles for an illustrative embodiment according

to the invention. PSD profile 143 depicts the power spectral density of signal 131

multiplied by PN chipping sequence 133. Similarly, PSD profile 144 conesponds to the

power spectral density of signal 131 multiplied by PN chipping sequence 134. Finally,

PSD profile 145 illustrates the power spectral density of signal 131 multiplied by PN

chipping sequence 135.

FIG. 13 also illustrates boundary 146 of the 2.4 GHz ISM band and boundary 147

of the UNII band. For two harmonics, a time displacement value \s\ = 0.25 provides a

substantially flat PSD profile 144. Persons of ordinary skill in the art who have the

benefit of the description of the invention understand, however, that one may use time

displacement values (s) in a range of approximately 0.1 and approximately 0.9 to provide

substantially similar PSD profiles for the third and fourth harmonics, as desired. In a

similar manner, one may use other values of time displacement ,s and appropriate

numbers of harmonics to implement communication systems having desired or prescribed

PSD profiles, as desired.

As an example, FIG. 14 depicts several PSD profiles that conespond to exemplary

embodiments of the invention that use increasing numbers of harmonics. FIG. 14

includes PSD profile 151, PSD profile 152, and PSD profile 153. A substantially flat

PSD profile 151 conesponds to a signal that includes the fundamental frequency through the seventh harmonic, using a time displacement value of \s\ = 0.375. Similarly, PSD

profile 152 pertains to a signal that includes the second through the seventh harmonics,

using a time displacement value of \s\ = 0.375. Finally, PSD profile 153 conesponds to a

signal that includes the third through the seventh harmonics and uses a time displacement

value of |s| = 0.375.

Note that values of time displacement s between approximately 0.1 and

approximately 0.9 provide substantially flat PSD profiles, similar to the PSD profiles that

FIG. 14 illustrates. As noted above, using larger numbers of harmonics while

conforming to PSD profiles (i.e., constrained to a maximum PSD value) results in an

increase in the total transmitted or radiated power.

One may generate and implement the time displacement s in variety of ways, as

persons of ordinary skill in the art who have the benefit of the description of the invention

understand. For example, one may implement s as digitally derived clock shift in timing

controller 42 of transmitter 4 and PN generator 55 in receiver 4. As another example, one

may implement the desired time shift by using a physical delay line in the path of the

digital input of mixer 47 in transmitter 4 and mixer 57 in receiver 5.

One may obtain the spectra shown in the figures by computing the Fourier

Transform of the composite signal S. More specifically, where the data pulses have a generally rectangular shape and have not been filtered (e.g., chipping signal 422 in FIG.

4), one may obtain the PSD as:

where/- denotes the chipping clock frequency, and «, and n₂ conespond to the order of

the harmonics used (i.e., the lower and upper boundaries of the range of harmonics used).

Note that one may omit selected harmonics within the range «, to n₂ to further shape the

spectrum, as desired. FIG. 15 shows an example of applying this technique.

Referring to FIG. 15, PSD profile 161 shows the power spectral density for an

embodiment of a communication system according to the invention that uses the third

through seventh harmonics of a 1.1 -GHz clock. In contrast, PSD profile 162 conesponds

to a system that employs the fifth through the seventh harmonics. As a result, the PSD

energy in the latter system lies mostly above 5 GHz.

As a third example, PSD profile 163 conesponds to a system that uses the third,

fourth, sixth, and seventh harmonics. Omitting the fifth harmonic in this system results

in a gap in the vicinity of 5 GHz to 6 GHz. As a result, the system may effectively coexist with systems that operate in the 5-GHz UNII band. Note that one may use

filtering to readily remove energy in the side lobes shown in FIG. 15.

The PSD profiles shown in FIG. 15 conespond to illustrative embodiments of

communication systems according to the invention. By judiciously employing selected

harmonics together with a chosen clock frequency, one may design and implement a wide

variety of communication systems with prescribed PSD profiles in a flexible manner.

The choice of design parameters (e.g., clock frequency and the number and order of

harmonics) depend on desired design and performance specifications and fall within the

knowledge of persons of ordinary skill in the art who have the benefit of the description

of the invention.

FIG. 16 shows PSD profiles for other exemplary embodiments of communication

systems or apparatus according to the invention. These embodiment conform with a PSD

mask in which the emissions at 3.1 GHz are at least -10 dB from the peak (marker labeled

as 265 in FIG. 16). Furthermore, the mask specifies emissions at 10.6 GHz of at least -10

dB from the peak (marker denoted as 266 in FIG. 16).

UNII band 267 extends from 5.15 GHz to approximately 5.9 GHz. FIG. 16

illustrates four PSD profiles (denoted as profiles 261, 262, 263, and 264, respectively)

fhat conespond to different choices of the order of harmonics used. All four PSD profiles conespond to a baseband chipping reference clock frequency of 1.4 GHz. Furthermore,

the PSD profiles assume time displacement s of approximately 0.375 between the

reference clock signal and the chipping sequences (see FIG. 13 and accompanying

description for an explanation of time displacement s and its effect on PSD profiles).

As noted above, PSD profiles 261, 262, 263, and 264 denote various choices of

the order of harmonics used. PSD profile 261 conesponds to a communication system

that uses the 3rd through the 7th harmonics of the chipping reference clock. Thus, such a

system effectively occupies the allowed bandwidth between 3.1 GHz and 10.6 GHz.

PSD profile 262 conesponds to a system that employs the 3rd, the 5th, the 6th,

and the 7th harmonics of the chipping reference clock. In other words, unlike the system

conesponding to PSD profile 261, it omits the fourth harmonic, which overlaps UNII

band 267.

The system conesponding to PSD profile 263 uses the 3rd through the 6th

harmonics of the chipping reference clock. Thus, this system omits the relatively higher

frequencies by not using higher-order harmonics.

PSD profile 264 pertains to a communication system that uses the 3rd, the 5th,

and the 6th harmonics of the chipping reference clock. This system omits the fourth harmonic, which overlaps UNII band 267. The system may switch its operation modes

between PSD profile 261 and PSD profile 262 or, alternatively, between PSD profile 263

and PSD profile 264, as described below in detail.

Table 1 below summarizes the harmonics used in the systems conesponding to

PSD profiles 261, 262, 263, and 264:

TABLE 1

As noted above, communication systems according to exemplary embodiments of

the invention may include multi-mode operation. Such systems may switch from one

mode of operation to another mode of operation based on desired or prescribed conditions

or stimulus. Referring to FIG. 3, controller input signal 40 enables mode switching in

transmitter 4. The state of controller input signal 40, transmitter 4 and, more specifically,

timing controller 42, determines the chipping duration relative to the reference clock cycle in a manner apparent to persons of ordinary skill in the art who have the benefit of

the description of the invention.

Communication systems according to the invention may perform mode switching

in response to virtually any stimulus, as desired. For example, a system user may

manually selection the mode and thus cause mode switching. As an alternative, the mode

switching may occur in an automatic manner, for instance, in response to predetermined

or selected system event.

As another example, the mode switching may occur in a semi-automatic manner,

but involve manual user selection in response to an event flagged or brought to the user's

attention. In other embodiments, an internal or external variable or quantity, for example,

time, may control mode switching. Alternatively, a remote signal received by the

communication system may switch the operating mode.

As yet another example, communications systems and apparatus according to

various embodiments of the invention may switch modes in response to the detection of

radio-signal energy in a desired band or bands. For example, in response to detecting the

presence of radio-signal energy in the UNU bands (between 5.15 GHz and 5.85 GHz), a

UWB communication apparatus or system according to the invention may switch its

mode of operation so that its transmissions have a prescribed spectral content. The new mode of operation may conespond to a PSD profile that tends to eliminate, reduce, or

minimize interference with any devices operating in the particular band of interest. For

example, the new PSD profile may constitute PSD profile 163 in FIG. 15.

Thus, the stimulus for the switching of modes in such systems is the detection of

the presence of RF signals from devices operating in a particular band or at a particular

frequency or plurality of frequencies, such as UNII band devices. The response of the

communication system or apparatus constitutes switching modes so as to eliminate or

minimize interference, for example, by omitting the harmonic component that would

result in transmitted energy in the affected frequency range or band. Such a feature

provides an additional measure of coexistence with devices operating in existing radio

frequency bands, such as UNII radio devices.

Note that the above examples constitute only a sampling of how one may switch

the operating mode. Depending on desired design and performance specifications, one

may use other techniques and mechanisms for mode switching, as persons of ordinary

skill in the art who have the benefit of the description of the invention understand.

Furthermore, one may apply any of these techniques to various embodiments of

communication systems and apparatus according to the invention, as desired. FIG. 17 shows an exemplary embodiment according to the invention of a

communication system that incorporates mode switching. High data-rate UWB

communication system 11 includes transceiver 111, which has internal power source 112

(e.g., a battery or other power source). System 11 also includes second transceiver 113,

with its internal power source 114 (e.g., a battery or other power source). The mode

switching in system 11 occurs depending on whether the system operates from its internal

power sources or from an external power source (not shown explicitly in FIG. 17).

When system 11 uses internal power source 112 and internal power source 114, it

may operate in a mode that conforms to a particular PSD profile, for example, PSD mask

81 in FIG. 8. This mode may conespond, for example, to system operation indoors. PSD

mask 81, conesponding to indoors operation, may have more relaxed requirements

because system 11 may cause less potential interference with other systems while it

operates indoors.

Conversely, when system 11 uses external power (supplied through port 115 to

transceiver 111 and supplied through port 116 to transceiver 113), it may operate in

another mode that conforms to a different PSD profile, for example, PSD mask 82 in

FIG. 8. The second mode may conespond, for example, to system operation outdoors.

Thus, by switching operation modes, UWB communication systems according to the invention can meet more stringent PSD masks outdoors and yet conform to a more

relaxed PSD mask while operating indoors.

To switch modes, system 11 senses the application of external power, and

supplies a trigger signal to controller input 40 of the transmitter (see FIG. 3). In response,

timing controller 42 and harmonic generator 49 adjust pre-determined timing parameters

to generate the desired PSD profile, as described above in reference to FIG. 8. An

analogous operation occurs in the receiver circuitry of the transceiver. Furthermore, a

companion or conesponding transceiver similarly adjusts parameters in its transmitter

circuitry and receiver circuitry in response to the particular PSD profile that the receiver

circuitry receives.

Note that, although FIG. 17 shows a pair of transceivers, alternative systems may

include a transceiver and a receiver, or a transmitter or receiver, as desired. Mode

switching in such systems occurs using a similar technique and mechanism as described

above, as persons skilled in the art with the benefit of the description of the invention

understand.

Another aspect of the invention relates to the shape of the pulses within chipping

signal 422 (reproduced in FIG. 18 for convenience). Chipping signal 422 includes pulses

with generally rectangular shapes. As a consequence, one may generally obtain the spectrum in the frequency domain of the chip as given approximately by the well-known

siπf ) sine function, . (A chip conesponds to the distance in time between the vertical x segments of signal 422, or the zero-crossings of signal 222.) The multiplication operation

in mixer 47 shifts that spectrum in the frequency domain and centers a copy of the

spectrum at each of the harmonic signals present in signal 421 (output signal of harmonic

generator 49).

Although the description above assumes a chipping signal with pulses that

generally have a rectangular shape (e.g., chipping signal 422), one may use other pulse

shapes, as desired. For example, the pulses may have a more "rounded" shape.

One example of a more "rounded" pulse shape is the Gaussian impulse.

Mathematically, one may represent a Gaussian impulse s(t) as:

-o.sr s(t) = e

where t represents time, and τ denotes a parameter that defines the pulse width. One may

obtain the shape of the spectrum in the frequency domain by using the Fourier transform

of s(t). Mathematically, one may express the Fourier spectrum of s(t) as: S(f) = e^{~2( )2}.

Using the above relationships, one may design a pulse of width conesponding to

frequency fβ (for example 1.1 GHz) where the magnitude of S(f) is below a reference

value by a desired amount (for example, by 10 log[S(/?)]= -10 dB). This technique

provides a design value for τ, which in turn allows one to evaluate s(t).

Note that FIG. 18 shows a Gaussian impulse as one example. One may use other

shapes, as desired, as persons of ordinary skill in the art who have the benefit of the

description of the invention understand. For example, one may use the trapezoidal shape

of chipping signal 133, chipping signal 134, and chipping signal 135 in FIG. 12, as

desired.

Furthermore, note that by shaping or filtering the pulses before mixing with a

signal having a relatively high frequency (a harmonic signal), one avoids designing or

shaping pulses at those relatively high frequencies. In the case of a filtered signal, one

may obtain the PSD from:

PSD dt,

where p(t) denotes the baseband filtered data signal. On example is a Gaussian filtered

signal, such as one chip of chipping sequence 222 in FIG. 18. Also, note that by using

multiple harmonics, one may shift the shaped pulses in the frequency domain and center

the shifted versions at the desired harmonic carriers.

Although FIG. 18 shows chipping sequence 422 and chipping sequence 222 as

having +1 and -1 amplitude swings, one may use other swings, as persons of ordinary

skill in the art who have the benefit of the description of the invention understand. For

example, one may implement chipping sequences that use +1, 0, and -1 amplitude

swings, as desired.

One may use various modulation schemes and techniques in communication

systems and apparatus according to the invention, as desired. For example, exemplary

embodiments of the invention may use techniques analogous to the conventional

quadrature phase shift keying (QPS ) systems. Other exemplary embodiments according

to the invention may use techniques analogous to offset QPSK (OQPSK).

More particularly, embodiments using QPSK use two harmonic carriers, which

requires two degrees of freedom so that both pairs of harmonically related signals have a

quadrature relationship. Specifically, the phase difference between the two reference clocks and an additional phase delay in one of the harmonic generator lines provide the

two desired degrees of freedom. A QPSK-like UWB system according to the invention

with two harmonic carriers has the desired property of providing a data rate twice the data

rate of a BPSK-like system, while still having an essentially flat PSD profile that

conforms to prescribed or desired criteria.

Providing an additional half chip length offset between the two data streams

modulating the quadrature harmonic carriers provides an OQPSK system. Such an

OQPSK system has the additional desirable property of a smoothed PSD spectrum or

profile relative to the PSD profile of the QPSK system.

FIG. 20 shows one example of the waveforms of an OQPSK UWB signal set in

an illustrative embodiment. Signal 2110 comprises sinusoidal harmonics, such as the

signal shown in FIG 11 A, while signal 2130 comprises cosinusoidal harmonics, like the

signal FIG. 11B illustrates. Data sfream 2120 modifies the polarity of signal 2110, and

data stream 2140 modifies the polarity of signal 2130, independent of data signal 2120.

The signal 2130 is furthermore shifted in time to the right of signal 2110 so that

the maximum envelope value 2135 of signal 2130 substantially conesponds with the

minimum envelope value 2115 of signal 2110. Additionally, to maintain quadrature, the

zero-crossings of signal 2110 conespond to the respective signal peaks of signal 2130. Conversely, the zero-crossings of signal 2130 conespond to the respective peaks of signal

2110.

Signal 2150 represents the sum of quadrature signals 2110 and 2130. Persons of

ordinary skill in the art with the benefit of the description of the invention appreciate that

the peak-to-average value of the composite signal is smaller than the peak-to-average

values of either signal 2110 or signal 2130. This property results in a smoother PSD

profile, and enables RF transmissions at a power level that requires less 'safety' margin to

the regulatory limit levels.

In other embodiments according to the invention, one may use a differential phase

shift keying (DPSK) scheme. One may modify a transmitter according to the invention,

for example, transmitter 4 in FIG. 3, to generate DPSK signals, as persons of ordinary

Transmitter 4 generates DPSK signals as follows. Referring to FIG. 3, transmitter 4

receives data at data input 44. Transmitter 4 encodes the data differentially, similar to

conventional DPSK. More specifically, transmitter 4 encodes the data as changes in the

bit stream.

For example, suppose the sequence starts with a binary "1" bit. If the next bit is a

"1," it indicates that transmitter 4 had sent a "0" previously (no change). On the other hand, if a "0" follows the original "1," then transmitter 4 encodes a "1." Thus, transmitter

4 encodes changes from 1 to -1 (or -1 to 1) as binary "l"s. Conversely, transmitter 4

encodes no bit-to-bit change (e.g., 1 followed by 1, or -1 followed by -1) as binary "0"s.

As the above description makes evident, to transmit m bits, one transmits m + 1 bits (a

starting bit, followed by m bits of data), because the changes in the input data bits encode

the data.

Referring to FIG. 3, data buffer 43 may perform the differential encoding

described above. PN generator 45 generates chip sequences associated with a delay or

time period D that equals the number of chips for a single data bit. The time delay D may

be one chip time in one exemplary embodiment, and may constitute a coded sequence of

bits in another illustrative embodiment (for example, D may be the number of chips

associated with a single data bit). Put another way, one may use a per-chip (time period

between starts of two chips) or per-bit (time period between the starts of two bits) time

delay D. Regardless of the choice of time delay D, one keeps D constant for that system.

In exemplary embodiments according to the invention, one may generate chip

sequences by using Barker codes or sequences. Each chip sequence is equal in length to

one of the known Barker sequences. Preferably, transmitter 4 uses Barker sequences of

length 13, 11, or 7, but as persons of ordinary skill in the art who have the benefit of the description of the invention understand, one may use other Barker sequences to provide

chip sequences, as desired. Table 2 below lists the known Barker codes:

TABLE 2

As persons skilled in the art understand, the reverse of the code sequences in

Table 2 also constitute Barker codes. Furthermore, the inverse of the listed code

sequences (i.e., code sequences obtained by replacing 1 with -1 and vice- versa) are

Barker codes.

Note that, rather than using Barker codes, one may use other types of code, as

understand. For example, one may use Kasami codes, as desired. Other examples

includes Hadamard codes, Walsh codes, and codes that have low cross-conelation

properties. PN generator 45 multiples each bit obtained from data buffer 43 with the Barker

sequence. Accordingly, the signal 424 (output signal of data/PN combiner 46) constitutes

either the Barker sequence or the inverse of a Barker sequence (i.e., obtained by

multiplying by -1 the code sequences in Table 2). Assuming, for example, that PN

generator uses a Barker code of length 11 , the time period or delay D equals the length of

11 chips. As another example, FIG. 12 illustrates one chip time, which relates to Barker

chips in FIG. 6 (signal 562), relating to a Barker code of length 4).

FIG. 19 illustrates an exemplary embodiment 19 of a differential receiver

according to the invention that is suitable for receiving DPSK signals. Receiver 19

includes antenna 910, mixer 916, integrator 918, sample-and-hold 920, and analog-to-

digital converter (ADC) 922. Receiver 19 may optionally include amplifier 912 and

amplifier 914.

Antenna 910 receives differentially encoded signals. Amplifier 912 amplifies the

received signal and provides the resulting signal to one input of mixer 916 and amplifier

914. Through delay device 916, the output signal of amplifier 916 (if used) couples to

another input of mixer 916. The delay D provided by delay device 916 equals one bit time. Accordingly,

mixer 916 multiplies the received signal by a version of the received signal delayed by a

time period D. Because of the differential coding of the signals (described above), a bit

sign in the delayed version of the received signal changes when receiver 19 receives a

binary "1."

The output of mixer 916 feeds integrator 918. The output of mixer 916

constitutes a +1 Barker sequence of Table 2 multiplied by an inverse Barker sequence,

thus resulting in a negative going voltage at the output of integrator 918 over the length

of the Barker code. Sample-and-hold 920 samples the output signal of integrator 918

when that signal crosses a threshold. Sample-and-hold 920 provides the sampled signal

to ADC 922. ADC 922 provides output data bits.

Note that, in illustrative embodiments, the length of the integration may be the

time period D. Based on design and performance specifications, however, one may use

longer or shorter time periods, as persons of ordinary skill in the art who have the benefit

of the description of the invention understand.

Optional amplifiers 912 and optional amplifier 914 may constitute either linear

amplifiers or limiting amplifiers, as desired. One may additionally use amplifier 914 to compensate for any losses in delay device 916. Note that one may place amplifier 912 as

shown in FIG. 19 or, alternatively, after delay device 916.

One may implement delay device 916 in a variety of ways, as persons of ordinary

example, a relatively simple delay device comprises a length of transmission line that has

electrical length D. One may use a length of coaxial line, printed strip-line, or microstrip

in various ways to realize such a device.

Implementing amplifier 912 and amplifier 914 as limiting amplifiers relaxes the

design demands on mixer 916. Mixer 916 may have a variety of structures and circuitry,

as persons of ordinary skill in the art who have the benefit of the description of the

invention understand. For example, mixer 916 may constitute a passive ring diode mixer

or a four-quadrant multiplier, as desired.

In conventional DPSK systems, the data bits constitute a length D equal to the

length of one data bit. Such systems modulate the phase of the carrier (0 or τd2 radians)

at the bit rate. In contrast, communication systems or apparatus according to the

invention use a Barker encoded sequence of harmonic wavelets (as shown, for example,

in FIG. 6) instead of the carrier in conventional systems. Communication systems or

apparatus according to the invention modulate the polarity of the wavelets (i.e., +1 or -1) at the chip rate. Furthermore, they polarity modulate the chip sequences at the bit rate.

Thus, in contrast to conventional DPSK systems, in communication systems and

apparatus according to the invention, the bit time (see signal 563 in FIG. 6) comprises a

coded sequence of wavelets.

Note that receiver 19 and associated circuitry may perform additional

functionality. For example, such circuitry may recover the data bits, recover timing of

the chip sequences, and fine tune the integration time of integrator 918 in response to

signal quality, as persons of ordinary skill in the art who have the benefit of the

description of the invention understand.

Note that the exemplary embodiments described above associate each data bit

with a spreading code of length N. More specifically, one may use Barker codes of

lengths N= 2, 3, 4, 5, 7, 11, and 13. Thus, one may associate N chips with a single data

bit. As an example, using a Barker code of length 7 (see Table 2, above), one may

transmit a "1" by using the sequence 1 1 1 -1 -1 1 -1. Similarly, to transmit a "0," one

may use the sequence -1 -1 -1 1 1 -1 1 (i.e., a sequence obtained by multiplying by -1

each number in the previous sequence).

In other embodiments according to the invention, one may use codes that have a

larger length than needed to encode a single bit. Doing so may have several advantages. First, the spectrum of the resulting signal more closely resembles white noise (i.e., the

benefit of spectrum "whiteness").

Second, one may use such codes to provide channelization. Longer codes have a

relatively large number of nearly-orthogonal family members. One may use such family

members to represent both various symbols (i.e., groups of bits) and to provide more

effective channelization.

As an example, one may use PN sequence generated which the TIA-95 code

division multiple access (CDMA). Such a sequence is 32,768 chips long. One may

define channels and symbols by multiplying (e.g., by using an exclusive-OR operation)

the PN sequence (at the chipping rate) with a Hadamard code or a Walsh code (i.e.,

repeated sequences like 1111111100000000, 1111000011110000, 1100110011001100,

and so on, as persons skilled in the art understand). Thus, groups of chips are uniquely

identify a symbol or channel. Such a techniques takes advantage of a code length of

32,767 to obtain a signal with a relatively smooth spectrum.

In addition to using relatively long codes to provide channelization, one may use

other techniques, such as such as time-division multiplexing and space-division

multiplexing (using directional antenna techniques to isolate links), as desired. Such techniques fall within the knowledge of persons of ordinary skill in the art who have the

benefit of the description of the invention.

In addition to coding the transmitted data in embodiments according to the

invention as described above, one may provide enor-conection coding (ECC), as desired.

For example, one may apply ECC to data input 44 in FIG. 3, as desired. Many such

codes exist in the art, and one may apply them to communication systems and apparatus

according to the invention as persons skilled in the art with the benefit of the description

of the invention understand. Examples of such codes include BCH codes, Reed-Solomon

codes, and Hamming codes.

As noted above, the carrier signal (e.g., carrier signal 21 in FIG. 2) may constitute

a sinusoidal or non-sinusoidal carrier signal. FIG. 21 shows examples of some signal

waveforms conesponding to a non-sinusoidal carrier signal. FIG. 21 includes a repeating

pattern "1010" of chips 2022. Signal 2021 conesponds to the "1010" repeating pattern of

chips. As FIG. 21 illustrates, signal 2021 may have a gap 2023 of an arbitrary length

(with the parameters of signal 2022, of course) between its segments.

Another aspect of the invention relates to multiple independently modulated

harmonic signals (e.g., harmonics of a given frequency, such as a clock frequency). In

other words, in communication apparatus according to the invention, one may modulate various harmonic signals with either the same data stream, or independently, each (or a

set) with a different data stream. Thus, the effective data rate constitutes the sum of all

the data rates that modulate the harmonic signals.

Furthermore, one may selectively enable or turn ON each harmonic signal, as

desired. Put another way, one may configure the harmonic signals independently. In one

configuration, the harmonic signals are not ON or enabled simultaneously. In effect, one

may hope from one harmonic signal or frequency to another harmonic signal or frequency

as a function of time, as desired.

Configuring the harmonic signals by turning them ON selectively has a benefit of

simplifying the communication apparatus or system. The communication apparatus or

system may operate in the presence of multipath interference without a need to resort to

coding. More specifically, such apparatus or systems may operate in an environment

where multipath effects are present without having to code the signals that modulate each

harmomc signal (as the embodiments described above do). Note, of course, that one may

still use coding, as desired, but one need not do so to combat the effects of multipath

interference.

To combat the effects of multipath interference, communication apparatus or

systems according to the invention transmit one impulse on a given harmonic frequency or channel and then wait for the multipath echoes on that channel to decay before

transmitting again. For example, suppose that multipath interference in a given

environment has a delay spread of 25 ns. Thus, it takes about 20 ns for the echoes present

because of multipath to decay, before one may receive the next impulse or signal (product

of the harmonic signal and a signal chip or bit that carries one datum bit).

By using multiple harmonic signals (i.e., two or more harmonics) or frequencies,

one may transmit multiple data bits. In other words, one may transmit a first datum bit

on the frequency of a first harmonic signal, then transmit a second datum bit on the

frequency of a second harmonic signal, and so on, until one transmits the final datum bit

(say, datum bit N) using the Nth harmonic signal. One may then repeat this cycle, as

desired.

The delay between subsequent transmissions using a given harmomc signal

allows the multipath echoes to decay, so that echoes from one transmission do not

interfere with a subsequent transmission that uses that harmonic signal. In effect, one

takes advantage of the fact that sufficient numbers of the frequency-time combinations

exist that before one transmits again using a given harmonic frequency, the multipath

echoes present at that frequency have decayed sufficiently. Furthermore, by spacing the

transmission frequencies sufficiently, one may reduce interference from multipath echoes

of one harmonic frequency with transmissions on another harmonic frequency. FIG. 22 shows an exemplary embodiment of a transmitter 2200 according to the

invention that uses independently modulated harmonic signals. Note that dashed lines in

FIG. 22 separate circuitry that operates at relatively lower frequency from other circuitry

that operates at relatively high frequency. One may include the lower-frequency circuitry

in one IC and include the higher-frequency circuitry in another IC, as desired.

Reference clock 41 generates a signal with a desired frequency, for example, a

sinewave with a frequency f_osc. One may implement reference clock 41 in a number of

ways and by using various techniques that fall within the knowledge of persons skilled in

the art with the benefit of the description of the invention.

Reference clock 41 couples to harmonic generator 2220. Based a clock signal it

receives from reference clock 41, harmomc generator 2220 generates an mtb harmonic

signal of the frequency of clock reference 41. For example, given a clock frequency foso

a second harmonic signal at the output of harmomc generator 2220 has a frequency

2f' osc_> ^an^ so on, such that, generally, the th harmonic signal has a frequency mf- _osc.

Note that one may vary m during operation of transmitter 2200, as desired. More

specifically, one may vary m per data bit, or on a chip-by-chip basis. By varying m, one

may generate a desired harmonic signal that has a given frequency. Thus, by using m = 3, one may generate the third harmonic or, by using m = 9, one may generate the ninth harmonic, and so on.

As persons of ordinary skill in the art who have the benefit of the description of

the invention understand, one may realize harmonic generator 2220 in a number of ways,

similar to harmonic generator 49, described above. As one example, one may use a

frequency synthesizer. By varying the control signal of the frequency synthesizer (e.g., a

control voltage), one may vary the output frequency of the frequency synthesizer. Thus,

by applying a level of the control signal that conesponds to a desired value of m, one may

generate the desired harmonic, as desired.

Harmonic generator 2220 generates the harmonic signals synchronously with

respect to the reference clock. One may use a variety of circuitry and techniques to

synchronize the one or more harmonics to the reference clock. Such circuitry and

techniques fall within the knowledge of persons of ordinary skill in the art who have the

benefit of the description of the invention, as discussed above.

Transmitter 2200 may also include signal shaping circuitry 2218 and mixer 2202.

Using signal shaping circuitry 2218 and mixer 2202, one may shape (or filter) data

signals 2206, as desired, and as described below in detail. In embodiments where one uses that option, mixer 2202 generates an output signal 2208 that constitutes shaped data

pulses.

Transmitter 2200 also includes mixer 2204 and antenna 48. Output signal 2208

feeds one input of mixer 2204. Output signal 2210 of harmonic generator 2220 feeds

another input of mixer 2204. The output signal of mixer 2204 constitutes modulated RF

signals 2212. Antenna 48 accepts modulated RF signals 2212 from mixer 2204 and

propagates them into a transmission medium.

Note that, by varying the value of m, one may cause transmitter 2200 to

heterodyne operating frequency of output signal 2208 of mixer 2202 (shaped data pulses)

to a different RF frequency. In other words, by varying the value of m as a function of

time, one may cause the output frequency of fransmitter 2200 to hop to various

frequencies as a function of time, as described above. One may vary the value of m in a

variety of ways, as persons of ordinary skill in the art who have the benefit of the

description of the invention understand. For example, one may use a controller (not

shown explicitly) to control various functions of transmitter 2200, including selecting the

value of m, as desired.

invention understand, one may use integer or non-integer (e.g., fractional) values of m, as desired. Thus, in general, one may derive operating frequency of output signal 2208 of

mixer 2202 by using integer or non-integer values of m, as desired. Put another way,

operating frequency of output signal 2208 of mixer 2202 need not (but may) constitute an

integer harmonic of the clock signal. Rather, it may relate to the clock frequency in any

desired or arbitrary way. For example, the clock frequency may constitute a fraction of

operating frequency of output signal 2208. Furthermore, one may use frequency

synthesizers, such as fractional-M synthesizers, to generate such operating frequencies, as

persons of ordinary skill in the art who have the benefit of the description of the invention understand.

Furthermore, one may modulate intelligence or information signals in a variety of

ways to generate data signals 2206, as desired. By way of illustration, one may apply

BPSK modulation, quadrature amplitude modulation (QAM), and QPSK modulation, and

the like, as described above and understood in the art. The choice of the modulation

scheme depends on design and performance specifications for a particular

implementation, as persons of ordinary skill in the art with the benefit of the description

of the invention understand.

FIG. 23 illustrates an exemplary embodiment of a receiver 2300 according to the

invention for receiving independently modulated harmomc signals. Receiver 2300

includes antenna 58, mixer 2314, mixer 2316, integrator/sampler (integrator/sarnple-and- hold) 2303, controller 2306, baseband template generator 2312, phase-locked loop (PLL)

2319, and harmonic generator 2220.

Antenna 58 receives RF signals and provides them to one input of mixer 2314.

Output signal of mixer 2316 constitutes a second input of mixer 2314. Baseband

template generator 2312 generates a template signal that constitutes one input of mixer

2316. The output of harmonic generator 2220 constitutes a second input of mixer 2316.

The output of baseband template generator 2312 matches the output of signal

shaping circuitry 2218 in transmitter 2200 (see FIG. 22). Baseband template generator

2312 generates its output under the control of PLL 2319. Using feedback within receiver

2300, PLL 2319 generates a first output signal, reference signal 2322, which it provides

to baseband template generator 2312.

When receiver 2300 locks onto a desired RF signal, reference signal 2322

constitutes the same as the reference signal used in the conesponding transmitter for the

RF signal. For example, referring to FIGS. 22 and 23, when receiver 2300 locks onto the

signal transmitted by transmitter 220, reference signal 2322 constitutes a signal similar to

the reference signal that clock reference 41 generates (see FIG. 22). In other words, PLL

2319 generates reference signal 2322 such that it has a frequency f_osc. PLL 2319 generates a second output signal 2328, which has a frequency f_osc, that

feeds harmonic generator 2220. Harmonic generator 2220 operates as described above in

connection with transmitter 2200 in FIG. 22. Thus, harmonic generator 2220 provides a

harmonic signal to mixer 2316 that has a frequency mf_osc.

As noted above, the output of mixer 2316 feeds one input of mixer 2314.

Receiver 2300 uses the output of mixer 2314 to control the feedback loop that includes

PLL 2319 so that the output of mixer 2316 matches the RF signals received from antenna

58. The control loop includes integrator/sampler 2303, controller 2306, and PLL 2319.

The output of mixer 2314 feeds the input of integrator/sampler 2303. Depending

on the datum value that receiver 2300 receives, integrator/sampler 2303 provides one of

two voltage levels as its output. For example, if receiver 2300 receives a binary zero, the

output of integrator/sampler 2303 may constitute a negative voltage. Conversely, if

receiver 2300 receives a binary one, integrator/sampler 2303 may provide a positive

voltage as its output.

The output of integrator/sampler 2303 feeds an input of controller 2306.

Controller 2306 generates a datum value depending on the voltage level it receives from

integrator/sampler 2303. For example, in response to a positive voltage present at the output of integrator/sampler 2303, controller 2306 may generate a binary one bit that has

desired digital level.

Note that controller 2306 may perform filtering, shaping, and the like, of the data

signals, as desired. Controller 2306 also provides feedback control signal 2325 to PLL

2319, thus affecting the frequency of the signals that PLL 2319 generates. Furthermore,

controller 2306 decides the value of m and provides that value to harmonic generator

2220.

As noted above, harmonic generator 2220 generates as its output the røfh

harmonic of output signal 2328 of PLL 2319. Note that one determines the sequence of

the values of m as a function of time for both the receiver and the transmitter. While

operating, the receiver and the transmitter use various values of m according to the pre¬

determined sequence.

Note that one may implement the feedback loop within receiver 2300 in a variety

of ways, as desired. The choice of implementation depends on a number of factors, such

as design and performance specifications and characteristics, as persons skilled in the art

with the benefit of the description of the invention understand. The feedback loop uses

baseband template generator 2312 to provide a locking mechanism for receiving a transmitted signal (i.e., a template receiver or matched template receiver), as persons

skilled in the art who have the benefit of the description of the invention understand.

As noted above, by varying the value of m, communication apparatus and systems

according to the invention may use various frequency channels. Furthermore, varying the

value of m as a function of time varies the use of those channels as a function of time.

Thus, one may specify a channel frequency plan and a channel timing plan for

communication apparatus and systems according to the invention. By varying the

frequency and channel timing plans, one may design and implement a wide variety of

communication apparatus and system, as desired.

Table 3 below shows an example of a channel frequency and timing plan in an

illustrative embodiment of a communication apparatus or system according to the

invention:

TABLE 3

The example in Table 3 conesponds to a communication apparatus or system that

uses six channels. Furthermore, the apparatus or system uses six time slots, each with an

8 ns duration. Thus, the time slots repeat at 48 ns intervals. The harmonics used range

from the 28th harmonic to the 38th harmonic. Put another way, m ranges from 28 to 38.

With a clock reference frequency of 125 MHz, the channels range in frequency from 3.50

GHz to 4.75 GHz.

More specifically, at time t = 0, m = 30 conesponds to a harmonic frequency of

3.75 GHz. That frequency conesponds to channel 2. Eight nanoseconds later, at t = 8 ns,

m = 34 conesponds to a frequency of 4.25 GHz, which conesponds to channel 4, and so

on. The frequency shown in the second column of Table 3 denotes the frequency of the harmonic signal that is ON or enabled (i.e., modulated and transmitted by the

transmitter).

Note that one may order the channels and their conesponding frequencies in a

variety of ways, as desired. In an embodiment that conesponds to Table 3, one may seek

to select the channel frequency conesponding to a time slot as far apart from neighboring

time slots possible. Referring to Table 3, note that a time period of at least two time slots

(i.e., 16 ns) separates adjacent channels. Selecting the channel and frequency and timing

plan in that manner tends to reduce or minimize interference among the channels, which

tends to increase or maximize channel separation and promote decay of multipath

interference.

Note, however, rather than using the channel plans in Table 3, one may use a wide

variety of other apparatus or systems that have other numbers of channels, frequencies,

time slots, and harmomc numbers, as desired, and as persons of ordinary skill in the art

who have the benefit of the description of the invention understand. Depending on the

desired system performance and design specifications, one may use channel and

frequency and timing plans to improve multipath rejection performance and to provide

channelization to accommodate multiple users in a communication system. With any given channel frequency and timing plan, one may use a variety of

modulation schemes, as desired, and as persons of ordinary skill in the art who have the

benefit of the description of the invention understand. Examples of modulation schemes include BPSK, QPSK, 8-QAM, and 16-QAM. The choice of channels and the type of

modulation technique used affects the overall data rate of the communication system or

apparatus.

Table 4 below shows an example of channels used and the approximate resulting data rates of throughput (in megabits per second) for various modulation techniques:

TABLE 4 Note that, rather than using six channels as Table 4 shows, one may use fewer or

more channels, as desired. The choice of the number of channels and the modulation

technique used depends on factors such as system performance and design specifications

and considerations, as persons skilled in the art with the benefit of the description of the

invention understand.

Furthermore, Table 4 conesponds to a UWB system with an approximately 500

MHz bandwidth per channel. One may apply the inventive concepts to a variety of UWB

systems with other bandwidths, as desired. The bandwidth of 500 MHz conesponds to

the smallest bandwidth defined as UWB in 47 C.F.R. Part 15 of the FCC rules and

regulations.

Note further that Table 4 conesponds to a system with a pulse repetition rate of

approximate 20.83 MHz. This pulse repetition rate conesponds to 8 nanosecond long

pulses (one cycle of the 125 MHz reference) sent every 48 nanoseconds at a 1/48 ns

(approximately 20.83 MHz) pulse repetition rate.

The last row in Table 4, labeled "Est. E^INo (dB)," denotes the estimated or

approximate energy used for each transmitted bit in the presence of noise. Referring to

Table 4, of the modulation schemes listed, that BPSK modulation has the lowest amount

of energy per bit to noise density ratio (7 dB for an approximate 0.1% bit enor rate), but also has the lowest overall data throughput. Conversely, 16-QAM has the highest energy

per bit to noise density ratio (16 dB for an approximately 0.1% bite enor rate), but has the

highest overall data throughput (roughly eight times higher than BPSK). Generally, the

more complex a modulation scheme, the higher the energy level it uses to transmit a bit

with a specified bit enor rate in the presence of noise.

One may use the information from Table 4 to design and implement

communication apparatus or systems that may meet the IEEE 802.15.3a proposed draft

standard. The proposed draft specifies data rates of about 110 megabits per second, about

200 megabits per second, and about 480 megabits per second. The cells highlighted with

bold numbers in Table 4 show combinations of modulation schemes and numbers of

channels that one may use to implement such apparatus or systems in a flexible manner.

Such flexibility is desirable because with the regulatory transmissions limits

specified as power spectral density limits, the total transmission power is proportional the

total bandwidth used (in other words on the number of channels used in Table 4). Thus,

one may transmit 125 Mb/s in three channels by using QPSK or, alternatively, in six

channels by using BPSK. With six channels the total radiated power may be twice that of

three channels for extended range communications. Hence, in a system, one may trade

bandwidth for range at a given or desired data rate. One aspect of apparatus or systems according to the invention concerns their

scalability. More specifically, one may design a plurality of 500 MHz-wide channels in

the frequency range of 3.1 to 5.2 GHz, using f_osc of 125 MHz, with the following center

frequencies:

A = Wfosc = 3-500 GHz, f₂ = 29f_osc = 3.625 GHz,

= 30/^ = 3.750 GHz,

and

₃ = 40/ _c = 5.000 GHz.

In a rulemaking, the FCC has limited UWB emissions to -41.3 dBm per MHz. For each

channel, one may determine the power from the following equation:

_c = -41.3 + 101og(2.374 /_∞ OS_ce)s '*

or -16.6 dBm per channel.

Thus, increasing the number of channels to provide higher data rates also

increases the total emitted power. For example, 2 channels would provide 3 dB more power than a single channel. As another example, 4 channels would provide 6 dB more

power than a single channel, and so on. Using multiple channels increases the total

emitted power by the same ratio as it increases the overall data rate or data throughput if

one uses a single modulation scheme (e.g., not switching from BPSK to QPSK, and so

on).

Consequently, the communication range remains approximately constant with an

increase in the data rate or data throughput or, as stated above, one may trade bandwidth

for communications range. Thus, apparatus or systems according to the invention

provide a desirable scalability feature such that increasing the data rate or data throughput

does not decrease the communication range. In other words, one may achieve

communication with a higher data throughput at a desired range by increasing the number

of channels and, hence, increasing the total emitted power.

Note that the examples described above with particular system parameters, such

as frequencies and frequency ranges, constitute illustrative embodiments of the inventive

concepts. As persons of ordinary skill in the art with the benefit of the description of the

invention understand, one may use a variety of other system parameters (e.g., frequencies

and frequency ranges), as desired, depending on various factors, such as desired design

and performance specifications. As an example, one may use two channels in the 3.1 to 5.1 GHz band, with

f_osc = 232 MHz (i.e., the channels are wider than 500 MHz), with the following center

frequencies:

/ = 16 f_osc = 3.712 GHz, and

= 20 ,^ = 4.640 GHz,

or m — 16 and 20, respectively, and where

/ -41.3+101og(2.374X ,

or -13.9 dBm.

By ruling, the FCC has allowed UWB emissions in the 3.1 to 10.6 GHz frequency

band or range. The exemplary channel plans described conform to the FCC rules while

allowing co-existing communications with the UNII band. Thus, the 3.1 to

approximately 5.2 GHz range constitutes an example of a desirable frequency range if

one wishes to avoid possible interference with communications in the UNII band.

The FCC rulemaking referenced above specifies one mask with a bandwidth

defined at -10 dB points. In the example given above, the -10 dB point occurs at 2 fosc ⁼ 464 MHz, while the -20 dB point occurs at 2.62 f_osc = 607.84 MHz. Thus, an

apparatus or system according to an illustrative embodiment based on this example meets

that FCC specification of -10 dB at 3.1 GHz. Note that, in this example, the two center

frequencies conespond to a bandwidth of 928 MHz, and that two channels fit in the

desired frequency range, here between 3.1 and 5.2 GHz.

Of course, one may implement other embodiments according to the invention

with a wide variety of parameters, such as frequericy plans, modulation schemes, and the

like, as persons skilled in the art with the benefit of the description of the invention

understand. In fact, one may use other frequency synthesis methods in which the value of

m does not constitute an integer, as noted above.

Another aspect of the invention relates to signal shaping in communication

apparatus. More specifically, signal shaping circuitry 2218 in FIG. 22 provides a way of

shaping, processing or filtering output signal 2202A of reference clock 41 to generate

shaped output signal 2204A. Shaped output signal 2204A feeds one input of mixer 2202,

as described above.

The signal shaping circuitry 2218 affects the spectrum of output signal of mixer

2204, which essentially constitutes the fransmitted signal of transmitter 2200. More specifically, rather than using signal shaping circuitry 2218 to mix shaped signal 2204 A

with data signals 2206, one may merely provide data signals 2206 to mixer 2204. As a

consequence of bypassing or not using signal shaping circuitry 2218, the spectrum of the

transmitted signal includes relatively high side lobe levels. Those side lobe levels may

fail to fit a desired mask, such as a mask that the FCC or another regulatory body has

prescribed.

By using signal shaping circuitry 2218, one may reduce or lower the side lobes of

the transmitted signal. Consequently, the spectrum of the transmitted signal tends to

more easily meet more stringent spectral radiation or mask requirements. Note that one

applies an analogous signal shaping function in a receiver that receives and processes

signals transmitted by transmitter 2200.

Referring to FIG. 23, receiver 2300 constitutes a matched template or matched

filter receiver. Baseband template generator 2312 provides the same or analogous signal

shaping functionality as does signal shaping circuitry 2218 in transmitter 2200. In other

words, as noted above, the output of baseband template generator 2312 matches the

output of signal shaping circuitry 2218 in transmitter 2200.

Note that signal shaping circuitry 2218 (and the conesponding signal shaping in

receiver 2300) may provide virtually any desired signal shaping, processing, or filtering function, as desired. By way of illustration, signal shaping circuitry 2218 may add a DC

component (such as a DC voltage), it may provide a magnitude function (e.g., by

performing full-wave rectification of the input signal), and the like, as persons skilled in

the art who have the benefit of the description of the invention understand.

Furthermore, one may combine various functions together, as desired. For

example, one may add a DC offset to a magnitude function. Generally, one may apply a

wide variety of signal shaping functions or combinations of functions by configuring the

transfer function of signal shaping circuitry 2218. The choice of the function(s) to use

depends on a variety of design and performance factors, such as desired spectral

characteristics and/or desired levels of out of band energy, and the like, as persons of

understand.

Rather than using analog circuitry to shape signals, one may use digital circuitry

or a mixed-mode circuitry, as desired. For example, one may store signal samples in a

memory, such as a read-only memory (ROM), and based on the input signal to signal

shaping circuitry 2218, use a counter to access various addresses in the memory in other

to generate a desired signal at the output of signal shaping circuitry 2218.

16 - By using an appropriate transfer function for signal shaping circuitry 2218, one

may smooth the spectrum of data signals 2206 or, put another way, reduce the high

frequency content of baseband data signals 2206. As noted above, data signals 2206

generally have pulse shapes (e.g., a square-wave or pulse train). Suppose, for example,

that signal shaping circuitry 2218 applies a magnitude function to output signal 2202 A of

reference clock 41.

Output signal 2204A of signal shaping circuitry 2218 constitutes a rectified cosine

signal, and its spectrum contains less high-frequency content than does the spectrum of

data signals 2206. Accordingly, output signal 2212 of mixer 2204 and, hence, the

transmitted signal, has side lobes with lower levels.

In the above example, note that one may implement the magmtude function

without using analog filtering components, as persons skilled in the art with the benefit of

the description of the invention understand. Thus, one may implement the magmtude

function in an IC that contains primarily digital circuitry, as desired. Doing so provides

more processing and manufacturing flexibility, which may result in higher reliability and

lower cost.

FIGS. 24 and 25 show sample waveforms for one example of a magmtude

function realized in an illustrative embodiment of a pulse shaping circuitry 2218 according to the invention. More specifically, FIG. 24 illustrates one cycle of output

signal 2208 of mixer 2202 in FIG. 22 (assuming rectangular data signals 2206). In other

words, reference clock 41 generates a cosine signal that it provides to signal shaping

circuitry 2218. Signal shaping circuitry 2218 processes that signal to generate its

magnitude, and provides the resulting signal (signal 2204A) to mixer 2202. Mixer 2202

mixes signal 2204A with input data signals 2206 to generate output signal 2208.

FIG. 25 depicts a Fourier transform of the signal in FIG. 24. Put another way,

FIGS. 24 and 25 provide time and frequency domain representations, respectively, of

output signal 2208 of mixer 2202. Thus, the waveform in FIG. 24 depicts the time

signal:

s(t) = cos(2πf₀ ),

and the spectrum in FIG. 25 illustrates the spectrum of s(t), or S(f):

In this exemplary embodiment, signal shaping circuitry 2218 realizes a magnitude

function. The magnitude of the cosine function (i.e., output signal 2204A of signal

shaping circuitry 2218), multiplied by the input data signals 2206, generates output signal

2208 of mixer 2202, as FIG. 24 illustrates (note, however, that FIG. 24 shows one cycle

of signal 2208). As noted above, FIG. 25 illustrates the Fourier transform of the signal in

FIG. 24. Effectively, in such an implementation, the input chip is weighted by a cosine

function.

Note that the maximum chip rate constitutes twice the frequency of the reference

clock, or 2f_osc. One may, however, send sparse chips at a rate of:

R = 2/, N

where O ≤ N ≤ [2f_gsc ^~] Furthermore,

^> _c=-41.3+101og(2.374/ ,

and the -10 dB point and the -20 dB point constitute, respectively, 2f_osc and 2.62f_osc.

One may write the closest frequency above 3.1 GHz (the edge of the FCC-prescribed

mask) where the signal level is -20 dB (or less),/, as: / = ( - 2.62) . /_OJC,

where m constitutes an integer.

Note that the signal in FIG. 24 and its associated spectrum in FIG. 25 constitute

baseband signals. In other words, the spectrum of the signal in FIG. 24 centers around

zero frequency, or DC. As the transmitter in FIG. 22 shows, one may heterodyne the

baseband signal so as to center it around a relatively high frequency (an RF frequency).

More specifically, one may use mixer 2204 to heterodyne output signal 2208 of mixer

2202 (by mixing it with signal 2210) and center it around a frequency mf_osc.

The heterodyning process shifts the spectrum of signal 2208 to a frequency band.

The frequency band may constitute a desired frequency band, such as a band prescribed

by a regulatory body (e.g., the FCC), or any other prescribed, specified, or designed

frequency band. Using the inventive concepts described here, one may design an RF

apparatus or system such that the shifted spectrum fits or satisfies a desired or prescribed

mask, for example, a mask specified in the FCC's rulemaking.

FIG. 26 illustrates sample waveforms in an exemplary embodiment of a

transmitter according to the invention, such as transmitter 2200 in FIG. 22. Waveform 2605 conesponds to output signal 2202 A of reference clock 41. Waveform 2610 denotes

shaped output signal 2204A, i.e., the output of signal shaping circuitry 2218. In this

particular embodiment, shaped output signal 2204A constitutes the magnitude of signal

2202 A. As noted above, however, one may configure signal shaping circuitry 2218 to

realize virtually any signal shaping or transfer function, as desired.

Waveform 2615 depicts input data signals 2206 (see FIG. 22). Waveform 2620

illustrates output signal 2208 as a function of time, i.e., the output signal of mixer 2202.

Note that, as described above, waveform 2620 conesponds to the product (by mixing) of

waveform 2610 and waveform 2615. As noted above, waveform 2620 conesponds to a

baseband signal (i.e., a signal centered around zero frequency, or DC).

Waveform 2625 denotes output signal 2210 of harmonic generator 2220. Note

that waveform 2625 conesponds to a particular value of m. As noted above, the value of

m varies as a function of time. Thus, the frequency of waveform 2625 also varies as a

function of time (in proportion with the value of m).

Waveform 2630 illustrates the output signal of mixer 2204, which constitutes

modulated RF signals 2212. Note that mixer 2204 mixes output signal 2208 (a baseband

signal) with output signal 2210 of harmomc generator 2220 (an RF signal) to generate

modulated RF signals 2212. As persons of ordinary skill in the art with the benefit of the description of the

invention understand, one may generate in-phase and quadrature orthogonal channels as

part of the heterodyning scheme. More specifically, by mixing output signal 2208, or a

pulse, with a cosine signal, one may generate an in-phase or I channel. Thus,

s, (t)=[cos(2πf_osct)] • cos(2πf₀t),

and

where

FIG. 27 illustrates an exemplary /-channel pulse (as mixed to generate a shifted or

heterodyned signal) as a function of time. FIG. 28 shows the magnitude of the spectrum of the signal in FIG. 27. Note that heterodyning has shifted the spectrum of the baseband

signal and has centered it around a relatively high frequency (approximately 4 GHz).

Conversely, mixing output signal 2208, or a pulse, with a sinusoid, one may

generate a quadrature or Q channel. Thus,

s_Q (t) = [c s(2πf_B )} ^■ sin(2πf ),

and

FIG. 29 illustrates an exemplary β-channel pulse (as mixed to generate a shifted

or heterodyned signal) as a function of time. FIG. 30 shows the magnitude of the

spectrum of the signal in FIG. 29. Note that heterodyning has shifted the spectrum and

has centered it around a relatively high frequency (approximately 4 GHz).

Note further that the formulae for S(f) for the /and Q channels, and the magnitude

of the spectra in FIGS. 28 and 30, are the same. As persons skilled in the art with the benefit of the description of the invention understand, the phase of the spectra are

different for the / and Q channels. (As noted above, however, FIGS. 28 and 30 depict the

magnitude of the respective spectra and therefore do not illustrate the phase differences.)

As noted above, by using signal shaping circuitry 2218, one may reduce the

magnitude of the side lobes present in the output spectra or profiles of transmitter 2200

(see FIG. 22). FIGS. 31 and 32 illustrate examples of how shaping the pulses affects the

side lobe magnitudes.

FIG. 31 shows two signals as a function of time that conespond to a cosine-

shaped pulse and a pulse with no shaping, in illustrative embodiments according to the

invention. Signal 3105 conesponds to a cosine-shaped pulse in output signal 2212 (see

FIG. 22) or, put differently, to a situation where one realizes a magnitude function by

using signal shaping circuitry 2218. Signal 3110, on the other hand, conesponds to a

situation where one does not apply any signal shaping to signal 2202A. In other words,

in the latter situation, signal 2204A constitutes a rectangular pulse or a DC level.

FIG. 31. Spectrum 3205 conesponds to the cosine-weighted pulse (shown as signal 3105

in FIG. 31). Spectrum 3210 conesponds to the situation where one does not apply any

signal shaping. Note that the side lobes of spectrum 3205 have a smaller magnitude than do the side lobes of spectrum 3210. Spectral mask 3215 denotes a desired or specified

mask, such as a mask prescribed by the FCC.

As noted above, one may apply virtually any signal shaping via signal shaping

circuitry 2218. FIGS. 33 and 34 provide additional examples of how shaping the pulses

affects the side lobe magnitudes.

FIG. 33 shows two signals as a function of time that conespond to a Gaussian-

invention. Signal 3105 conesponds to a Gaussian-shaped pulse in output signal 2212

(see FIG. 22) or, put differently, to a situation where signal shaping circuitry 2218

generates a Gaussian-shaped signal (or an approximation to a Gaussian-shaped signal), as

-0.5 (V described by the equation s(t) = e ^/τ at its output. Signal 3110, similar to FIG. 31,

conesponds to a situation where one does not apply any signal shaping to signal 2202A.

FIG. 34 illustrates the spectra resulting from using the signal shaping shown in

FIG. 33. Spectrum 3405 conesponds to the Gaussian-weighted pulse (shown as signal

3305 in FIG. 33). Spectrum 3410 conesponds to the situation where one does not apply

any signal shaping. Similar to FIG. 32, note that the side lobes of spectrum 3405 have a

smaller magmtude than do the side lobes of spectrum 3410. Spectral mask 3415, similar to spectral mask 3215 in FIG. 32, denotes a desired or specified mask, such as a mask

prescribed by the FCC.

Note that the choice of signal shaping function realized or applied by signal

shaping circuitry 2218 tends to not affect the characteristics of the main lobe in the

spectrum of the resulting output signal of the transmitter. In other words, although

certain signal shaping schemes (for example, the techniques described above) tends to

reduce the magnitude of the spectral side lobes, the main lobe characteristics tend to

remain relatively unaltered. As an example, note that in FIG. 32, the main lobe in

spectrum 3205 has a substantially similar shape and magnitude as does the main lobe in

spectrum 3210.

Depending on the signal shaping implemented or realized by signal shaping

circuitry 2218, the side lobes in the spectrum of the resulting signal at the output of mixer

2202 (i.e., signal 2212 in FIG. 22) may have too high a magmtude. In other words, the

side lobes in the spectrum of the resulting signal may exceed a limitation prescribed by a

particular mask. In such cases, one may filter signal 2212 before providing it to antenna

48. One may configure or design the transfer function (i.e., the filtering characteristics)

of the filter to remove or reduce energy at certain frequencies or within certain frequency

bands. Doing so reduces the side lobe magnitudes that would otherwise not fit within the

constraints of the particular mask. Apparatus and methods according to the invention are flexible and lend

themselves to a broad range of implementations, as persons of ordinary skill in the art

who have the benefit of the description of the invention understand. One may design,

implement, and manufacture communication apparatus and systems according to the

invention using a wide variety of semiconductor materials and technologies. For

example, one may use silicon, thin-film technology, silicon-on-insulator (SOI), silicon- germanium (SiGe), gallium-arsenide (GaAs), as desired.

Furthermore, one may implement such systems and apparatus using n-type metal

oxide semiconductor (NMOS), p-type metal oxide semiconductor (PMOS),

complementary metal oxide semiconductor (CMOS), bipolar junction transistors (B JTs),

a combination of B JTs and CMOS circuitry (BiCMOS), hetero-junction transistors, and

the like, as desired. The choice of semiconductor material, technology, and design

methodology depends on design and performance specifications for a particular

application, as persons of ordinary skill in the art who have the benefit of the description

of the invention understand.

Note that, by taking advantage of standard semiconductor devices and fabrication

technology, one may manufacture communication systems and apparatus according to the

invention with high yield, high reliability, and low cost. For example, one may manufacture such systems and apparatus using standard mixed-signal CMOS processes.

This flexibility allows manufacture and marketing of high data-rate consumer products,

professional products, health-care products, industrial products, scientific

instrumentation, military gear, and the like, that employ communication systems and

apparatus according to the invention.

Although the above description of communication systems and apparatus relates

to wireless communications, one may use the disclosed inventive concepts in other

contexts, as persons of ordinary skill in the art who have the benefit of the description of

the invention understand. For example, one may realize high data-rate land-line (i.e.,

using cables, fiber optics, house wiring, coaxial lines, twin-lead lines, telephone lines,

cable television lines, and the like) communication systems and apparatus, as desired.

Put another way, one may omit the antennas (and any associated circuitry) and

couple the transmitter and receiver together via a transmission line such as a wire line or

an optical fiber. In such systems, one obtains the same or similar benefits as the wireless

counterparts. More specifically, the UWB signal can coexist with other signals on the

same transmission medium.

The spectra shown in various figures (e.g., FIG. 13-16) are representative of

transmitted and emitted spectra. Radio wave propagation in free space exhibits no frequency dependency, so the field strength PSD at the receiver is the same as the

transmitted PSD, and the signal attenuates as l/^π r²). As noted above, if one receives

the signal with a constant-aperture type of antenna, then the received spectrum equals the

fransmitted spectrum. An example of a constant-aperture antenna is a wide-band horn or

a wide-band parabola whose gain increases as the square of frequency.

On the other hand, if one receives the signal with a constant-gain type of antenna,

then the received spectrum will have an imposed

characteristic. An example of a

suitable constant-gain antenna is a wide dipole whose gain is essentially flat with

frequency. Non-free-space environments may exhibit some frequency dependencies.

Those effects, however, are essentially equal whether one employs a constant-aperture or

a constant-gain antenna is employed.

Referring to the figures, the various blocks shown (for example, FIG. 3 or FIG. 5)

depict mainly the conceptual functions and signal flow. The actual circuit

implementation may or may not contain separately identifiable hardware for the various

functional blocks. For example, one may combine the functionality of various blocks

into one circuit block, as desired. Furthermore, one may realize the functionality of a

single block in several circuit blocks, as desired. The choice of circuit implementation

depends on various factors, such as particular design and performance specifications for a given implementation, as persons of ordinary skill in the art who have read the disclosure

of the invention will understand.

Other modifications and alternative embodiments of the invention in addition to

those described here will be apparent to persons of ordinary skill in the art who have the

benefit of the description of the invention. Accordingly, this description teaches those

skilled in the art the manner of carrying out the invention and are to be construed as

illustrative only.

The forms of the invention shown and described should be taken as the presently

prefened embodiments. Persons skilled in the art may make various changes in the

shape, size and anangement of parts without departing from the scope of the invention

described in this document. For example, persons skilled in the art may substitute

equivalent elements for the elements illustrated and described here. Moreover, persons

skilled in the art who have the benefit of this description of the invention may use certain

features of the invention independently. of the use of other features, without departing

from the scope of the invention.

Claims

I claim:

1. A radio-frequency (RF) transmitter, comprising:

a reference signal generator, the reference signal generator configured to provide a

reference signal having a frequency;

a signal generator, the signal generator configured to provide an operating signal

in response to a selection signal, wherein the operating signal has a

frequency equal to the frequency of the reference signal multiplied by a

number; and

a first mixer, the first mixer configured to mix the operating signal with a first

signal to generate a transmission signal.

2. The transmitter according to claim 1, wherein the selection signal varies as a

function of time.

3. The transmitter according to claim 2, further comprising a second mixer, the

second mixer configured to mix an input data signal with a second signal to generate the

first signal.

4. The transmitter according to claim 3, further comprising a signal shaping

circuitry, the signal shaping circuitry configured to generate the second signal by shaping

the reference signal.

5. The transmitter according to claim 4, wherein the input data signal comprises a

modulated data signal.

6. The transmitter according to claim 5, wherein the input data signal comprises a

binary phase shift keyed (BPSK) signal.

7. The transmitter according to claim 5, wherein the input data signal comprises a

quadrature amplitude modulated (QAM) signal.

8. The transmitter according to claim 5, wherein the input data signal comprises a

quadrature phase shift keyed (QPSK) signal.

9. The transmitter according to claim 4, wherein the signal shaping circuitry is

configured to full-wave rectify the reference signal to generate the second signal.

10. The transmitter according to claim 9, wherein the reference signal comprises a

sinewave signal.

11. The transmitter according to claim 4, wherein the signal shaping circuitry is

configured to generate a Gaussian-shaped signal as the second signal.

12. The transmitter according to claim 4, wherein the signal shaping circuitry

comprises a filter.

13. The transmitter according to claim 2, wherein the selection signal varies according

to a channel frequency plan.

14. The transmitter according to claim 13, wherein the selection signal further varies

according to a channel timing plan.

15. The transmitter according to claim 1, further comprising a filter, the filter

configured to receive the transmission signal, the filter further configured to generate a

filtered transmission signal.

16. The transmitter according to claim 1 , wherein the frequency of the operating

signal equals the frequency of the reference signal multiplied by an integer number.

17. The transmitter according to claim 1, wherein the frequency of the operating

signal equals the frequency of the reference signal multiplied by a non-integer number.

18. A receiver, comprising: a first mixer, the first mixer configured to mix an input radio-frequency (RF)

signal with a first signal to generate a first mixed signal;

a integrator/sampler, the integrator/sampler configured to receive the first mixed

signal and to provide an output signal;

in response to a selection signal, wherein the operating signal has a

frequency equal to the frequency of a reference signal, multiplied by a

number; and

a second mixer, the second mixer configured to mix the operating signal with a

template signal to generate the first signal.

19. The receiver according to claim 18, wherein the selection signal varies as a

function of time.

20. The receiver according to claim 19, further comprising a reference signal

generator, the reference signal generator configured to generate the reference signal.

21. The receiver according to claim 20, further comprising a template signal

generator, the template signal generator configured to provide the template signal.

22. The receiver according to claim 21, wherein the template signal generator is

further configured to shape the reference signal to generate the template signal.

23. The receiver according to claim 19, wherein the output signal of the

integrator/sampler comprises a signal derived from the first mixed signal.

24. The receiver according to claim 23, further comprising a confroller, the controller

configured to generate a datum value from the output signal of the integrator/sampler.

25. The receiver according to claim 24, wherein the input radio-frequency (RF) signal

comprises a binary phase shift keyed (BPSK) signal.

26. The receiver according to claim 24, wherein the input radio-frequency (RF) signal

comprises a quadrature amplitude modulated (QAM) signal.

27. The receiver according to claim 24, wherein the input radio-frequency (RF) signal

comprises a quadrature phase shift keyed (QPSK) signal.

28. The receiver according to claim 22, wherein the template signal generator is

further configured to shape the reference signal by full- wave rectifying the reference

signal.

29. The receiver according to claim 22, wherein the template signal generator is

further configured to shape the reference signal by generating a Gaussian-shaped signal

from the reference signal.

30. The receiver according to claim 22, wherein the template signal generator is

further configured to shape the reference signal by filtering the reference signal.

31. The receiver according to claim 24, wherein the reference signal generator

comprises a phase locked loop (PLL).

32. The receiver according to claim 31, wherein the controller is further configured to

provide a control signal to the phase locked loop.

33. The receiver according to claim 24, wherein the controller is further configured to

provide the selection signal.

34. The receiver according to claim 19, wherein the selection signal varies according

to a channel frequency plan.

35. The receiver according to claim 34, wherein the selection signal further varies

according to a channel timing plan.

36. The receiver according to claim 18, wherein the frequency of the operating signal equals the frequency of the reference signal multiplied by an integer number.

37. The receiver according to claim 18, wherein the frequency of the operating signal

equals the frequency of the reference signal multiplied by a non-integer number.