ULTRA-WIDEBAND COMMUNICATION SYSTEMS AND METHODS TECHNICAL FIELD OFTHE INVENTION
The invention relates generally to ultra-wideband communications, and more particularly to systems and methods for communication using ulira-vvideband technology.
BACKGROUM)OFTHEINVEMION
The electromagnetic spectrum used to convey radio communications is a precious commodity. Communication systems seek to use this spectrum as efficiently as possible to maximize the capacity or quantity of information, which canbe conveyed using the spectrum
Various multiple access techniques have been developed to transfer information among a number of users, all while efficiently using spectrum. Time division multiple access (TDMA) techniques assign different users to different timesbts. Capacity is hardlimited by the number of time slots available. To prevent intolerable interference, the portion of the spectrum used in one radio coverage area or cell has conventionally been unusable in adjacent ceϋs. Thus, only a fraction, typically less than αie-third, of the entire spectrum available for conveying communications has been conventionally usable in any one location. Ih other words, conventional TDMA systems employ a fiequency reuse pattern of at least three, indicating an inefficient use of spectrum
Conventional direct sequence spread spectrum (DSSS) code division multiple access (CDMA) techniques theoretically use the spectrum more efficiently than TDMA techniques. However, in practice conventional DSSS- CDMA techniques typically fill to provide results significantly better ton TDMA. DSSS-CDMA techniques assign different users to different codes. The different codes have conventionally been selected because of orthogonality or low cross correlation properties wiflitiie codes of other users. These properties rnirώnize interference. AU communications are broadcast using ύie same spectrum, so the fiequency reuse pattern equals one. While the commonly used spectrum conveys a composite of αmmunications for all users, each individual user's communications are extracted fiorn the ∞rrφcstebycorolatingareceived signal against the individual user's assigned code.
Capacity in conventional DSSS-CDMA systems is interference limited. Ih other words, more andmore codes can be assigned so that the given amount of spectrum can service more and more users until interference reaches a level where oitiy arninimally acceptable quality of service results. Inpractice,most conventionalDSSS-CDMA systems can assign far fewer codes ton appear theoretically possible due to a near-far effect and multipafh. The near-far effect results when signals from different users are received with greatly differing field strengths, but this detrimental effect may be ameliorated somewhat bypower control.
Multipath results when the transmitted signal takes mulφle pate totherecmerduetobekig reflected from and deflected around obstacles in the environment As the signal propagates over the multiple paths, different propagation delays are experienced. Thus, a signal transmitted at aprecise instant in time is rawed spread ovσ an interval, causing the signal to interfere with itself In conventional DSSS-CDMA communication systems, multipath tends to destroy the orthogonality of spreading codes, resulting in dtanatic^y increased interferaice.
SUMMARY OFTBDE INVENTION
Inordαto combat the above problems, systems and methods described herein provide anovel ultra-wideband communication system. In one embodiment, an ultra-wideband communication system divides a stream of data conveying symbols into a plurality of unspread substieams. A common spreading code is generated at the ultra- wideband transmitter, and each of the unspread suhstreams are spread using the common spreading code to form a plurality of spread substreams. The spread sulsstreams are ambine^
In another embodiment, an ultra-wideband communication system comprises a demultiplexer tor dividing a stream of data conveying symbols into a plurality of unspread substreams. A spreading section is coupled to the demultiplexer and configured to generate a plurality of spread substreams fiom the plurality of unspread substreams. A cαiabining section is coupled to the spreading section and configured to form a composite signal fiom the plurality of spread substreams, and a transmission section is coupled to flie combining section and configured to transmit the composite signal over an ultra-wideband communication channel.
These and other features and advantages of the present invention will be appreciated fiom review of the following Detailed Description of the Preferred Embodiments, along with the accompanying figures in which like reference numerals areusedto describe the same, similar or correspondingparts in the several views of the drawings.
BREEFDESCRIPπθNOFTHEDR4WINGS
A more complete understanding of the present invention may be derived by referring to the detailed description and claims when considered in connection wifti the Figures, wherein like reference numbers refer to similar items tiiroughoutihe Figures, and:
HG. 1 shomakyoutdiagtHnofanexerrφlatyen
FIG.2 shows a timing diagram, which depicts a temporal format of a TDMA communication signal;
FIG.3 shows ablock diagram ofatransnitterandareedverconfigiiredh present invention;
HG. 4 shows a timing diagram depicting how a cyclic spreading code is applied to blocks of unspread data streams in accordance with first, second and third embodiments of a DSSS modulation section in the transmitter of the present invention;
FIG.5 shows ablock diagram of the first embodiment ofthe DSSS modulation section;
FIG.6 shows ablock diagram oflhe second embodiment of the DSSS modulation section;
FIG.7 shows ablock diagram offhe third embodiment of tiie DSSS modulation section;
FIG.8showsafirstembc)c%ientofaCDMtoTDM converter section in Iherecdverofflie present invention;
FIG.9 shows an exemplary spectral analysis of a suitable spreading code usable in connection with the present invention, flie spectral analysis showing a substantially flat response;
FIG. 10 shows an exemplary timing diagram of various individual signal components present in a composite signal output firm amatohedfilter portion of amismak^ed filter kflie CDM to TDM converter,
FTG. 11 shows a timing diagram depicting how a cyclic spreading code is applied to blocks of υnspread data steams in fourth and fifth embodiments of lhe DSSS modulation section;
FlG. 12 shows ablock diagram of the fourih and fiffli embodiments of tie DSSS modulation section;
HG. 13 shows a second embodiment of the CDM to TDM converter for use with the fourth embodiment of the DSSS modulation section;
HG. 14 shows a fhiiri embodiment of flie CDM to TDM converter for use wilh lhe fifth embodiment of the DSSS modulation section;
FIG. 15isanillustt^onofdiffej^ntcoriimunicationme11iods;and
FIG. lόisanfflustrationoftwoultra-widebandpulses.
B will be recognized that some or all of the Figures are schematic representations for purposes of illustration and do not necessarily depict the actual relative sizes or locations of the elements shown. The Figures are provided for the purpose of illustrating one or more embodiments of the invention with the explicit understanding that they will not be used to limit tfie scope or the meaning ofthe claims.
DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS
In the following paragraphs, the present invention will be described in detail by way of example with reference to the attached drawings. While this invention is capable of enibodiment in many different forms, there is shown in the drawings aid will herein be described in detail specific ernbodiments, with lhe understood to be considered as an example of the principles of the invention and not intended to limit the invention to the specific embodiments shown and described. That is, throughout fliis description, the embodiments and examples shown should be considered as exemplars, rather than as limitations on the present invention. As used herein, the "present invention" refers to any one of the embodiments of the invention described herein, and any equivalents. Furthermore, reference to various featiirφ) of the "present invention" throughout this document does not mean lhat all claimed embodiments or methods must include flie referenced features).
The present invention provides several advantages and features, for example, the present invention combines TDMA and spread spectrum techniques so fhat wireless αammunications capacity is increased over the capacities achievable through conventional TDMA and/or CDMA systems using an equivalent amount of spectrum.
Another advantage of the present invention is that robust, simple, and inexpensive processing techniques are usable, making the present invention suitable for hubs, subscriber units, mobile stations/ fixed stations, portable stations, and the lice.
Another advantage is that the present invention may be adapted to and used in conjunction with a variety of modulation and multiple access techniques, such as frequency division multiple access (FDMA) and orthogonal frequency divisionmultrplexing (OFDM).
Another advantage of the present invention is 1hat a composite RF communication signal includes signal components obtained by modulating diverse branches of a single user's data stream using cyclic variants of a common
spreading code.
Another advantage is that the present invention is configured to tolerate self-interference and is better able to tolerate multipathtlian conventional DSSS<13^^
These and other features and advantages of the present invention will be appreciated fiom review of the following discussion:
FIG. 1 shows a layout diagram of an exemplary environment in which a communication system 20 configured in accordance with the teaching of the present invention may be practiced Ccrømunication system 20 includes any number of transmitters (TXs) 22 (three shown) and any number of receivers (EOCs) 24(five shown). Transmitters 22 wirelessly broadcast messages through RF time domain multiple access (IDMA) communication signals 26 which are receivable by receivers 24 located within radio coverage areas 28 for the transmitters 22. Radio coverage areas 28 may also be called cells or sectors. As illustrated in FIG. 1, various ones of radio coverage areas 28 may be adjacent to one another aid even overlap to some extent Ih the preferred embodiment, a common spectrum is used in all radio coverage areas 28 so that communication system 20 has a frequency reuse pattern substantially equal to one.
For the sake of clarity, FIG. 1 depicts only a forward link in which radio equipment is viewed as being only a transmitter 22 or a receiver 24. However, those skilled in the art will appreciate that a reverse link may also be implemented and that each item of equipment may have both a transmitter and receiver. The reverse link may use the same or a different spectrum fiom the forward link. If a forward link confoims to the teaching of the present invention, then the reverse linkmayormaynot conform, and vice versa.
FIG. 2 shows a timing diagram, which depicts an exemplary temporal format for TDMA communication signal 26. FIG.2 specifically depicts two frames 30, each ofwMchis temporally subdividedinto any number of timeslots 32. Diferent timeslots 32 are preferably assigned to dirler^ so that different recipients are distinguished from one another by being assigned to flie different time slots 32. Ih the prefened embodiments, TDMA communication signal 26 consumes the entire common spectrum for each time slot 32. Nothing requires atirne slot 32 tobe assigned to receivers 24 for an indefinite period or to be ofthe same duration as other time slots 32.
Each time slot 32 of TDMA communication signal 26 is subdivided into successive blocks 34 of symbols 36. HG.2 labels blocks 34 with the identifiers Bjς ibrk=OtoK-l, whereKis anintegernumber. Any number ofblocks 34 maybe mcludedmeachtmeslot 32. EachblockBkincludesMsyrnbolsSόjlabeledasakjmfor m=OtoM-l,whereMis an integer number. HG.2 illustrates each of symbols 36 within ablock 34 as being conαnientiy present throughout the entire duration of ablockperiodbecause αrtainpreferred ernbodiments discussed below configure symbols 36 to remain present forblockperioda
FIG.2 further illustrates that the M symbols 36 of each block 34 are spread using an N<;hip spreading code 38, labeled as Cn, for n= 0 to N-I, whereN is an integer number. As discussed in more detail below, each symbol 36 is independently spread using cyclic variations ofthe same common code 38. The number M of symbols 36 in a block
may equal the number N of chips in a spreading code, in which case tie spreading factor equals one. However, performance irnprovements result whenNis greaterflianM.
HG.3 shows ablock diagram of a single transmitter 22 and a single receiver 24 configured in accordance with the teaching of Ihe present invention. Those skilled in Ihe art will appreciate that all transmitters 22 and receivers 24 may be configured similarly. IQ addition, any number of receivers 24 may, at any given instant, receive TDMA communication signal 26 fiom a given transmitter 22 and, in feet, may receive TDMA communication signals 26 from more than one transmitter 22.
Transmitter 22 includes a TDMA modulation section 40, which generates a TDMA-configured stream 42 of data conveying symbols 36. Stream 42 feeds a direct sequence spread spectrum (DSSS) modulation section 44, which generates a composite signal 46. Composite signal 46 feeds a Iraπsmission section 48, which forms TDMA communication signal 26 from composite signal 46 and wirelessly broadcasts TDMA communication signal 26 for reception by receivers 24 located withinradio coverage area 28 (FIG. 1) oftansmitter22.
Within TDMA modulation section 40 any number of data sources 50 supply digital data to a multiplexer (MUX) 52. The digital data fiom data sources 50 may be intended for any number of receivers 24. Multiplexer 52 groups the digital data so that data intended for different receivers 24 are serially fed to a cyclic redundancy check (CRQ section 54 in accordance with the assignment of timeslots 32 (FIG.2) to receivers 24. CRC section 54 provides forward error correction inarnarmer well understood bythose skilled in Ihe art
From CRC section 54, Ihe input data stream may be fed through a scramble 56 which lmtorrώesihe data to an encode and interleave section 58. Section 58 may apply another type of error correction, such as convolutional or turbo encoding, to tie input stream, and interleave the data CRC section 54 and section 58 may utilize a fomi ofblock encoding. The block size or boundaries of such encoding need have no relationship to blocks 34 (FIG. 1), discussed above.
However, the output of section 58 feeds an optional peak-to-average (P/A) block encoding section 60. P/A block encoding section 60 applies a type of encoding which primarily reduces the peak-to-average power ratio in composite signal 46 and thereby lessens the demands placed on apower amplifier included in transmission section 48 to faithfully reproduce communication signal 26 with a minimum amount of distortion. This type of encoding may, but is not required to, provide additional coding gain. In the preferred embodiments, when P/A block encoding section 60 is included, it applies block encoding so that encoded blocks coincide with successive blocks of symbols 36 (FIG. 2), discussed above. IQ other words, the data are encoded so thatP/A encodedblocks begin with symbol 36 a® (FIG.2) and end with symbol36 a^fFIG.2).
P/A block encoding section 60 feeds a constellation encoding section 62 which converts the data into complex symbols in accordance with apredeterminedphase constellation As an example, each four-bit group of data output ftom P/A block encoding section 60 may be mapped by section 62 into a single complex symbol having in-phase and quadrature components in accordance with a 16-QAM phase constellation. However, those skilled in the art will
appreciate that 1he present inverώco maybe us^
The stream of complex symbols output fc>m constellation encode section 62 passes Ihtough a syncluonization multiplexer (SYNC MUX) 64, where a preamble 66 is insedM into tie steam at appitpi^ intervals. Preamble66isa known code which helps receivers 24 obtain synchronization and determine the timing of flames 30 and time slots 32 (FIG.2). The resulting TDMA-configured complex stream 42 serves as the output from TDMA modulation section 40 and feeds DSSS modulation section 44.
Within DSSS modulation section 44, a demultiplexer (DEMJX) 68 divides TDMA-configured stream 42 of complex symbols 36 into blocks 34 (FIG.2) of symbols 36. As a result, M unspread complex symbol substreams 70 are provided by demultiplexer 68 so 1hat each unspread substream 70 contributes a single complex symbol 36 during each block 34, and each block34 has ablockpeήcdT*M, whereT is the symbol periodofTDMA-configuredstream42.
Unspread substreams 70 feed a spreading section 72. Within spreading section 72, cyclic variations of common spreading code 38 (FIG.2) are applied to the M unspread substreams 70 to form M spread substreams 74 of "chips." The chip period in each spread substream 74 is T*M/N. The M spread substreams 74 may be passed through an optional peak-to-average (P/A) reduction section 76 which adjusts phase angles of the complex chips conveyed in the spread substreams 74 in amanner understood by those skilled in lhe art toreducspeds-to-aveiage power ratio and lessen demands placed on a power amplifier. Following P/A reduction section 76, a combining section 78 combines spread substreams 74 to form composite signal 46. Various αnbodirnents of DSSS modulation section 44 are discussed in more detail below.
Transmission section 48 includes any number of components and functions well known to those skilled in the art For example, scrambling section 56 and/or synchronization multiplexer 64, discussed above, may be included in ttansmission section 48 rather ten in TDMA modulation section 40. A pulse shaping section (not shown) is desirably included in transmission section 48 to spread the energy from each chip over a number of chip intervals using a suitable filter which minimizes inter-symbol or inter-chip interference so that spectral constraints maybe observed. Transmission section 48 may also include digital-to-analog conversion, quadraturemodulation, up-conveision, andpower amplification functions, all implemented in conventional fashion Power cαiliolmaybeinplementediniransrnissioii section 48 atthe power amplifier to ameliorate a potential near-far problem, which should be much less pronounced in communication system 20 (FIG. 1) ten in traditional CDMA communication systems. After pulse shaping, analog conversion, up- conversion, and amplification, TDMA communication signal 26 is formed from composite signal 46 and wirelessly broadcast from transmission section 48.
Receiver 24 receives TDMA communication signal 26. Within receiver 24, communication signal 26 is processed through a receiving section 80 and passed to a code division multiplex (CDM) to time division multiplex (TDM) converter 82. CDM to TDM converter 82 produces a baseband signal 84, which is further demodulated in a TDM demodulation section 86, with individual users receiving their respective data streams 88. Qf course, nothing requires a receiver 24 to serve multiple users and TDM demodulation section 86 may simply provide a data stream
intended for a single user.
Receiving section 80 includes any number of components and functions well known to those skilled in the art. For example, amplifying, filtering, and down-conversion may be performed to form an intermediate frequency (IF) signal The F signal maybe converted from an analog form into a digital form, and automatic gain control (AGQ may be provided. In the prefened embodiments, the digitized form of the down-converted communication signal 26 passes to CDMto TDM converter 82.
Generally, CDM to TDM converter 82 performs despreading and optionally performs equalization on Hie communication signal Various embodiments of CDM to TDM converter 82 are discussed in more detail below.
TDM demodulation section 86 includes any number of components and functions well known to those skilled inflieart For example, cliannel estimation and synchronization may be performed in TDM demodulation section 82. A rake receiver and/or equalizer may be included. De-interleaving, error correction decoding, and descrambϋng are desirablyperibmied, and preambles and other control data ate evaluated to detect time slots assigned to the receiver 24. These and other components and functions conventionally used in digital demodulators may be included in TDM demodulation section 86.
FIG. 4 shows a timing diagram depicting how common spreading code 38 (FIG. 2) is applied to blocks 34 (FIG. 2) of unspread data substreanis 70 (FIG. 3) in accordance with first, second and third embodiments of DSSS moduMonsec£cm44(πG.3)ibitransrnittα22(FIG.3). RG.4ispiesmtedhtabularfomi,wi1hrowsrepresenting1he application of the chips of common spreading code 38 to symbols 36 (FIG. 2). Columns in FIG.4 depict successive blocks 34. As indicated by a shaded region in FIG.4, spreading code 38 is applied to unspread substreams 70 so that composite signal 46 is influenced, at least for a portion of Ihe time, by symbols 36 from two different blocks 34. In particular, in flie specific embodiment depicted by FIG.4, for only a single chip of each block period is composite signal 46 influenced by symbols 36 from a common block 34 of symbols. The manner of application of common spreading code 38 (FIG. 2) to blocks 34 (FIG. 2) of unspread data substteams 70 depicted in FIG. 4 may be contrasted with an alternate embodiment, discussedbelow in connection withFIG. 11.
FIG. 5 shows a block diagram of the first embodiment of DSSS modulation section 44. Demultiplexer 68 is omitted fiomFIG.5 for convenience. The unspread subsiream 70 conveying symbols a^, experiences no delay before being fed to a first input of amultipliα 90. However, flie unspread substreams 70 conveying symbols a^ through a^ are respectively delayed in delay elements 92 by 1 Ihrough M-I symbol periods (T) before being fed to respective first inputs of othermultipliers 90.
A spreading code generation section 94 generates cyclic variations of common spreading code 38. FIG. 5 illustrates code generation section 94 in matrix form, which matrix takes on a cyclic Toεplitz form because flie matrix elements hold cyclic variations of flie same spreading code 38. As depicted in HG. 5, different columns of flie matrix supply code chips C0 through Q^1 to second inputs of respective multipliers 90. Different rows of the matrix indicate different code chips to apply during different chip intervals. So long as flie number (N) of chips in spreading code 38 is
greater ten or equal to the number (M) of symbols 36 per block 34, diflferait code chips offlie same cxxfe are qpKed to difleient symbols during any and all chip intervals.
Outputs of multipliers 90 provide respective spread substreams 74. FIG. 5 omits depiction of optional P/A reduction section 76 (FIG.3) for convenience. Combining section 78 takes the form of an adder, so 1hat composite signal 4 6 during each chip interval equals the sum of M symbols 36, with each of the M symbols being premultiplied by designated chips of common spreading code 38. Accordingly, DSSS modulating section 44 temporally onsets application of common spreading code 38 to unspread substreams 70 so that the resulting spread substreams 74 correspond to unspread substreams 70 modulatedby cyclic variations of common spreading code 38.
FIG. 6 shows a block diagram of the second embodiment of DSSS modulation section 44. This second embodiment is equivalent to the first embodiment ofFIG.5, but it is implemented differently. Demultiplexer 68 (FIG.3) is omitted fiom FIG. 6 for convenience. Ih this embodiment, spreading code generation section 94 need not be implemented as a two-dimensional matrix having a different row to define the different chips to be applied during different chip intervals, as discussed above in connection with FIG.5. Rather, spreading code generation section 94 may be implemented as a one-dimensional matrix having different columns, aid only one of those columns is simultaneously appftedtoclifferaitiinspread substreams 70. Spreading code generation section 94 maybe implemented as a shiflregister (Orώgiπ^toshiftcyclicallyatthechiprate. ihordertoacMeveiheappropriaietemrx^ now positioned between multipliers 90 and the adder of combining section 78. Accordingly, in this second embodiment of DSSS modulating section 44, DSSS modulating section 44 temporally offsets the application of common spreading code 38 to unspread substreams 70 so that the resulting spread substreams 74 correspond to unspread substreams 70 modulatedby cyclic variations of common spreading code 38.
FIG.7 shows ablock diagram of the IhM embodiment ofDSSS modulating section 44. This third embodiment is also equivalent to the first embodiment of FIG. 5, but is implemented differently. This third embodiment is a finite impulse response (HR) implementation. In this third embodiment, symbol stream 42 (FIG.3) is led to a series of delay elements 96, each of which imparts a one-chip interval delay. The series of delay elements 96 serves the role of demultiplexer 68 (FIG.3) in teflαird embodiment, with theiφuttothe first deky element 96 and the outouts of all delay elements 96 providing unspread substreams 70. Delay elements 92 (FIGs.5-6) fiom the first and second embodiments of spreading section 72 are omitted.
Spreading code generation section 94 simply provides common spreading code 38, and need not be cycled beca^eiinspiead substreams 70 to wMch spreading αx^ requirements. Accordingly, symbol delay elements 92 are omitted and spreading code generating section 94 need not cycle the common spreading code or explicitly provide separate versions of spreading code 3 8 to separate unspread substreams 70. Nevertheless, in this third embodiment ofDSSS modulation section 44, spreading section 72 temporally offsets application of common spreading code 38 by sequentially delaying symbols 36 to form unspread substreams 70 and applying spreading code 38 to the delayed symbols in unspread substearris 70 so triat the resulting spr^
74 correspond to unspread substreams 70 modulated by cycHc variations ofcommon spreading code 38.
FIG. 8 shows a fist embodiment of CDM to TDM converter 82 included in receiver 24 (FIGs. 1 and 3). Desirably, CDM to TDM converter 82 is configured to complement DSSS modulation section 44 of transmitter 22 (FIGs. 1 and 3). fii particular, this first embodiment of CDM to TDM converter 82 is configured to complement any of the first through ftM embodiments ofDSSS moduMcm section 44 dscussed above in αmnecticnwiih H
CDM to TDM converter 82 includes a pulse shaping matched filter 98, the output of which feeds a mismatched filter 100. Pulse shaping matched filter 98 complements a pulse shaping filter (not shown) desirably implemented in transmission section 48 of transmitter 22 (FIG. 3) to optimize signal-to-noise ratio and band-limit the signal. Pulse sriaping matched filter 98 is desirably implemented using conventional techniques knowntolhosesldlledhtheaΛ
Mismatched filter 100 accomplishes two functions. One function is despreading and the oihσ function is sidelobe suppression. Ia tact, mismatched filter 100 is desirably implemented to correspond to a spreader matched fitter 102 upstream of a sidelobe suppression filter 104. One technique for implementing mismatched filter 100 is simply to implement two filters coupled in series for the despreading and sidelobe suppression Junctions. Ih aiother technique (not shown) the two functions maybe αmbmedinaαmmon filter.
Mismatched filter 100 experiences a signal-to-noise ratio typically worse than thatofamatchedfilter. However, h thepreferred embodiments, mismatched filter 100 is desirably configured to achieve arelative efficiency of greaterfhan 60%, andmore preferably greater than 90%, compared to amatehed filter.
Those skilled in the art will appreciate that the configuration of common spreading code 38 is a strong determinant of the relative efficiency of mismatched filter 100. For example, conventional orthogonal pseudonoise (PN) codes commonly used in conventional CDMA applications are unacceptable because their mismatched filters achieve relative efficiencies roughly around only 50%.
Whileawide variety of difiereri codes may be used with the present inver£on,cc<ies which have low aperiodic autocorrelation sidelobes and a substantially flat spectral analysis are preferred in this embodiment Barker codes make suitable codes because of aperiodic autocorrelation sidelobes having magnitudes less than or equal to one. However, for many applications the limited length (i.e., N <13) and/or prime numbered length of many Barker codes proves a detriment In such cases, other codes having a greater length and slightly greater aperiodic autocorrelation sidelobes, such as magnitudes less than or equal to two or three are aixφtableandrrmybeeasflydenVedbythc^sMedinflieart
FIG. 9 shows an exemplary spectral analysis of a suitable spreading code usable in connection with the present invention, In particular, FIG.9 represents an arbitrary code for which a spectral analysis can be performed using a time- frequency domain transformation, such as a Fourier transform. While a code having a precisely flat spectral analysis result is not a requirement, better results are achieved whenno frequency bin shows substantially more or less signal level 1hanoiherbins,asdepictedinFIG.9. As an example* Ihesignallevelmeachbmiscles^ signallevel taken overall bins, ^particular, forbestresultsno bins should exhibit anearly zero signal level
The implementation of mismatched filter 100 illustrated in FIG. 8 will be readily understood by those skilled in
the art. Spreader matched filter 102 may be implemented using the complex conjugate of spreading code 38 (FIG. 2) presented in a reverse order. Sidelobe suppression filter 104 may be implemented using well-known EFT or linear prograrrrming techniques.
The output of spreader marched fitter 102 in mismatched filter 100 is a composite signal 106 equivalent to the autocorrelation tunction applied to each of the M unspread and spread substreams 70 and 74 (FIG.3 and 5-7) discussed above.
EIG. 10 shows an exemplary timing diagram of the various individual signal components present in composite signal 106 output fiom the matched filter 102 portion of rnisrnatched filter 100. For convenience, FIG. 10 depicts an exemplary situation where the number of substreams 70 and 74 (Le., M) equals seven and the number (Le., N) of chips in spreading code 38 equals seven. Thικ,eachrowinFIG.10 represents one oftheseven substreams, and eachtow depicts autocoπelation with an assumed rectangular pulse. Qf course, composite signal 106 is the sum of all rows in HG. 10 rather than seven distinct signals.
Assuming ideal synchronization where samples are taken at the integral chip intervals of 0, 1, 2, ..., then in this example, seven successive samples yield the signal levels of the seven substreams. However, each sample in composite signal 106 is corrupted by self-mterference 108, caused by sidelobes of the autocorrelation function. Accordingly, sidelobe suppression filter 104 (FIG. 8) substantially attenuates the self-interference 108 of the sidelobes while not severely attenuating the autocorrelationpeak.
Referring back to FIGs.3 and 8, the output sidelobe suppression filter 104 provides baseband signal 84, which also serves as the output of CDM to TDM converter 82. Baseband signal 84 is routed to TDM demodulation section 86. Depending upon the severity ofmulφathiernaininginbaseband signal 84 afterprocessing through sidelobe suppression filter 104, a rake receiver (not shown) or equalizer (not shown) may be used in TDM demodulation section 86 to compensate for the multipath. While some inefficiency may result firm using a rnisrnatched filter to despread communication signal 26, any such inefficiency is more than compensated for by a marked improvement in multipath tolerance.
While receiver 24 receives a communication signal 26 fiom one transmitter 22, it may simultaneously receive other ∞mmimtcation signals 26 fiom other transmitters 22 in adjacent radio coverage areas 28 (FIG. 1). Conventional CDMA techniques maybe used to prevent interference between such diverse rorcimurication signals 26. For example, different spreading codes 38 maybe selected for use at different transmitters 22. Such different spreading codes 38 are desirably configured to have low cross-coireMon sidelobes among all spreading codes 38. If this option is selected, only
communication signals 26 from non-adjacent radio coverage areas 28 (FIG. 1). Alternatively, transmitter 22 and receiver 24 may include other stages to further scrarnble/descramble spread spectrum signals using other spreading codes.
The embodiments of the present invention discussed above and characterized by the timing depicted in FIG 4, wherein composite signal 46 is influenced, at least for aportion of 1hetime,bysyrnbols36fiomtwodifrerentblocks34,
show advantageous resilience in Hie presence of multipath. However, alternate embodiments, discussed below, may provide evenbetterperformance in Ihepiesenceofmultrpathfersαiie ^plications.
HG. 11 shows a timing diagram depicting how common spreading code 38 (FTG.2) is applied to blocks 34 (FIG.2) of unspread data substrεams 70 (FIG.3) in accordance with fourth and fifth embodiments ofDSSS modulation section 44 (FIG.3) in transmitter 22 (FIG. 3). like HG.4 discussed above, HG. 11 is presented in tabular form, with rows representing the application of the chips of common spreading code 38 to symbols 36 (HG.2). Columns in HG. 11 depict successive blocks 34. As indicated by a shaded region in HG. 11, spreading code 38 in the fourth and fifth embodiments is applied to unspread substreams 70 so that ccmpcdte signal 46 is influence^ at aU times, by syrr^ from only commonblocks 34.
HG. 12 shows a block diagram of the fourth and fifth embodiments of DSSS modulation section 44. Demultiplexer 68 (HG.3) is omitted from HG. 12 for convenience. In addition, in order to enable matrix multiplication operations discussed below, HG. 12 represents that N symbols 36 (i.e. ai$ through a^i) areprovided from demultiplexer 68 during each block 34 (HG.2). In other words, HG. 12 represents that the number (M) of symbols 36 per block 34 equals the number (N) of chips in spreading code 38. Those skilled in the art will appreciate that when M < N, the number M of symbols 3 6 per bl∞k 34 rnay be made equal to N by padding with zeros so that the zeros are evenly distributed among the symbols 36. As an example, ifM equals 4 andN equals 12, then 12 symbols 36 maybe provided by following each symbol 36 in each block 34 with two zeros.
Unspread substreams 70, which provide N symbols 3 6 per block 34, pass to an optional time-frequency domain transformation section 110. Time-fiequency domain transformation section 110 may be implemented as an inverse fast Fourier transfonn QFFT). For purposes of the present discussion, the fburih embodiment of DSSS modulation section 44 shall be deemed to omit section 110, while the fifth embodiment shall be deemed to include section 110. Thus, unspread substreams 70 convey time domain data to spreading section 72 in Hie fourth embodiment and frequency domain data to spreading section 72 in the fifth embodiment
While section 110 is not arequirement of thepresent invention, certain benefits maybe achieved by the addition of section 110 as will be discussed below. Moreover, section 110, or the equivalent, is conventionally included in digital communicalion transmitters which implαnent an orthogonal frequency division multiplexed (OFDM) modulation format Ih such situations, section πOmaybepreserjtforuseinccαmecticmwiihtherjresentw complexity or expense.
Delay elements 92 (HGs.5-6) are omitted in the fourth and fifth embodiments ofDSSS modulation section 44 to permit only symbols 36 coroirrently present during a common block 34 to influence composite signal 46. However, spreading section 72 and spreading code generating section 94 are implemented in a manner similar to that discussed above in connection with the first and second embodiments ofDSSS modulation section 44 (HGs.5-6). In particular, cyclic variations of a single common spreading code 38 are applied in the form of a cyclic Toeplitz matrix (see HG. 5). While spreading code generating section 94 acts to multiply the 1 X N matrix of symbols 36 in each block 34 by
spreading code 38 effectively in Hie form of an N X N cyclic Toeplitz matrix, it may do so simply through a one- dimensional matrix having different columns applied to different unspread substcεams 70 at different multipliers 90. Instead of selecting a spreading code 38 with low aperiodic autocorrelation sidelobes as discussed above in connection with the first, second and third embodiments ofDSSS modulation section 44, the fourth and fifth embodiments ofDSSS modulation section 44 are better served with a spreading code 3 8 having low periodic autocorrelation sidelobes and a substantially flat spectrum. Spreading code generation section 94 maybe implemented as a shift register configured to shift cyclically at the chip rate. Spread substreams 74 outout from multipliers 90 are combined man adding circuit 78 to form a pre-composite signal 46", which is converted back into parallel streams at a demultiplexer (DEMUX) 112 into N cb^perblcx±34,labeledb^1hroughbiςN.l inFIG.12.
Demultiplexer 112 provides one technique for forming a cyclic prefix 114. Chips ttø through 1^
1 aid cyclic prefix 114 are routed in parallel to inputs of a multiplexer (MUX) 116 for conversion into serial composite signal 46. In
composite signal 46", and chips b^, ... b
&m, hm
> W
b occur last in pre-composite signal 46'. FIG. 12 illustrates an example where thep =2 first-occiπiing spread substrearm that they also occur last in composite signal 46. Qf course, those skilled in the art will appreciate that the clock rate of multiplexer 116 is desirably suffideMyrngheririanfceclock chip rateto accommodate c^ 114.
Transmission section 48 forms blocks 34 of communication signal 26 from blocks 34 of composite signal 46. Blocks 34 of communication signal 26 propagate to receiver 24 Ihrough a communication channel, which may be unique to a specific transmitter 22 location and receiver 24 location. Blocks 34 of communication signal 26 experience muMpath and other types of distortion when propagating Ihrough ttiis channel. The mafliematical effect of this distortion is equivalent to multiplying composite signal 46 by the transfer function of the channel, which imposes the multipafli
As discussed above, each block 34 of composite signal 46 is formed from the matrix multiplication ofthe spreading code 38 with a block 34 of symbols 36. The effect of multipath distortion is then the matrix multiplication of thematrix expression oftlie channel transfer function with Ihis matrix product Normally, amatrixmultiplication does not observe a communicative mathematical property. In other words, the product of the channel transfer function by the spreading matrix does notnecessarily equal theproduct of spreading matrkbyte
Due to the Mure of the mathematical αmmuricative properly in matrix multiplication, normally equalization to compensate for multipath should be performed before despreading in receiver 22. Urifortunately, such equalization is exceedingly difficult to successfully perform, due at least in part to requiring the implementation of a filter with characteristics equivalent to the inverse ofthe channel transfer function. The characteristics ofthe channel cannot be easily controlled, and channel transfer fiinction quitepossibly has elements near zero. Attempting to form inverse filters of such characteristics often leads to unstable implementations.
However, the use of cyclic variations of common spreading code 38, when combined with cyclic prefix 114 and processed as discussed below in receiver 24, enables the mathematical communicative property. Consequently,
despreading may now occur prior to equalization tor mutttpafc, Ihereby making effective equalization a relatively simple task
FIG.13 showsaserøndembcx3imαtofCDMtoTDMconverte82forυse
DSSS modulation section 44 (i.e., the time domain embodiment). The digitized IF form of communication signal 26, after being distorted through the communication channel, is applied to a demultiplexer (DEMUX) 118 and a synchronization (SYNC) section 120. An output of synchronization section 120 feeds a cyclic prefix removal section 122 of demultiplexer 118. Synchronization section 120 identifies the start ofblocks 34, and cyclic prefix removal section 122 removes the fiist-occurring p chips fiom each block 34. As discussed above, the last<κxurring p chips duplicate the first-crømng p chips, and the last-occurring p chips and all other chips remain in each block 34. The fiist-occurring p chips are removed because theyareMumcedbymultipathfirmihepreviou^^ AH chips, which remain in eachblock 34, are influenced only by multipath fiom that block 34.
The block 34 of chips, with cyclic prefix 114 (FIG. 12) removed, passes to mismatched filter 100 for despreading and equalization As discussed above, due to the use of cyclic variations of spreading code 3 8 to spread symbols 3 6 and the inclusion of cyclic prefix 114, the matrix multiplication which characterizes the channel now observesthecxjmmunicativemafhematicalproperty. Consequently, despreadingmayocrurbetoreequalizatioα
Despreading may take place using a despreading code generator 124, a despreading section 126, and a combining section 128. Despreading code generator 124, despreading section 126, and combining section 128 ate identical in structure to spreading code generator 94, spreading section 72, and αambining section 78 in DSSS modulator section 44 of transmitter 22 (FIG. 12), with the despreading code generated in despreading code generator 124 being related to spreading code 38. ^particular, despreading code chips Dn = IFFT(IZFFT(Cn)), where IFFT and FFT denote inverse iastFouriertransfoirn and SstFourier teans&rm, respectively, and Cn represents the chips of spreading code 38.
Spread substreams 130 are provided by demultiplexer 118 to multipliers 132 in despreading section 126 along with tile despreading code matrix fiom despreading code generator 124. The despreading code is applied in 1he form of a cyclic Toeplitz matrix due to the use of cyclic variations of a common spreading code to which the despreading code is related. Multipliers 132 provide despread subslieams 134 to combiner 128 to add despread substreams 134 on a chip by chip basis into a serial pre-composite baseband signal 136. Pre-composite baseband signal 136 is converted into parallel symbol subsfcreams 140 at a demultiplexer (DEMUX) 13 8, and symbol substreams 140 are applied to a maximum likelihood sequence estimation (MLSE) equalizer 142 or the equivalent MLSE equalizer 142 may also be called a Viteώi equalizer. Parallel outputs fiom MLSE equalizer 142 feed amultiplexer (MUX) 144 which converts Hie parallel symbol substreams into baseband signal 84 for furlherprocessing by TDM demodulation section 86 (FIG.3).
Those skilled in the art will appreciate Ihat an MLSE equalizer is a simple structure, which is stable and can be effectively configured to compensate for multipath. The coupling of MLSE equalizer 142 downstream of despreading section 126 is possible duetotiie use of cyclic variations of αmmm spreading ccide 38 htiHismife 114 to enable matrix multiplication exhibiting the mathematical communicative property.
FlG. 14 shows a third embodiment of CDM to TDM converter 82 for use with foe fiffli embodiment of DSSS modulation section 44 (Le., the frequency domain embodiment), discussed above in connection with FIG. 12. The digitized IF form of communication signal 26, after being distorted through the communication channel, is applied to demultiplexer (DEMLDQ 118 and synchronization (SYNQ section 120, as discussed above in connection wifhFIG. 13. likewise, cyclic prefix 114 (FIG. 12) is removed at cyclic prefix removal section 122 of demultiplexer 118, as discussed above in connection withFIG. 13.
Spread substeams 130 are provided by demultiplexer 118 to a time-fiequency domain transformation section 146, which complements time-frequency domain transformation section 110 (FIG. 12). Thus, if time-fiequency domain transformation section 110 in DSSS modulation section 44 implements an inverse fast Fourier transform QFFT), then time-fiεφjencyαbmam transformation section 146 desirably implements a fast Fourier transform (FFT).
Mismatched filter 100 couples downstream of time-fiequency domain transformation section 146. In this third embodiment of CDM to TDM converter 82 mismatched filter 100 may be implemented in a manner that joins despreading and equalization functions in a common frequency domain equalizer. As illustrated in FIG. 14, coefficients for the frequency domain equalizα lake Ihe fomi D*H[n)^ spreading code 38 in the manner discussed above in connection with FIG. 13 and D*^ represents the complex conjugate of the transfer function of the channel. One reason why a frequency domain equalizer is easy and effective to implement is that coefficients ate notproportional to the inverse of the transfer function of ftie channel. While despreading code chips are related to the inverse of the FFT of the spreading code, such coefficients do not pose problems because the designer controls the FFT of the code through code selection, and a spreading code having a substantially flat spectral response may be selected, as discussed above in connection withFIG.9.
Parallel outputs of mismatched filter 100 pass in parallel to hard decision sections 148, and parallel outputs of hard decision sections 148 are combined into serial baseband signal 84 in multiplexer (MUX) 144 for further processing in TDM demodulation section 86 (FIG.3).
Due to the enabling of the mathematical rørrmunicative property for matrix multiplication discussed above, mismatched filter 100 may reside downstream of time-frequency tiaisformation section 146, which improves the efficacy and simplicity of the equalization and despreading functions.
The present invention provides an improved method and apparatus for wireless Communications. The present invention contemplates tfie combination of TDMA and spread spectrum techniques so that wireless communications capacity is increased over the capacities achievable through conventional TDMA and/or CDMA systems using an equivalent amount of spectrum. Furthermore, robust, simple, and inexpensive processing techniques are usable in the present invention, making the present invention suitable for hubs, subscriber units, mobile stations, fixed stations, portable stations, and the like. The present invention may be adapted to and used in conjunction with a variety of modulation and multiple access techniques, such as frequency division multiple access (FDMA) and orthogonal frequency division multiplexing (OFDM). The advantages and improvements of flie present invention are achieved, at least in part through
the use of a composite RF communication signal which includes signal components obtained by modulating diverse branches of a single user's data steam using cyclic variants of a common spreading code. The present invention is configured to tolerate self-interference and is better able to tolerate multipath lhan conventional DSSS-CDMA communication systems.
With reference to FIGS. 15 and 16, additional embodiments of the present invention will now be described The embodiments described below employultra-wideband commuiication technology. Referring to FIGS. 15 and 16, one type of ultra-wideband (UWB) cornmunication technology employs discrete signals of electromagnetic energy that are emitted at, for example, nanosecond or picosecond intervals (generally tens of picoseconds to hundreds of nanoseconds in duration). For this reason, this type of ultra-wideband is often called "impulse radio." That is, the UWB signals may be taansπώted without modulation onto a sine wave, or a sinusoidal carrier, in contrast with conventional carrier wave cornmunication technology. Thus, UWB generally requires neither an assigned frequency nor a power amplifier.
Anoihα example of sinusoid carrier wave commun^^ 15. IEEE8Q2.11a is a wireless local area network (LAN) protocol, which transmits a sinusoidal radio frequency signal at a 5 GHz center frequency, with a radio frequency spread of about 5 MHz. As defined herein, a carrier wave is an electromagnetic wave of a specified frequency and amplitude that is emitted by a radio tarβrrώter in order to carry information. The 802.11 protocol is an example of a carrier wave communication technology. The carrier wave comprises a substantially continuous sinusoidal waveform having a specific narrow radio frequency (5 MHz) that has a duration that may range from seconds to minutes.
M contrast, an ultra-wideband (UWB) signal may have a 2.0 GHz center frequent, with a fisquency spread of approximately 4 GHz, as shown in HG. 16, which illustrates two typical UWB signals. HG. 16illustratesfhattheshorter the UWB signal in time, the broader the spread of its frequency spectrum. This is because bandwidth is inversely proportional to the time duration of file signal A 600-picosecond UWB signal can have about a 1.8 GHz center frequency, with a frequency spread of approximately 1.6 GHz and a 300-picosecond UWB signal can have about a 3 GIfø(ra3terirequency,withairequ Thus,UWBsignalsgeneraUycfonotoperate within a specific frequency, as shown in FIG. 15. Ih addition, either of the signals shown in HG. 16 may be frequency sniffed, for example, by using heterodyning, to have essentially the same bandwidth but centered at any desired frequency. And because UWB signals are spread across an extremely wide frequency range, UWB communication systems allow communications at very high datarates, such as lOOmegabits per second or greater.
Also, because the UWB signals are spread across an extremely wide frequencyrange, the power sanpled in, tor example, a one megahertz bandwidth, is very low. For example, UWB signals of one nano-second duration and one milliwatt average power (0 dBm) spreads the power over the entire one gigahertz frequency band occupied by the signal. The resulting power density is thus 1 milliwatt divided by the 1,000 MHz signal bandwidth, or 0.001 milliwatt per megahertz (-30 dBm/MHz).
Generally, in the case of wireless communications, a multiplicity of UWB signals may be transmitted at relatively low power density (milliwatts per megahertz). However, an alternative UWB cαυmunication system may transmit at ahigher power density. For example, UWB signals may be transmittedbetween 3OdBm to -5OdBm.
Several different methods ofutoa-wideband (UWB) communications have been proposed For wireless UWB communications in the United States, all of these methods must meet the constraints recently established by the Federal CornmunicatiorB Cbmrnission (FCQ in their Report and Order issued April 22, 2002 (ET Docket 98-153). Currently, the FCC is allowing limited UWB communicatiorjs, but as UWB systems are deployed, and additional experience with this new technology is gained, the FCC may expand the use ofUWB communication technology. It will be appreciated thatthepreseπt inventionmaybe applied to current forms ofUWB communications, as well as to future variations and/or varieties ofUWB communication technology.
For example, the April 22 Report and Order requires that UWB pulses, or signals occupy greater than 20% fractional bandwidth or 500 megahertz, whichever is smaller. Fractional bandwidth is defined as 2 times the difference between the high and low 10 dB cutoff frequencies divided by the sum of the high and low 10 dB cutoff frequencies. However, these requirements for wireless UWB ccαrrmumcations in the Urited Slates rrmy change in the f^^
Communication standards committees associated with the International Institute of Electrical and Electronics Engineers (IEEE) are considering a number of ultra-wideband (UWB) wireless αmmurication methods that meet the αrrrent constraints established by the FCC. One UWB cornrnurricMonme&cdrnaytCTisrnitUWB signals that occupy 500 MHz bands within the 7.5 GEHz FCC allocation (fiom 3.1 GHz to 10.6 GHz). Ih one embodiment of this communication method, UWB signals have about a 2-nanosecond duration, which corresponds to about a 500 MHz bandwidth The center frequency of the UWB signals can bevariedtoplacefliemwhereverdesired within the 7.5 GHz allocation. In another embodiment of this communication method, an Inverse Fast Fourier Transform QFFT) is performed on parallel data to produce 122 carriers, each approximately 4.125 MHz wide. In this embodiment, also known as Orthogonal Frequency Division Multiplexing (OFDM), the resultant UWB pulse, or signal is approximately 506 MHz wide, and has a 242 nanosecond duration. It meets the FCC rules for UWB communications because it is an aggregation of manyielatively narrowband carriers rather thanbecause of the duration of each signal
Another UWB ∞rrrrnunication method being evaluated by the IEEE standards αmmitfees comprises transmitting discrete UWB signals that occupy greater than 500 MHz of frequency spectrum. For example, in one embodiment of this (xrairnunication method, UWB signal durations may vary from 2 nanoseconds, which occupies about 500 MHz, to about 133 picoseconds, which occupies about 7.5 GHz ofbandwidth. That is, a single UWB signal may occupysubstantially all ofthe entire aflc)catiαiforcorrrmιjnications(fiom3.1 GHzto 10.6GHz).
Yet another UWB communication method being evaluated by the IEEE standards committees comprises transmitting a sequence of signals that may be ar)proximately 0.7 nanoseconds or less in duration, and at a diip^ arpoximately 1.4 giga signals per second. The signals are modulated using a Direct-Sequence modulation technique, andiscalledDS-UWB. Operation in two bands is contemplated, with one band is centered near4GHzwitha 1.4GHz
wide signal, while the second band is centered near 8 GHz, with a 2.8 GHz wde UWB sigiM Operation may occur at either or both of the UWB bands. Data rates between about 28 Megabits/second to as much as 1,320 Megabitsfeecond are contemplated.
Thus, described above are three different methods of wireless ultra-wideband (UWB) communication. It will be appreciated that the present inventionmay be employ using anyone of Ihe above-describedmethods, variants ofthe above methods, or other UWB communicatimmefhods yet to be developed.
Certain features of Hie present invention may be employed by an ultra-wideband (UWB) communication system. For example, one embodiment of an UWB communication system divides a stream of data conveying symbols into a plurality of unspread substreams. A common spreading code is generated at the ultra-wideband transmitter, and each ofthe unspread substreams are spread using the common spreading code to form a plurality of spread substreams. The spread substreams are combined to fomi a composite signal that is transmitted using a plurality of discrete electromagnetic signals.
In another embodiment, an ultra-wideband communication system comprises a demultiplexer for dividing a stream of data conveying symbols into a plurality of unspread substreams. A spreading section is coupled to the demultiplexer and configured to generate a plurality of spread substreams fiom <he plurality of unspread substreams. A combining section is coupled to the spreading section and configured to form a composite signal fiomthe plurality of spread substreams, and a transmission section is coupled to the combining section and configured to transmit the composite signal over anultra-wideband communication channel.
The UWB devices, systems and/or methods in ihe above-described embodiments rømmunicate with each other by transmitϋng and receiving ultra-wideband signals, as opposed to a substantially røntinuous carrier wave. Each ulira-wideband signal may have a duration that can range between about 10 picoseconds to about 1 microsecond, and a power that may range between about +30 dBmto about -6OdBm, as measured atasinglefiequency.
The present invention may be employed in any type of network, be it wMess, wire, oramixofwire media and wireless components. That is, a network may use bo1h wire media, such as coaxial cable, and wireless devices, such as satellites, or cellular antennas. As defined herein, a network is a group of points or nodes connected by communication paths. Thecximmunicationpaftisrmyusewiresortheymaybewireless. A network as defined herein can interconnect with other networks and contain sub-networks. A network as defined herein can be characterized in terms of a spatial distance, for example, such as a local area network (LAN), apersonal area network (PAN), ametropolitan area network (MAN), a wide area network (WAN), and a wireless personal area network (WPAN), among others. A network as defined herein can also be characterized by the type of data transmission technology used by the network, such as, for example, a Transmission Control Protocol/fntemet Protocol (TCP/IP) network, a Systems Network Architecture network, among oihers. A network as defined herein can also be characterized by whether it carries voice, data, or both kinds of signals. A network as defined herein may also be characterized by users ofthe network, such as, for example, users of a public switched telephone network (PSTN) or other type of public network, and private networks (such as
wilfain a single room or home), among others. Anetwork as definedherein can also be characterized by the usual nature of its connections, for example, a dial-up network, aswitehednetwork, a dedicated network, andanon-switehednetwork, among others. A network as defined herein can also be characterized by the types of physical links that it employs, for example, optical fiber, coaxial cable, amix ofboth, unshielded twistedpair, and shielded twistedpair, among others.
The present inventionmay be employed in any type of wireless network, such as awireless PAN, LAN, MAN, or WAN. In addition, the present invention may be employed in wire media, as the present invention dramatically increases the bandwidth of conventional networks that employ wire media, such as hybrid fiber-coax cable networks, or CATV networks, yet it can be inexpensively deployed without extensive modification to the existing wire media network.
Thus, it is seen that systems and methods of ultta-wideband communications are provided One skilled in the art will appreciate 1hat the present invention can be practiced by other than Hie above-described embodiments, which are presented in this description for purposes of illustration and not of Imitation. For example, those skilled in file art will appreciate that Ihe order of time-fiequency domain transformation and spreading functions may be reversed from that shown in Fig. 12. The specification and drawings are not intended to limit tiie exclusionary scope of this patent document It is noted that various equivalents for the particular embodiments discussed in this description may practice the invention as well That is, while Represent inventionhas been described in conjunction with specific embodiments, it is evident that many alternatives, modifications, permutations and variations will become apparent to those of ordinary skill in the art in light of the foregoing description. Accordingfy, it is intended that flie present invention embrace all such alternatives, modifications and variations as fall wilhin the scope of the appended claims. The fact that a product, process or method exhibits differences from one or more of the above-described exemplary embodiments does notmean that the product or process is outside the scope (literal scope and/or other legally-recognized scope) of the following claims.