WO2006137007A1

WO2006137007A1 - Transmitter in wireless communication system

Info

Publication number: WO2006137007A1
Application number: PCT/IB2006/051973
Authority: WO
Inventors: Xuecheng Qian
Original assignee: Koninklijke Philips Electronics N.V.
Priority date: 2005-06-24
Filing date: 2006-06-20
Publication date: 2006-12-28

Abstract

The invention relates to a transmitter (200) for use in a wireless communication system. The transmitter (200) comprises a first and a second digital-to-analog converters (204, 214), for converting a first and a second digital signal to a first and a second anolog signal, the first and the second digital signal being the in-phase component and the quadrature component of a digital baseband signal, a summator (220) for adding up the first and second analog signal to form a third signal, and a band pass filter (236) suppressing harmonics in the third signal while passing the harmonic at a predetermined carrier frequency as a modulated signal. In this way, the necessary of a mixer and a 90° phase shifter are circumvented and the anolog circuits are simplified so as to reduce cost and increase flexibility of implement of a transmitter.

Description

TRANSMITTER IN WIRELESS COMMUNICATION SYSTEM

FIELD OF THE INVENTION

The invention relates to a transmitter for use in a wireless communication system, more particularly, as defined by the preamble of claim 1.

BACKGROUND OF THE INVENTION

Such a transmitter is generally known. For wireless digital communications, digital user signal is usually at first converted into analog domain and then modulated to radio frequency by one or two steps. Conventional dual up-conversion transmitter usually employs an intermediate frequency (IF) mixer and a radio frequency (RF) mixer to convert baseband user signal to IF and RF in sequence . Following the mixers, an IF filter and a RF filter are used to attenuate unwanted harmonics or interference respectively. Preferably, lowpass filter, automatic-gain-control and power amplifier are often used in order to have a clean modulated signal within certain dynamic range. The performance depends much on the linearity of the mixer used for up-conversion and the selectivity of filters. Furthermore, the cost of a transmitter depends much on the mentioned analog circuits.

In order to increase the flexibility for re-dimension and decrease the cost, direct up-conversion transmitter is widely used in recent years for multi-mode communication system. Fig. 1 is a block schematic diagram of a direct up-conversion transmitter 100. The digital users signals comprising in-phase component and quadrature component are converted to analog baseband signals by digital-to-analog converters 104 and 114, which are followed by lowpass filters 106 and 116 for removing harmonics at unwanted frequencies. Afterwards, the analog signals are modulated directly to a carrier frequency signal in IF mixers 108 and 118 by multiplying the analog signals with quadrature local oscillating signals generated by a carrier frequency local oscillator 112 and 90° phase shifter 110. The modulated signals are added up in a summator 120 to form a real signal, which is filtered by a RF filter 122 for attenuating out-of-band noise and spurious interference. Automatic gain contol 124 and PA 126 might be used to tune the signal power within certain dynamic range. After that the antenna 128 transmits the modulated signal at carrier frequency.

This transmitter architecture simplifies analog circuits and thus reduces the cost and increases flexibility of reconfiguration. Based on this architecture, an US patent US2004/0137862A1 titled "Direct-conversion transmitter circuit and transceiver system" disclosed a direct conversion type transmitter circuit suitable for a mobile communication device, which does not necessitates any high-performance low noise voltage control oscillator and RF filter, and is capable of reducing a number of components and the cost of a direct up-conversion transmitter.

However, the direct up-conversion transmitter still suffers many problems similar to conventional dual up-conversion transmitter. First of all, the existing analog RF mixer is expensive for good performance. Secondly, the lowpass filter located between digital-to-analog converter and RF mixer needs to have a low cutoff frequency and is difficult to be re-dimensioned for other modes. Besides these, there exists other problems including local oscilator pulling, leakage of local oscilator signal to output of RF mixers and I/Q phase and amplitude mismatching.

OBJECT AND SUMMARY OF THE INVENTION Amongst others it is an object of the invention to provide a transmitter for use in wireless communication system as defined in the opening paragraph with simplified analog circuits and thus further reduce cost and increase integration level of a transmitter.

To this end, the invention provides a transmitter comprising a first and a second digital-to-analog converters, for converting a first and a second digital signal to a first and a second anolog signal, the first and the second digital signal being the in-phase component and the quadrature component of a digital baseband signal, a summator and a band pass filter, characterized in that said first and second digital-to-analog converters operate at the same operating rate f_s but with a time shift τ , for keeping 90° phase difference between the harmonics at the carrier frequency in the first and second analog signal, the summator adding up the first and second analog signal to form a third signal, the band pass filter suppressing harmonics in the third signal while passing the harmonic at a predetermined carrier frequency f_c as a modulated signal, whereby the carrier frequency f_c is a multiple N of the operating rate f_s of said first and second digital-to-analog converters. In this way, the necessary of a mixer and a 90° phase shifter are circumvented and the anolog circuits are simplified thereby achieving the object of the invention.

In a preferred embodiment of the transmitter according to the invention, the time shift is selected that the time shift τ , bandwidth of the digital baseband signals B and the carrier frequency f_c satisfy τ « — and τ ^■ f_c = n + — , wherein n is an integer. In this

way, the harmonics at the carrier frequency in the first and second analog signals can keep 90° phase difference exactly to remove the need of a 90° phase shifter.

In another preferred embodiment of the transmitter according to the invention, the transmitter further comprises a lowpass filter, for suppressing harmonics at frequencies higher than the carrier frequency in the third signal. The cutoff frequency of the lowpass filter is normally a bit higher than the carrier frequency and much higher than that in a direct up-conversion transmitter and thus the lowpass filter is very easy to be implemented in integrated circuits and re- dimensioned for other modes. The invention further provides a communication system comprising a transmitter and a receiver, the transmitter being the transmitter according to the invention.

It is a further object of the invention to provide a method of modulating digital signals to a predetermined carrier frequency for use in wireless communication system as defined in the opening paragraph with simplified processing of analog circuits and thus further reduce cost and increase integration level of a transmitter.

To this end, the invention provides a method of modulating digital signal to a predetermined carrier frequency, the method comprising: converting a first and a second digital signal to a first and a second analog signal via first and second digital-to-analog converters, the first and the second digital signal being the in-phase component and the quadrature component of a digital baseband signal, characterized in that said first and second digital-to-analog converters operate at the same operating rate f_s but with a time shift τ , for keeping 90° phase difference between the harmonics at the carrier frequency in the first and second analog signal, and the method further comprisng adding up the first and second annalog signal to form a third signal and suppresing harmonics in the third signal while passing the harmonic at a predetermined carrier frequency f_c as a modulated signal, whereby the carrier frequency f_c is a multiple N of the operating rate f_s of said first and second digital-to-analog converters. In this way, the necessaries of a mixing and 90° phase shifting are circumvented and the anolog circuits processing are simplified thereby achieves the object of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS The above and other objects and features of the present invention will become more apparent from the following detailed description considered in connection with the accompanying drawings, in which:

Fig. 1 is a block schematic diagram of a conventional direct up-conversion transmitter.

Fig. 2 is a block schematic diagram of a transmitter for use in a wireless communication system in accordance with the invention.

Fig.3 is a block schematic diagram of another embodiment of a transmitter in accordance with the invention. Fig.4 is a block schematic diagram of a communication system comprising a transmitter in accordance with the invention.

Fig.5 is a flow chart illustrating a method of modulating a digital signal to a carrier frequency in accordance with the invention.

Fig.6 is a flow chart illustrating another embodiment of a method of modulating a digital signal to a carrier freqency in a transmitter provided in this invention.

In these figures identical parts are identified with identical references.

DETAILED DESCRIPTION OF THE INVENTION

Fig.2 is a block schematic diagram of a transmitter 200 for use in a wireless communication system in accordance with the invention. The in-phase component DBS-I and quadrature component DBS-Q of a digital baseband signal are passed to digital-to- analog converters 204 and 214, where they are converted to a first and a second analog signal. The first and the second analog signal are added up in a summator 220 to form a third signal having a plurality of harmonics. The third signal is passed to a bandpass filter 236, which suppresses harmonics in the third signal while passing the harmonic at a predetermined carrier frequency ( f_c ) as a wanted modulated signal (MS) to be transmitted.

To enable the function of the transmitter provided in this invention, properly selecting the operating rate of and time shift between digital-to-analog converters 204 and 214 is important. Assuming the carrier frequency predetermined for modulating digital signal is f_c and the operating rate of the two digital-to-analog converters is f_s , then the operating rate f_s should be selected in such a way that the carrier frequency f_c is multiple of f_s , that is

wherein N is an integer and a wanted modulated signal is obtainable from the N - th harmonic of the first and second analog signal. Since the impulse response of a digital-to- analog converter is a rectangular function, there are period zeros in its amplitude frequency response. Hence, N shall be selected that the N - th harmonic in the first and the second analog signal is not located at the zeros of the frequency response.

Besides the selection of the operating rate of the digital-to-analog converters, the time difference between the digital-to-analog converters 204 and 214 shall be selected properly to keep 90° phase difference between the harmonics at the carrier frequency in the analog signals outputted by the digital-to-analog converters. This is explained below in detail.

Assuming I(t) and Q(t) are time-domain in-phase component and quadrature component of a digital baseband signal to be transmitted and /(/) and Q(f) are respectively spectrum signals, then/(/ + Nf _s ) and Q(f + Nf _s ) are their N - th harmonics at the carrier frequency f_c in the first and second analog signal. Assuming all harmonics excepted the one at the predetermined carrier frequency have been suppressed by the bandpass filter 236. The signal processing could be concentrated on the harmonics at the carrier frequency.

Since digital-to-analog converters 204 and 214 operate with a small delay τ, the N - th harmonics of quadrature component is Q(f + Nf _s )e^~l2φ .

The first and second analog signal comprises harmonics at both/_c and -f_c, the sum of the in-phase and quadrature harmonics could be expressed respectively as I(f + Nf_s) - Q(f + Nf_s )e-^J2*^T and /(/ - Nf, ) - Q(f - Nf_s)e^~llφ with f_c = N ^■ f_s , the sum is further expressed as

/(/ + f_c ) - Q(f + f_c )e^~]2φ and /(/ - f_c ) - Q(f - f_c )e^~^^τ The sum of these two components obtained at the summator 220 corresponds to a real time-domain signal, which can be expressed as /(O cos(2πf_ct) - Q(t - τ) cos[2πf_c (t - τ)] In order to keep l(t)cos(2πf_ct) and Q{t - r)cos[2^f_c (t - τ)] quadrature, the time difference should be selected properly to satisfy

τ « — and τ ^■ f = n ± — B ^c A wherein n is an integer and B is bandwidth of the baseband signal and then obtain

2πf τ = 2πn + -

2

From this, 90° phase difference between l(t)cos(2πf_ct) and Q(t - τ)cos[2πf_c (t - τ)] can be generated and thus remove the necessity of 90° phase shifter in the transmitter. Addition to

the consideration of 90° phase difference, the requirement of τ « — , e.g., τ is far smaller

B than bandwidth of the baseband signal is based on the consideration of signal

approximation. That means when τ is much smaller than — , the value of ζ)(t - τ)

B approximates to ζ)(t), e.g., Q(t-τ) «<2(t). Hence, it is reasonable to rewrite the real time- domain signal as below:

/(O cos(2πf_ct) - Q(t ~ T) cos[2πf_c (t - r)] « /(O cos(2#_f 0 - 2(0 ύn(2πf_ct) It is obvious that l(t)cos(2πf /)- Q(t)ύn(2πf_ct) is exactly the wanted modulated signal at the carrier frequency f_c as expected.

Fig.3 is a block schematic diagram of another embodiment of a transmitter 300 in accordance with the invention. Besides the components comprised in transmitter 200 in fig.2 as described above, the transmitter 300 further comprises a highpass filter 232, perferably formed by an AC coupling, for removing baseband components and suppressing harmonics at frequencies lower than the carrier frequency and a lowpass filter 234 for suppressing harnomics at frequencies higher than the carrier frequency. The highpass filter 232 and lowpass filter 234 are arranged after the summator 220. They cooperate with the bandpass filter 236 for further removing or attenuating unwanted harmonics and out-of- band interference. Followed the filters, the transmitter 300 further comprises an automatic- gain-control 244, for tuning the power of the modulated signal with certain dynamic range, a power amplifier 246, for amplifying power of the modulated signal, and an antenna 248, for transmitting the modulated signal, wherein the the power amplifier 246 is arranged after the automatic-gain-control 244 and is followed by the antenna 248.

An example of the transmitter 300 for UMTS terminal is explained below. The wanted signal is supposed to be modulated in a band centered at f_c = 1992.4 MHz. When selecting N - 1 , the operating rate of digital-to-analog converters is f_s = f_c = 1992.4MHz. The duty-cycle is chosen to be 0.5, which causes a loss of about

4dB at the carrier frequency due to lowpass response of digital-to-analog converter. Besides, amplitude frequency response of the digital-to-analog converter has zeros at 2/_c and Af _c , while its attenuation on 3^rd-order harmonics at 3/_c is 13.5dB and becomes bigger at higher frequencies. The time difference between the digital-to-analog converters

204 and 214 is selected properly to keep 90° phase difference between the harmonics at the carrier frequency f_c = 1992.4 in the analog signals outputted by the digital-to-analog converters 204 and 214.

The lowpass filter 234 is chosen to be a 3^rd-order Butterworth type and its 3- dB cutoff frequency is also f_c . It has an attenuation of 28.5dB on the 3^rd-order harmonic at

3/_c . From bandpass filter 236, additional 2OdB is quite easily available and therefore total suppression on the closest harmonic, e.g., the 3^rd-order harmonic is

(13.5 -4)+ (28.5 - 3)+ 20 = 55 dB (8) which is higher than 48dB required by 3GPP specification, even when the transmitter 300 operates at the maximum power level of 24 dBm.

The fig.4 is a block schematic diagram of a communication system 10 comprising a transmitter and a receiver. The transmitter is a transmitter in accordance with the invention.

Based on above description, it can be concluded that the transmitter provided by this invention has same functions as a conventional up-conversion transmitter but it removes the needs of any mixer or 90° phase shifter, thereby simplifies the analog circuits, reduces cost and increases the flexibility of integrateion.

Fig.5 is a flow chart illustrating a method of modulating a digital signal to a carrier frequency in a transmitter provided in this invention. In step 302 of the process, a first and a second digital signal are converted to a first and a second anolog signal in the first and second digital-to-analog converters 204 and 214, the first and the second digital signal being the in-phase component and the quadrature component of a digital baseband signal. It is assumed that the first and second digital-to-analog converters operate at same operating rate f_s but with a time shift τ , keeping 90° phase difference between the harmonics at the carrier frequency in the analog signals.

In step 303, the first and second analog signal are added up in a summator 220 to form a third signal having a plurality of harmonics. It is assumed that the carrier frequency f_c is multiple N of the operating rate f_s of the first and second digital-to- analog converters 204 and 214 and the operating rate is selected in such a way that the third signal comprises harmonics at the carrier frequency and the harmonics at the carrier frequency are not located at its zeros of the amplitude frequency response of the digital-to- analog converters. In step 304, harmonics in the third signal are suppressed but the harmonic at the carrier frequency f_c is passed as a wanted modulated signal through a band pass filter 236.

Fig.6 is a flow chart illustrating another preferred method of modulating a digital signal to a carrier frequency in a transmitter provided in this invention. The processing in step 302 in fig.6 is the same as the one in fig.5.

The processing in step 303 in fig.6 is the same as the one in fig.5. In step 304', besides the same processing of step 304 in fig.5, the harmonics at frequencies higher than the carrier frequency in the third signal are further suppressed via a lowpass filter 234. The cutoff frequency of the lowpass filter is the carrier frequency, which is much higher than that of lowpass filter in a conventional direct up-conversion transmitter 100, and thus easier to be implemented as integrated circuit and re- dimensioned for other modes.

Furthermore, in step 304' the baseband components and harmonics at frequencies lower than the carrier frequency in the third signal are further suppressed via a highpass filter 232. In this case, the highpass filter is preferably formed by a coupling capacitor, which is easy to be implemented. In step 305, the power of the modulated signal is tuned within a predetermined dynamic range in an automatic-gain-control 244.

In step 306, the power of the modulated signal outputted by automatic- gain-control 244 is amplified in a power amplifier 246. In step 307, the amplified modulated signal is transmitted by an antenna 248.

Based on above description, it can be concluded that the method of modulating a digital signal to a predetermined carrier frequency provided by this invention removes the needs of mixing and 90° phase shifting and is much simplier than that in a conventional up-conversion transmitter 100, and thus can reduce cost and improve flexiblity of a transmitter.

The embodiments of the present invention described herein are intended to be taken in an illustrative and not a limiting sense. Various modifications may be made to these embodiments by those skilled in the art without departing from the scope of the present invention as defined in the appended claims.

Claims

CLAIMS:

1. A transmitter, for use in a wireless communication system, comprising: first and second digital-to-analog converters (204, 214), for converting a first and a second digital signal to a first and a second anolog signal, the first and the second digital signal being the in-phase component and the quadrature component of a digital baseband signal; a summator (220); a band pass filter; characterized in that said first and second digital-to-analog converters (204, 214) operate at the same operating rate ( f_s ) but with a time shift (τ ), for keeping 90° phase difference between the harmonics at the carrier frequency in the first and second analog signal, the summator (220) adding up the first and second analog signal to form a third signal, the band pass filter (236) suppressing harmonics in the third signal while passing the harmonic at a predetermined carrier frequency ( f_c ) as a modulated signal, whereby the carrier frequency ( f_c ) is a multiple ( N ) of the operating rate ( f_s ) of said first and second digital- to-analog converters (204, 214).

2. A transmitter as claimed in claim 2, wherein the time shift (τ ), bandwidth

of the digital baseband signals ( B ) and the carrier frequency ( f_c ) satisfy τ « — and

B

τ ■ f_c = n ± — , wherein n is an integer.

3. A transmitter as claimed in claim 1, further comprising a lowpass filter (234) arranged after the summator (220), for suppressing harmonics at frequencies higher than the carrier frequency in the third signal.

4. A transmitter as claimed in claim 3, whereby the cutoff frequency of the lowpass filter (234) is at least the carrier frequency.

5. A transmitter as claimed in claim 1, further comprising a high pass filter

(232) for removing baseband components and harmonics at frequencies lower than the carrier frequency in the third signal.

6. A transmitter as claimed in claim 5, the high pass filter (232) is formed by a coupling capacitor, arranged after the summator (220).

7. A transmitter as claimed in claim 1, further comprising an automatic-gain- control (244), for tuning the power of the modulated signal within a predetermined dynamic range, preferably, the automatic-gain-control (244)being arranged after the bandpass filter (236).

8. A tranmitter as claimed in claim 7, further comprising a power amplifier (246), for amplifying power of the modulated signal, the power amplifier being arranged after the automatic-gain-control (244).

9. A transmitter as claimed in claim 8, further comprising an antenna (248), for transmitting the modulated signal, the antenna (248) being arranged after the power amplifier (246).

10. A communication system comprising a transmitter and a receiver, characterized in that the transmitter is a transmitter as claimed in any of the preceding claims.

11. A method of modulating a digital signal to a predetermined carrier frequency, comprising: converting a first and a second digital signal to a first and a second analog signal via first and second digital-to-analog converters (204, 214), the first and the second digital signal being the in-phase component and the quadrature component of a digital baseband signal; characterized in that said first and second digital-to-analog converters operate at the same operating rate ( f_s ) but with a time shift (τ ), for keeping 90° phase difference between the harmonics at the carrier frequency in the first and second analog signal, and further comprisng adding up the first and second annalog signal to form a third signal in a summator (220) and suppresing harmonics in the third signal while passing the harmonic at a predetermined carrier frequency ( f_c ) as a modulated signal via a bandpass filter (236), whereby the carrier frequency ( f_c ) is a multiple ( N ) of the operating rate ( f_s ) of said first and second digital-to-analog converters (204, 214).

12. A method as claimed in claim 11, wherein the time shift (τ ), bandwidth of

the digital baseband signals ( B ) and the carrier frequency ( f_c ) satisfy τ « — and

B

τ - f_c = n + — , wherein n is an integer.

13. A method as claimed in claim 11, further comprising suppressing harmonics at frequencies higher than the carrier frequency in the third signal via a lowpass filter (234).

14. A method as claimed in claim 13, whereby the cutoff frequency of the lowpass filter (234) is at least the carrier frequency.

15. A method as claimed in claim 11, further comprising removing baseband components and harmonics at frequencies lower than the carrier frequency in the third signal via a high pass filter (232).

16. A method as claimed in claim 15, the high pass filter (232) is formed by a coupling capacitor.